Classifications

• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
  • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
  • H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
  • H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
• • B—PERFORMING OPERATIONS; TRANSPORTING
  • B62—LAND VEHICLES FOR TRAVELLING OTHERWISE THAN ON RAILS
  • B62D—MOTOR VEHICLES; TRAILERS
  • B62D5/00—Power-assisted or power-driven steering
  • B62D5/04—Power-assisted or power-driven steering electrical, e.g. using an electric servo-motor connected to, or forming part of, the steering gear
  • B62D5/0457—Power-assisted or power-driven steering electrical, e.g. using an electric servo-motor connected to, or forming part of, the steering gear characterised by control features of the drive means as such
  • B62D5/046—Controlling the motor
• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
  • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
  • H02P21/06—Rotor flux based control involving the use of rotor position or rotor speed sensors
• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
  • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
  • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
  • H02P25/022—Synchronous motors
  • H02P25/03—Synchronous motors with brushless excitation
• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
  • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
  • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
  • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using DC to AC converters or inverters
  • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using DC to AC converters or inverters with pulse width modulation
• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
  • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
  • H02P2205/00—Indexing scheme relating to controlling arrangements characterised by the control loops
  • H02P2205/01—Current loop, i.e. comparison of the motor current with a current reference
• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
  • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
  • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

Definitions

• the present invention relates to a motor optimal for use in an electric power steering device, an improved drive control device thereof, and an electric power steering device using the same.
• the motor used in the slave power steering device is generally a permanent magnet synchronous motor (PMSM), and the permanent magnet synchronous motor is driven by a three-phase sine wave current.
• PMSM permanent magnet synchronous motor
• vector control As a control method for driving a motor, a control method called vector control is widely used.
• brushless DC motors as motors suitable for miniaturization.
• the configuration is such that after a current command value calculation unit 100 for controlling the current of the motor 1, a subtractor 20-1, 20 for detecting an error between the current command values: tayref, Ibvref, Icvref and the motor current la, lb, Ic. -2, 20-3 and PI control unit 21 for inputting error signals from subtracters 20-1, 20-2, 20-3, and voltage va, vb, vc from PI control unit 21
• a main path to the motor 1 is connected via a PWM control unit 30 and an inverter 31 for converting DC to AC.
• a current detection circuit for detecting the motor currents Ia, lb, and Ic is provided between the inverter 31 and the motor 1. In this way, feedback circuits are provided in which the detected motor currents Ia, lb, and Ic are fed back to the subtractors 20-1, 20-2, and 20-3, respectively.
• Control system B is configured.
• a torque command value Tref calculated from a torque detected by a torque sensor (not shown) and a rotation angle 0 e of a rotor in the motor 1 detected by a position detection sensor 11 connected to the motor 1
• the electrical angular velocity ⁇ e calculated by the differentiating circuit 24 is input.
• the conversion unit 101 uses the electrical angular velocity ⁇ e and the rotation angle ⁇ e of the rotor as inputs, the conversion unit 101 calculates the back electromotive voltages ea, eb, and ec.
• the three-phase Z-two-phase converter 102 converts the d-axis component voltage ed and the q-axis component voltage eq into a d-axis component voltage ed and a q-axis component voltage eq, and the q-axis command current calculator
• the q-axis current command value Iqref is calculated.
• the current command values Iavref, Ibvref, Icvref are calculated based on the current command value Iqref from the q-axis command current calculation unit 108 and the advance angle ⁇ of the advance angle control described later. That is, the q-axis command current calculator 108 receives the ⁇ advance ⁇ calculated by the lead angle calculator 107 and the current command value Iqref, and the current command values Iav ef, Ibvref, Icvref in the two-phase / 3-phase converter 109. Is calculated.
• the motor base angular velocity co b is the limit angular velocity of the motor when the motor 1 is driven without using the field-weakening control.
• Figure 2 shows the relationship between the torque and the motor speed n (angular velocity coe), and shows an example of the limit angular velocity cob without field-weakening control.
• the current command values Iavref, Ibvref, and Icvref are independent of the advance angle ⁇ . If the control is performed according to the value calculated from the Iqref by the two-phase / three-phase converter 109, the torque according to the torque command value Tref can be output. In other words, this means that the driver's steering operation is performed smoothly as an electric power steering device.
• the feature of the advance angle control method is that the phase of the current command value Iqref is advanced by the angle ⁇ to create a field weak component.
• Iqref X sin is generated as the d-axis component
• Iqref X cos is generated as the q-axis component.
• Iqref X.s in «J> acts as a field weakening component
• Iqref X cos I> acts as a torque component.
• the controller controls the rotation magnetic field via an inverter based on the rotation position of the rotor, thereby controlling the rotation of the rotor.
• the vector control is adopted. That is, the vector control is performed by sequentially switching the excitation of each excitation coil by a control circuit according to the rotor position to a plurality of excitation coils arranged at a predetermined angle interval on the outer peripheral surface of the rotor. Is to be controlled.
• FIG. 3 is a block diagram showing an example of drive control of the motor 56 by vector control.
• a command current determination unit 51 that determines a control command value of a motor 56 is supplied to a motor 56 via a PI control unit 52, a two-phase three-phase coordinate conversion unit 53, a PWM voltage generation unit 54, and an impeller 55.
• the main route of the command signal to reach is formed.
• current sensors 571 and 572 are arranged between the inverter 55 and the motor 56, and the motor current detected by the current sensors 571 and 572 is converted into a two-phase by a three-phase / two-phase coordinate conversion unit 59, and A feedback path is formed in which the phase current components Iq and Id are knocked by the subtraction circuits 581 and 582 arranged between the command current determination unit 51 and the PI control unit 52.
• the command current determination unit 51 detects the torque command value Tref calculated from the torque detected by the torque sensor and the position detection sensor.
• the current command values Idref and Iqref are determined based on the output rotor rotation angle 0 and electrical angle ⁇ .
• These current command values I dref, I qref are feedback-corrected by the two-phase current components I q, I d converted to two phases by the three-phase Z two-phase coordinate converter 59 in the feedback path in subtraction circuits 581, 582, respectively. You. That is, the errors between the two-phase current components Id and Iq and the current command values Idref and Iqref are calculated by the subtraction circuits 581 and 582.
• the PI control units 521 and 522 calculate the signals indicating the duty of the PWM control as the command values Vd and Vq in the form of the d component and the q component.
• the component and q component are inversely transformed into three-phase components Va, Vb, and Vc.
• the inverter 55 is PWM-controlled based on the three-phase command values Va, Vb, and Vc, and an inverter current is supplied to the motor 56 to control the rotation of the motor 56.
• 61 is a vehicle speed sensor
• 62 is a sensitive area determination circuit
• 63 is a coefficient generation circuit
• 64 is a basic assist force calculation circuit
• 65 is a return force calculation circuit
• 66 is an electrical angle converter
• 67 is an angular speed converter
• 68 Denotes a non-interference control correction value calculation unit.
• the current command values I dref and Iqref are determined based on the torque command value Tref, the electrical angle ⁇ , and the rotation angle 0. Further, the feedback currents Iu, Iw of the motor 56 are converted into three-phase currents Iu, Iv, Iw, and then converted into two-phase current components Id, Iq, and thereafter, are subtracted by the subtraction circuits 582 and 581. An error between the two-phase current components Id and Iq and the current command values Idref and Iqref is calculated, and the error is used to execute the current control by the PI control, so that the command values Vd and Vq to the Is required.
• the command values Vd, Vq are inversely converted again into the three-phase command values Va, Vb, Vc by the two-phase / three-phase coordinate conversion part 53, and the inverter 55 is controlled to control the driving of the motor 56. It has become.
• the d-axis component and the q-axis component generated by the advance angle control only advance the current command value Iqref by the phase ⁇ , the d-axis Iqref X sin and the q-axis Iqref X cos! Therefore, the quantitative balance is not always optimized.
• the motor terminal voltage saturates during high-speed rotation, the motor current cannot follow the current command value, the torque ripple increases, and the motor noise increases. For this reason, the electric power steering apparatus has problems such as an abnormal vibration felt through the handle and an uncomfortable feeling of the driver due to the motor noise when the handle is rapidly steered.
• the detected current of the motor 56 and the output of the inverter 55 are three-phase, and the feed pack control system is two-phase. In this way, it is necessary to drive and control the motor 56 by performing the inverse conversion from the two-phase to the three-phase again in the two-phase Z-three-phase coordinate conversion unit 53. Therefore, there is a problem that the entire control system becomes complicated.
• the force S and the drive control of the motor 56 include various factors of non-linearity.
• One of the causes of the non-linearity of the motor drive is, for example, inverter control dead time.
• FETs are used as switching elements of inverters.
• FETs are not ideal switching elements.
• the motor current generated by the switching of the FET having such a dead time includes a nonlinear element in a switching transient state.
• a non-linear element is included in a detection element or a detection circuit for detecting a motor current. This means that, for example, the nonlinear element generated in the a-phase current la is converted into the d-axis current component I d and the q-axis current component I by the d-q conversion in the three-phase / two-phase coordinate converter 59 of the feedback system. contained in q.
• the feedback current control is calculated in two phases, and the command values Vd and Vq determined in two phases are formally formed, even though the motor is driven in three phases. Because the control is performed by converting the three-phase command values to Va, Vb, and Vc, the non-linear elements diffuse.
• the present invention has been made in view of the circumstances described above, and an object of the present invention is to control a nonlinear element included in motor control in a state where the nonlinear elements are separated into respective phases, thereby reducing torque ripple and reducing noise and noise.
• an object of the present invention is that even when the motor is rotating at high speed, the motor terminal voltage is not saturated, the torque ripple is small, the motor noise is small, and in the case of the electric power steering device, even when the handle is rapidly steered, It is an object of the present invention to provide a motor drive control device and an electric power steering device that generate low noise and can smoothly follow a steering operation. Disclosure of the invention
• the present invention also relates to a motor drive control device that controls a motor having three or more phases, and a vector control phase command value calculation unit that calculates a phase current command value for each phase of the motor using vector control.
• a motor current detection circuit that detects a motor phase current of each phase of the motor; and a current control unit that controls a phase current of the motor based on the phase current command value and the motor phase current.
• the vector control phase command value calculation unit calculates each phase back electromotive voltage by each phase back electromotive voltage calculation unit, and calculates the d axis and the q axis component of the back electromotive voltage from each phase back electromotive voltage.
• a d-q voltage calculator for calculating certain voltages ed and eq; a q-axis command current calculator for calculating a current command value Iqref which is a q-axis component of the current command value from the voltages ed and eq; a current command value
• a d-axis command current calculation unit for calculating a current command value I dref which is a d-axis component of This is achieved by having a phase current command calculator for calculating a phase current command value for each phase from the command values iqref and Idref.
• phase current command values Iavref, Ibvref, and Icvref are calculated by using the current command values Idref, Iqref and a constant that depends on the rotation angle 0 e of the motor. .
• the object of the present invention is that the current control circuit includes an integral control, or that the motor is a brushless DC motor, or that the current of the motor is a rectangular wave or a quasi-rectangular wave. This is achieved more effectively by an electric power steering device using the motor drive control device.
• the present invention relates to a motor drive control device that controls a motor current based on a current command value Idref and an Iqref calculated using vector control, wherein the detected mechanical angular velocity com of the motor is
• the current command value Idref can be calculated from the torque command value Tref of the motor, the base angular speed ⁇ b, and the mechanical angular speed ⁇ m.
• the object of the present invention is that the current command value Idref is obtained as a function of the torque command value Tref and sin ⁇ , and the advance angle ⁇ is derived from the base angular velocity cob and the mechanical angular velocity com.
• the torque command value Iqref is calculated by substituting the torque command value I dref into a motor output equation, or by the motor current of the brushless DC motor being a rectangular wave current or a quasi-rectangular wave current. Achieved more effectively.
• FIG. 1 is a control block diagram based on conventional advance angle control.
• FIG. 2 is a diagram showing a base angular velocity which is a limit angular velocity when field weakening control is not used.
• FIG. 3 is a control block diagram showing a control method of the conventional vector control.
• FIG. 4 is a sectional structural view showing an example of a brushless DC motor to be controlled by the present invention.
• FIG. 5 is a diagram showing the principle of rotor position detection.
• FIG. 6 is a diagram for explaining the definition of trapezoidal wave current (voltage).
• FIG. 7 is a diagram showing an example of an induced voltage waveform (rectangular wave).
• FIG. 8 is a block diagram showing an example of a control system of the brushless DC motor according to the present invention.
• FIG. 9 is a block diagram showing a configuration example of calculating a current command value Idref according to the field weakening control of the present invention.
• FIG. 10 is a diagram showing a vector relationship between a current command value Idref and a current command value Iqref according to the control method of the present invention and the conventional advance control method.
• the three-phase brushless DC motor 1 is arranged along a cylindrical housing 2 and the axis of the housing 2, and is rotated by bearings 3a and 3b at upper and lower ends.
• a rotating shaft 4 freely supported, a motor driving permanent magnet 5 fixed to the rotating shaft 4, and fixed to the inner peripheral surface of the housing 2 so as to surround the permanent magnet 5;
• the rotor 7 includes a stator 6 around which 6a, 6b and 6c are wound, and a rotating shaft 4 and a permanent magnet 5 constitute a rotor 7.
• a ring-shaped permanent magnet 8 for phase detection is fixed, and the permanent magnet 8 is alternately magnetized to the S pole and the N pole at equal intervals in the circumferential direction.
• a support substrate 10 made of a ring-shaped thin plate is provided via a stay 9 on the end surface of the housing 2 on the side where the bearing 3b is provided.
• a rotor position detector 11 such as a resolver or an encoder is fixed to the support substrate 10 so as to face the permanent magnet 8.
• the rotor position detector 1 1 as shown in FIG. 5.
• the exciting coils 6a to 6c are disposed so as to surround the outer peripheral surface of the rotor 7 at an electrical angle of 120 degrees, and the coil resistances of the exciting coils 6a to 6c are all equal. Has become.
• the rotor position detector 11 outputs a position detection signal according to the magnetic pole of the opposing permanent magnet 8.
• the rotor position detector 11 detects the rotation position of the rotor 7 by utilizing the change caused by the magnetic pole of the permanent magnet 8.
• the vector control phase current command value calculation unit 20 which will be described later, sequentially energizes the excitation coils 6a to 6c one phase at a time while energizing two phases simultaneously to the three-phase excitation coils 6a to 6c.
• the rotor 7 is driven to rotate by the switching two-phase excitation method.
• the drive control of the motor 1 is controlled using a rectangular wave current or a pseudo-rectangular wave current as a motor current or a rectangular wave voltage or a pseudo-rectangular wave voltage as a motor induced voltage.
• the control with the rectangular wave voltage or pseudo-rectangular wave current or the induced voltage square wave voltage or pseudo-rectangular wave voltage is the same as the sine wave current or sine wave voltage when the current peak value or voltage peak value is the same. If so, square wave This is because a larger output value (power) can be obtained because the effective value of the current or the rectangular wave voltage is larger.
• a rectangular wave current or pseudo-rectangular wave current as the motor current or a rectangular wave voltage or pseudo-rectangular wave voltage as the motor induced voltage can reduce the size of the motor.
• control using a rectangular wave current or a pseudo-rectangular wave current or an induced voltage using a rectangular wave voltage or a pseudo-rectangular wave voltage reduces the torque ripple as compared with control using a sine wave current or a sine wave voltage.
• control using a sine wave current or a sine wave voltage There is a disadvantage that it is difficult.
• Fig. 6 shows an example of the motor current waveform controlled by the current (Id) control.
• Fig. 6 (B) shows the motor 1 The figure shows the motor current waveform when field-weakening control by current (Id) control is performed at high speed.
• Fig. 6 (A) shows the motor current waveform, and the corresponding induced voltage waveform is a rectangular (trapezoidal) wave as shown in Fig. 7 (A).
• the actual current waveform for [A] is shown in Fig. 7 (C).
• a square wave current or a square wave voltage as referred to in the present invention is different from a perfect square wave (trapezoidal wave) and is different from a concave portion shown in FIG. 6 (A) or FIG. B) or a waveform having a peak as shown in Fig. 7 (C), or a current waveform (quasi-rectangular wave current) or a voltage waveform (quasi-rectangular wave voltage) as shown in Fig. 7 (A). .
• the order wave component n of 5% or more of the amplitude component is expressed by the following equation (3), n XPZ 2 X 0) ⁇
• P is the number of poles and c is the actual speed.
• an angle sensor is provided, and a current waveform is given at least as a function of an induced voltage waveform of a rectangular wave or a pseudo-rectangular wave.
• the electrical time constant of the motor correlation may be equal to or longer than the control cycle, and an angle estimating means may be provided so that the motor current waveform is given by the estimated angle from the angle estimating means.
• the reason for setting the above equation (3) for the order wave component n of 5% or more of the amplitude component is as follows.
• the current command value is multiplied by an order wave component n that cannot be responded by the current control unit, it appears as a motor torque ripple. It is known that if the torque ripple of the motor is within 10%, the torque is not felt by the steering wheel in the torque control system (for example, Japanese Patent No. 3298006 (B2)). Therefore, the higher-order wave component of the back electromotive voltage can be determined so that the current value (torque) is 10% or less.
• PWM control is usually performed at 20 KHz.
• the frequency is lower than 20 KHz, motor noise is a problem, and when the frequency is higher than 20 KHz, there is a problem of electromagnetic radiation noise and heat generation. This depends on the performance of the FET as a driving means.
• the upper limit of the current control response frequency is 1/20 / 1000Hz
• 40KHz PWM control 1/20 / 2000Hz is the current limit. This is the upper limit of the control response frequency.
• the present invention constitutes a motor drive control device as shown in FIG.
• the motor drive control device includes a vector control phase current command value calculation unit 20, current command values Iavref, Ibvref, and Icvref from the vector control phase current command value calculation unit 20.
• a PI control unit 21 for performing the following operations.
• Each phase command current is supplied from the inverter 31 to the motor 1 by the PWM control of the PWM control unit 30 to control the rotational driving of the motor 1.
• Area A shown by the broken line constitutes the current control unit.
• the vector control phase command value calculation circuit 20 determines the current command values of the vector control d and q components using the excellent characteristics of the vector control, and then calculates the current command values.
• the feed pack control unit is configured to close all phases using phase control instead of d and q control. Therefore, since the theory of vector control is used at the stage of calculating the current command value, this control method is called pseudo vector control (hereinafter, referred to as “PVC control”).
• the current control unit A of the present embodiment includes subtraction circuits 20-1, 20 for calculating each phase current error from the phase current command values Iavref, Ibvref, Icvref of the motor 1 and the motor phase currents Ia, lb, Ic. -2, 20-3, and a PI control unit 21 that receives the phase current error of each phase. Also, between the inverter 31 and the motor 1, current detection circuits 32-1, 32-2, 32-3 are arranged as motor current detection circuits, and the current detection circuits 32-1, 32-2, 32-3 A feed-pack circuit B is provided that supplies the motor phase currents Ia, lb, and Ic detected in step (1) to the subtraction circuits 20-1, 20-2, and 20-3.
• the vector control phase current command value calculation unit 20 includes a conversion unit 101 as each phase back electromotive voltage calculation unit, a three-phase / two-phase conversion unit 102 as d-axis and q-axis voltage calculation units, and q Q-axis command current calculation unit 103 for calculating axis current command value Iqref, 2-phase to 3-phase conversion unit 104 as each phase current command calculation unit, and d-axis current command value for calculating d-axis current command value Idref A calculation unit 105; and a conversion unit 106 for converting the base angular velocity cob of the motor from the torque command value Tref.
• the rotation angle 0 e of the rotor 7 detected by the rotor position detector 11 such as a resolver, and the rotation angle 0 e is received by the rotor position detection signal composed of the electric angular velocity co e calculated by the differentiating circuit 24 and the torque command value Tref determined based on the torque detected by the torque sensor (not shown).
• a command value signal is calculated.
• the rotor position detector 11 has a function as an angle sensor, and can be replaced with angle estimation means.
• Torque command value Tref is input to q-axis command current calculation unit 103, conversion unit 106 and d-axis command current calculation unit 105, and rotation angle 0 e is converted unit 101, 3-phase / 2-phase conversion unit 102 and 2-phase.
• the electrical angular velocity ce is input to the three-phase conversion unit 104, and is input to the conversion unit 101, the q-axis command current calculation unit 103, and the d-axis command current calculation unit 105.
• the drive control of the motor 1 is performed as follows.
• the vector control phase current command value calculation unit 20 inputs the rotation angle 0 e of the rotor 7 and the electrical angular velocity co e to the conversion unit 101, and inverts each phase based on the conversion table stored in the conversion unit 101.
• the electromotive voltages ea, eb, and ec are calculated.
• the back electromotive voltages ea, eb, and ec are rectangular or pseudo-rectangular waves of the nth harmonic, and the frequency of the nth harmonic is obtained by multiplying the electric angular velocity of the motor by n.
• the electric angular velocity of the motor is expressed as ⁇ ⁇ 2 ⁇ ⁇ ⁇ ⁇ , where ⁇ is the actual motor speed.
• the back electromotive voltages ea, eb, and ec are calculated by the three-phase Z two-phase converter 102 as a d-q voltage calculator based on the following equations (4) and (5). Converted to component voltages ed and eq.
• d-axis current command value Idref is based angular velocity omega b of the conversion unit 106, the electrical angular velocity coe from the differentiation circuit 24, the d-axis command current calculating unit IO 5 as input the torque command value Tref from the torque sensor following (6 ) Calculated according to the formula.
• Kt is the torque coefficient
• cob is the base angular velocity of the motor
• the base angular velocity cob is obtained by the conversion unit 106 by inputting the torque command value Tref.
• Idref ⁇ I Tref / Kt
• Idref 0.
• the current command value Idref changes depending on the rotation speed of the motor 1, there is an excellent effect that control during high-speed rotation can be performed smoothly without any connection. .
• Motor phase voltage V is generally
• V E + R ⁇ I + L (di / dt) ⁇ (7)
• E is the back electromotive voltage
• R is the fixed resistance
• L is the inductance
• the back electromotive voltage E increases as the motor rotates at a higher speed
• the power supply voltage such as the battery voltage is fixed.
• the voltage range that can be used for controlling the power supply becomes narrow.
• the angular velocity at which this voltage saturation is reached is the base angular velocity ⁇ b.If voltage saturation occurs, the duty ratio of the PWM control reaches 100%, and beyond that the current command value cannot be followed, resulting in a large torque ripple. .
• the current command value Idref expressed by the equation (6) has a negative polarity
• the induced voltage component of the current command value Idref for L (di / dt) in the above equation (6) has a polarity opposite to the back electromotive voltage E.
• the effect of reducing the back electromotive voltage E which becomes larger as the rotation speed increases, is reduced by the voltage induced by the current command value Idref.
• the voltage range in which the motor can be controlled by the effect of the current command value Idref is widened.
• FIG. 9 shows a block configuration of a circuit related to the calculation of the above-described current command value Idref.
• the torque command value Tref is input to the conversion unit 106 and the torque coefficient unit 105d, and the electric angular velocity ⁇ e of the motor is input to the mechanical angle calculation unit 105a.
• the sin calculation unit 105c calculates a sin from the input advance angle and inputs it to a multiplier 105f that multiplies the value by 11, and the multiplier 105f receives the advance angle ⁇ from the sin calculation unit 105c and the advance angle ⁇ from the absolute value unit 105e. Multiply by the absolute value I Iqb I and multiply by 11 to obtain the current command value Idref.
• the current command value Idref is obtained by the following equation (8), and this is the output of the d-axis command current calculation unit 105.
• Idref one I Iqb
• the current command value Idref calculated according to the above equation (8) is calculated by the q-axis command current calculator 103 and the two-phase three-phase converter 104. Is input to
• the current command value Iqref is derived from the motor output equation that the motor output is equivalent to the electric power, so that the current command value Iqref can be easily calculated. Further, it is possible to calculate the current command value Idref for obtaining the required command torque Tref and the optimal current command value Iqtef balanced. Therefore, even when the motor rotates at high speed, the terminal voltage of the motor does not saturate, and control that minimizes torque ripple becomes possible.
• FIG. 10 (A) shows the relationship between the current command values Idref and Iqref of the present invention as described above.
• Fig. 10 (B) shows the relationship in the case of the conventional advance control method.
• the current command values Idref and Iqref are input to a two-phase / three-phase conversion unit 104 as a current command value calculation unit for each phase, and are converted into current command values Iavref, Ibvref, and Icvref for each phase. That is, they are expressed as in equations (12) and (13).
• the suffix is, for example, “avref ⁇ ” of the current command value Iavref represents the current command value of the a-phase determined by the vector control.
• the determinant C 2 is given by the following equation (13). As shown, it is a constant determined by the rotation angle 0 e of the motor. la ref
• the current command values lavref, Ibvref, and Icvref were calculated by the two-phase / three-phase conversion unit 109 in FIG. 1 using the current command value Iqref and the advance angle ⁇ , but in the present invention, the current With the command values Idref and Iqref as inputs, the two-phase / three-phase converter 104 calculates the current command values lavref, Ibvref, and Icvref. Then, subtraction circuits 20-1, 20 and 20 subtract each phase current Ia, lb, Ic of the motor detected by the current detection circuits 32-1, 32-2, 32-3 from the current command values lavref, Ibvref, Icvref. Subtract by -2, 20-3 to calculate each error.
• the error of each phase current is controlled by the PI control unit 21 to calculate the command value of the inverter 31, that is, the voltage values va, vb, vc representing the duty of the PWM control unit 30, and the voltage values va, vb,
• the motor 1 is driven by the PWM control unit 30 performing PWM control of the inverter 31 based on vc, and a desired torque is generated.
• the motor and the drive control device of the present invention do not saturate the terminal voltage of the motor even during high-speed rotation of the motor, and can perform control to minimize the torque ripple. Therefore, when the present invention is applied to an electric power steering device, rapid steering can be smoothly performed. It has the excellent effect of not giving the driver any uncomfortable feeling such as handle vibration.
• the present invention is completely different from the conventional feed-pack control by d and q control in that the feed-pack control is performed only by each phase control.
• the nonlinear element generated in the a-phase is dispersed to each of the ', b, and c phases in the process of performing the feedback control by the conventional d and q control, and there is a problem that the corrective control cannot be performed.
• the non-linear elements of the a-phase are subjected to the feedback control only in the a-phase, and are not dispersed in the b-phase and the c-phase.
• the motor can be controlled with the non-linear elements included in the control separated into each phase, and as a result, a motor control with small torque ripple and low noise can be performed. For this reason, when applied to an electric power steering device, a handle operation with low noise and smooth vibration with little vibration becomes possible even during parking or emergency steering.
• the phase voltages ea, eb, and ec are used. However, the same effect can be obtained by converting the voltages into line voltages eab, ebc, and eca.
• the motor of the present invention even when the motor is rotating at high speed, the terminal voltage of the motor is not saturated, the torque ripple is small, and the motor noise is small.
• the steering device there is an excellent effect that an electric power steering device can be provided which can smoothly follow the rapid steering of the steering wheel, has no uncomfortable feeling in the steering wheel operation, and has low noise.
• the brushless DC motor is calculated by calculating the current command value for each phase based on the vector control, and using the PVC control for controlling the current feedback individually for each phase. Control to reduce torque ripple and motor noise. It is possible to provide a motor drive control device capable of performing the above operation, and to provide an electric power steering device with smooth handle operation and low noise.
• the motor of the present invention since the frequency of the n-th harmonic is equal to or lower than the upper limit value of the response frequency of the current control, the motor is driven by a rectangular wave current or a quasi-rectangular wave current, or a rectangular wave voltage or a quasi-rectangular wave voltage. Even so, the torque ripple is small, small, and low noise.
• the present invention even when the motor rotates at high speed, the terminal voltage of the motor does not saturate, the torque ripple is small, and the motor noise is small. Therefore, when applied to the electric power steering device, the steering wheel can be smoothly steered. It is possible to provide an electric power steering device that follows the operation of the steering wheel without feeling uncomfortable and with low noise.
• the brushless DC motor is reduced in size by calculating the current command value for each phase based on the vector control and using the PVC control for controlling the current feedback individually for each phase.
• a motor drive control device capable of controlling the torque ripple and the motor noise to be small, and to provide an electric power steering device with smooth handle operation and low noise.

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• Engineering & Computer Science (AREA)
• Power Engineering (AREA)
• Chemical & Material Sciences (AREA)
• Combustion & Propulsion (AREA)
• Transportation (AREA)
• Mechanical Engineering (AREA)
• Control Of Ac Motors In General (AREA)
• Control Of Motors That Do Not Use Commutators (AREA)
• Power Steering Mechanism (AREA)
• Valve Device For Special Equipments (AREA)
• Vehicle Body Suspensions (AREA)
• Control Of Electric Motors In General (AREA)
• Steering Control In Accordance With Driving Conditions (AREA)
• Electrical Control Of Air Or Fuel Supplied To Internal-Combustion Engine (AREA)

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• 2003
  • 2003-11-06 JP JP2003376428A patent/JP2004201487A/ja not_active Withdrawn
  • 2003-11-27 AT AT03775921T patent/ATE508528T1/de not_active IP Right Cessation
  • 2003-11-27 KR KR1020057009232A patent/KR20050083958A/ko not_active Ceased
  • 2003-11-27 US US10/536,990 patent/US7339346B2/en not_active Expired - Lifetime
  • 2003-11-27 AU AU2003284466A patent/AU2003284466A1/en not_active Abandoned
  • 2003-11-27 WO PCT/JP2003/015166 patent/WO2004049554A1/ja not_active Ceased
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• 2007
  • 2007-09-25 US US11/860,873 patent/US7463006B2/en not_active Expired - Fee Related
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