US7848118B2 - Bi-directional DC-DC converter and method for controlling the same - Google Patents

Bi-directional DC-DC converter and method for controlling the same Download PDF

Info

Publication number
US7848118B2
US7848118B2 US12/193,244 US19324408A US7848118B2 US 7848118 B2 US7848118 B2 US 7848118B2 US 19324408 A US19324408 A US 19324408A US 7848118 B2 US7848118 B2 US 7848118B2
Authority
US
United States
Prior art keywords
switching element
switching
power source
lower arm
switching elements
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Expired - Fee Related, expires
Application number
US12/193,244
Other languages
English (en)
Other versions
US20090059622A1 (en
Inventor
Takae Shimada
Yoshihide Takahashi
Kimiaki Taniguchi
Hiroyuki Shoji
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Hitachi Information and Telecommunication Engineering Ltd
Original Assignee
Hitachi Computer Peripherals Co Ltd
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Hitachi Computer Peripherals Co Ltd filed Critical Hitachi Computer Peripherals Co Ltd
Assigned to HITACHI COMPUTER PERIPHERALS CO., LTD. reassignment HITACHI COMPUTER PERIPHERALS CO., LTD. ASSIGNMENT OF ASSIGNORS INTEREST (SEE DOCUMENT FOR DETAILS). Assignors: TAKAHASHI, YOSHIHIDE, TANIGUCHI, KIMIAKI, SHIMADA, TAKAE, SHOJI, HIROYUKI
Publication of US20090059622A1 publication Critical patent/US20090059622A1/en
Priority to US12/896,176 priority Critical patent/US7911810B2/en
Application granted granted Critical
Publication of US7848118B2 publication Critical patent/US7848118B2/en
Priority to US13/024,434 priority patent/US8130515B2/en
Assigned to HITACHI INFORMATION & TELECOMMUNICATION ENGINEERING, LTD reassignment HITACHI INFORMATION & TELECOMMUNICATION ENGINEERING, LTD MERGER (SEE DOCUMENT FOR DETAILS). Assignors: HITACHI COMPUTER PERIPHERALS CO., LTD.
Expired - Fee Related legal-status Critical Current
Adjusted expiration legal-status Critical

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33584—Bidirectional converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to a bi-directional DC-DC converter having an insulating function and a method for controlling such a converter.
  • the hybrid automobile has a main battery for driving a motor and a battery for auxiliaries. If an electric power can be mutually supplied between the two batteries of different voltages, design flexibility of a vehicle power supply systems can be increased.
  • a bi-directional DC-DC converter for bi-directionally converting electric powers between two power sources of different voltages has been disclosed in JP-A-2002-165448.
  • a voltage type circuit on a high voltage side and a current type circuit on a low voltage side having a choke coil are connected through a transformer.
  • the electric power is supplied from the high voltage side power source to the low voltage side power source.
  • the electric power is supplied from the low voltage side power source to the high voltage side power source.
  • a bi-directional DC-DC converter in which a voltage clamping circuit including an object connection in series of a switching element and a capacitor is connected to a low voltage side circuit has been disclosed in JP-A-2006-187147.
  • a loss caused by a circulating current is reduced by the voltage clamping circuit.
  • a high efficient and small bi-directional DC-DC converter in which, at the time of the step-up/down operations, the generation of a surge voltage in the low voltage side circuit is prevented and a withstanding voltage of the switching element is reduced is provided.
  • Another object of the invention is to provide a small bi-directional DC-DC converter in which a large electric power can be outputted even at the time of a step-down operation.
  • a bi-directional DC-DC converter which comprises a high voltage side switching circuit, connected between a first DC power source and a transformer, for executing an electric power conversion between a direct current and an alternating current, a low voltage side switching circuit, connected between a second DC power source and the transformer, for executing an electric power conversion between a direct current and an alternating current, and a control circuit for controlling ON/OFF of switching elements included in each of the switching circuits and in which an electric power is transmitted and received between the first and second DC power sources, wherein the high voltage side switching circuit includes a first vertical arm in which a first upper arm switching element and a first lower arm switching element are serially connected, a second vertical arm in which a second upper arm switching element and a second lower arm switching element are serially connected, a first smoothing capacitor connected in parallel to the first and second vertical arms and the first DC power source, and a series connector which is connected between a series node of the first upper arm switching element and the first lower arm switching
  • control circuit further includes means for switching the first lower arm switching element and the second upper arm switching element to ON while keeping the switching elements switched to ON by the first switching means in the ON state and, after a direction of a current flowing in the primary winding was reversed, switching the switching elements in the ON state to OFF.
  • control circuit further includes: third switching means for switching the switching elements in the OFF state in the second switching element group to ON for a period of time during which both of the first lower arm switching element and the second upper arm switching element are in the ON state; and fourth switching means for switching the first upper arm switching element and the second lower arm switching element to ON while keeping the switching elements switched to ON by the third switching means in the ON state.
  • control circuit further includes means for switching the first upper arm switching element and the second lower arm switching element to ON while keeping the switching elements switched to ON by the third switching means in the ON state and, after a direction of a current flowing in the primary winding was reversed, switching the switching elements in the ON state to OFF.
  • a voltage which is substantially twice as large as the first DC power source is applied to the series connector of the resonance reactor and the resonance capacitor for a period of time during which the switching elements switched to ON by the first or third switching means are in the ON state.
  • a bi-directional DC-DC converter which comprises a high voltage side switching circuit, connected between a first DC power source and a transformer, for executing an electric power conversion between a direct current and an alternating current, a low voltage side switching circuit, connected between a second DC power source and the transformer, for executing an electric power conversion between a direct current and an alternating current, and a control circuit for controlling ON/OFF of switching elements included in each of the switching circuits and in which an electric power is transmitted and received between the first and second DC power sources, wherein the high voltage side switching circuit includes a first vertical arm in which a first upper arm switching element and a first lower arm switching element are serially connected, a second vertical arm in which a second upper arm switching element and a second lower arm switching element are serially connected, a first smoothing capacitor connected in parallel to the first and second vertical arms and the first DC power source, and a series connector which is connected between a serial node of the first upper arm switching element and the first lower arm switching element and a serial no
  • the control means switches the upper or lower arm switching element in the ON state to OFF and, after a direction of a current flowing in the primary winding was reversed, switches the switching elements in the ON state in the first switching element group to OFF, and switches a state of the energy in the smoothing reactor from the accumulation to the emission.
  • control means controls a time which is necessary until the upper or lower arm switching element in the ON state is switched to OFF and the energy is supplied to the first DC power source after the switching elements in the ON state in the first switching element group were switched to OFF, thereby adjusting an amount of energy which is supplied to the first DC power source.
  • control circuit further includes: means for switching two of the first and second upper or lower arm switching elements to ON for the period of time during which the energy is supplied to the first DC power source; means for switching one of the upper and lower arm switching elements in the ON state to OFF after a direction of a current flowing in the primary winding was reversed; and means for switching the upper or lower arm switching elements in the ON state to OFF after the switching elements in the ON state in the first switching element group were switched to OFF and a state of the energy in the smoothing reactor was switched from the accumulation to the emission.
  • the first switching element group includes fifth to eighth switching elements
  • the second switching element group includes a ninth switching element
  • the voltage clamping circuit includes a series connector of the ninth switching element and the clamping capacitor
  • the low voltage side switching circuit includes a third vertical arm in which the fifth and sixth switching elements are serially connected and a fourth vertical arm in which the seventh and eighth switching elements are serially connected and is constructed in such a manner that the secondary winding is connected between a serial node of the fifth and sixth switching elements and a serial node of the seventh and eighth switching elements, the third and fourth vertical arms and the voltage clamping circuit are connected in parallel, one end of the smoothing reactor is connected to one end of the voltage clamping circuit, one end of the second smoothing capacitor is connected to the other end of the smoothing reactor, and the other end of the voltage clamping circuit is connected to the other end of the second smoothing capacitor.
  • the first switching element group includes fifth and sixth switching elements
  • the second switching element group includes seventh and eighth switching elements
  • the voltage clamping circuit is constructed by connecting one end of each of the seventh and eighth switching elements and one end of the clamping capacitor
  • the secondary winding has a connector of one end of a first secondary winding and one end of a second secondary winding
  • the low voltage side switching circuit is constructed in such a manner that one end of the fifth switching element and the other end of the seventh switching element are connected to the other end of the first secondary winding, one end of the sixth switching element and the other end of the eighth switching element are connected to the other end of the second secondary winding, the other end of the fifth switching element and the other end of the sixth switching element are connected to the other end of the clamping capacitor, and a series connector of the smoothing reactor and the second smoothing capacitor is connected between a node of the fifth and sixth switching elements and a node of the first and second secondary windings.
  • the first switching element group includes fifth and sixth switching elements
  • the second switching element group includes seventh and eighth switching elements
  • the voltage clamping circuit is constructed by connecting one end of the seventh switching element, one end of the eighth switching element, and one end of the clamping capacitor
  • the smoothing reactor is constructed by connecting one end of a first smoothing reactor and one end of a second smoothing reactor
  • the low voltage side switching circuit is constructed in such a manner that one end of the fifth switching element, the other end of the seventh switching element, and the other end of the first smoothing reactor are connected to one end of the secondary winding
  • one end of the sixth switching element, the other end of the eighth switching element, and the other end of the second smoothing reactor are connected to the other end of the secondary winding of the transformer
  • the other end of the fifth switching element and the other end of the sixth switching element are connected to the other end of the clamping capacitor
  • the second smoothing capacitor is connected between a node of the fifth and sixth switching elements and a node of the first and second smoothing reactors.
  • the clamping capacitor and the second smoothing capacitor are connected.
  • the resonance reactor has a leakage inductance and a wiring inductance of the transformer and includes first, second, and third resonance reactors which are respectively serially connected to the primary winding and the secondary winding and are magnetically coupled therewith, and the resonance capacitor includes first, second, and third resonance capacitors which are respectively serially connected to the primary winding and the secondary winding.
  • each of the switching elements has diodes which are connected in inverse parallel and snubber capacitors which are connected in parallel.
  • a control method of controlling a DC-DC converter which comprises a high voltage side switching circuit, connected between a first DC power source and a transformer, for executing an electric power conversion between a direct current and an alternating current, a low voltage side switching circuit, connected between a second DC power source and the transformer, for executing an electric power conversion between a direct current and an alternating current, and a control circuit for controlling ON/OFF of switching elements included in each of the switching circuits and in which the high voltage side switching circuit includes a first vertical arm in which a first upper arm switching element and a first lower arm switching element are serially connected, a second vertical arm in which a second upper arm switching element and a second lower arm switching element are serially connected, a first smoothing capacitor connected in parallel to the first and second vertical arms and the first DC power source, and a series connector which is connected between a serial node of the first upper arm switching element and the first lower arm switching element and a serial node of the second upper arm switching element and the second lower arm switching element
  • a control method of controlling a DC-DC converter which comprises a high voltage side switching circuit, connected between a first DC power source and a transformer, for executing an electric power conversion between a direct current and an alternating current, a low voltage side switching circuit, connected between a second DC power source and the transformer, for executing an electric power conversion between a direct current and an alternating current, and a control circuit for controlling ON/OFF of switching elements included in each of the switching circuits and in which the high voltage side switching circuit includes a first vertical arm in which a first upper arm switching element and a first lower arm switching element are serially connected, a second vertical arm in which a second upper arm switching element and a second lower arm switching element are serially connected, a first smoothing capacitor connected in parallel to the first and second vertical arms and the first DC power source, and a series connector which is connected between a serial node of the first upper arm switching element and the first lower arm switching element and a serial node of the second upper arm switching element and the second lower arm switching element
  • control processes of the control circuit included in the bi-directional DC-DC converter according to any one of the above embodiments are selectively switched in accordance with a propagating direction of the energy, an input voltage, an input current, an output voltage, and an output current.
  • the small and high efficient bi-directional DC-DC converter constructed in such a manner that even when the step-up voltage ratio/output electric power is large at the time of the step-up operation, the voltage which is applied to the switching element of the low voltage side circuit is reduced can be provided.
  • the small and high efficient bi-directional DC-DC converter which can output the large electric power even at the time of the step-down operation can be provided.
  • FIG. 1 is a circuit constructional diagram of a bi-directional DC-DC converter according to an embodiment of the invention
  • FIG. 2 is a voltage/current waveform diagram for describing the step-down operation 1 of the bi-directional DC-DC converter according to the embodiment of the invention
  • FIG. 3 is a voltage/current waveform diagram for describing the step-down operation 2 of the bi-directional DC-DC converter according to the embodiment of the invention
  • FIG. 4 is a voltage/current waveform diagram for describing the step-up operation 1 of the bi-directional DC-DC converter according to the embodiment of the invention.
  • FIG. 5 is a voltage/current waveform diagram for describing the step-up operation 2 of the bi-directional DC-DC converter according to the embodiment of the invention.
  • FIG. 6 is a voltage/current waveform diagram for describing the step-up operation 3 of the bi-directional DC-DC converter according to the embodiment of the invention.
  • FIG. 1 is a circuit constructional diagram of a bi-directional DC-DC converter 5 according to the embodiment of the invention.
  • a smoothing capacitor 12 a load 14 , a first switching arm in which an emitter of an IGBT 101 and a collector of an IGBT 102 are connected, and a second switching arm in which an emitter of an IGBT 103 and a collector of an IGBT 104 are connected are connected in parallel to a power source 10 on a high voltage side.
  • Each of diodes 111 to 114 is connected between a collector and an emitter of each of the IGBTs 101 to 104 so as to allow a current to flow from the emitter side to the collector side.
  • body diodes may be used as diodes 111 to 114 .
  • Each of snubber capacitors 121 to 124 is connected between the collector and the emitter of each of the IGBTs 101 to 104 .
  • a primary winding 31 of a transformer 30 , a resonance reactor 20 , and a resonance capacitor 22 are serially connected between a node of the IGBTs 101 and 102 and a node of the IGBTs 103 and 104 .
  • the resonance reactor 20 may be replaced by a leakage inductance and a wiring inductance of the transformer 30 here.
  • a smoothing capacitor 42 and a load 44 are connected in parallel to a power source 40 on the low voltage side.
  • One end of a secondary winding 32 of the transformer 30 , one end of a secondary winding 33 of the transformer 30 , and one end of a smoothing reactor 46 are connected.
  • the other end of the smoothing reactor 46 is connected to a positive polarity of the power source 40 .
  • the other end of the secondary winding 32 is connected to a drain of a MOSFET 201 .
  • the other end of the secondary winding 33 is connected to a drain of a MOSFET 202 .
  • a source of the MOSFET 201 and a source of the MOSFET 202 are connected to a negative polarity of the power source 40 .
  • a source of the MOSFET 203 is connected to the drain of the MOSFET 201
  • a source of the MOSFET 204 is connected to the drain of the MOSFET 202
  • the other end of the clamping capacitor 48 is connected to the negative polarity of the power source 40 .
  • Each of diodes 211 to 214 is connected between the drain and the source of each of the MOSFETs 201 to 204 so as to allow a current to flow from the source side to the drain side.
  • Body diodes of MOSFETs can be also used as diodes 211 to 214 .
  • Each snubber capacitors may be connected between the drain and the source of each of the MOSFETs 201 to 204 .
  • the IGBTs 101 to 104 and MOSFETs 201 to 204 are switching-controlled by a control circuit 100 .
  • Voltage sensors 51 to 54 and current sensors 61 to 63 are connected to the control circuit 100 .
  • drain-source voltages V( 201 ) to V( 204 ) of the MOSFETs 201 to 204 the drain is set to be positive and, as for gate-source voltages Vg( 201 ) to Vg( 204 ), the gate is set to be positive.
  • Synthetic currents flowing in the MOSFETs 201 to 204 and the diodes 211 to 214 connected in parallel therewith are respectively assumed to be I( 201 ) to I( 204 ) when a direction of the current flowing from the drain to the source of each of the MOSFETs 201 to 204 is set to be positive.
  • a direction of the voltage applied from the second switching arm to the first switching arm is set to be positive.
  • a current I( 20 ) flowing in the resonance reactor 20 a direction of the current flowing from the first switching arm to the second switching arm is set to be positive.
  • a direction of the current flowing from a node of the MOSFETs 203 and 204 to the negative polarity of the power source 40 is set to be positive.
  • a voltage of the drain of each of the MOSFETs 203 and 204 in the case of setting the negative polarity of the power source 40 to a reference is assumed to be a voltage V( 48 ) of the clamping capacitor 48 .
  • a direction of a current I( 46 ) flowing in the smoothing reactor 46 is defined as follows.
  • a direction of the voltage applied from a node of the secondary winding 32 and the secondary winding 33 to the positive polarity of the power source 40 is set to be positive.
  • a direction of the voltage applied from the positive polarity of the power source 40 to the node of the secondary winding 32 and the secondary winding 33 is set to be positive.
  • a direction adapted to accelerate the current I( 46 ) in the smoothing reactor 46 in the positive direction is set to be positive.
  • the operation of the bi-directional DC-DC converter 5 according to the embodiment of the invention will be described in detail hereinbelow with reference to the drawings.
  • the operation for feeding the energy of the power source 10 to the power source 40 is assumed to be the step-down operation and the operation for feeding the energy of the power source 40 to the power source 10 is assumed to be the step-up operation.
  • FIG. 2 is a voltage/current waveform diagram for describing the step-down operation 1 .
  • the step-down operation 1 will be explained in detail hereinbelow with reference to FIG. 2 .
  • a 1 to h 1 denote periods of time.
  • the IGBTs 101 and 104 are in the ON state, the IGBTs 102 and 103 are in the OFF state, and the voltage of the power source 10 is applied to primary winding 31 of the transformer 30 through the IGBTs 101 and 104 , resonance capacitor 22 , and resonance reactor 20 .
  • the MOSFETs 202 and 203 are in the OFF state, the voltage developed in the secondary winding 32 is applied to the smoothing reactor 46 through the power source 40 and diode 211 , the current I( 46 ) increases gradually, and the energy is supplied to the power source 40 .
  • the voltages developed in the secondary windings 32 and 33 are applied to the clamping capacitor 48 through the diodes 214 and 211 , so that the clamping capacitor 48 is charged.
  • the IGBT 104 is turned off and, thereafter, the IGBT 103 is turned on.
  • the current I( 20 ) which has been flowing in the IGBT 104 discharges the snubber capacitor 123 while charging the snubber capacitor 124 .
  • the voltage V( 103 ) reaches the zero voltage, the diode 113 is made conductive.
  • the IGBT 103 is turned on (zero voltage switching).
  • the current I( 20 ) is refluxed by a path passing through the resonance reactor 20 , primary winding 31 , diode 113 , IGBT 101 , and resonance capacitor 22 .
  • the current flowing in the primary winding 31 as mentioned above is hereinbelow referred to as a circulating current.
  • the MOSFET 204 Since the MOSFET 204 is in the ON state, the current I( 201 ) is negative, and the voltage V( 201 ) is equal to the zero voltage, the voltage V( 48 ) of the clamping capacitor 48 is applied to the secondary windings 32 and 33 .
  • the voltage developed in the primary winding 31 is applied to the resonance reactor 20 and gradually decreases the current I( 20 ). Therefore, the circulating current decreases and the energy which is lost on the path through which the circulating current flows can be reduced. In association with the decrease in circulating current, a discharge current of the clamping capacitor 48 increases.
  • the MOSFET 204 is turned off and, thereafter, the MOSFET 202 is turned on.
  • the MOSFET 204 is turned off, the discharge of the clamping capacitor 48 is finished and the decrease in circulating current also becomes gentle.
  • the circulating current decreases gently.
  • the current which has been flowing in the MOSFET 204 is commutated to the diode 212 .
  • the synchronous rectification is performed.
  • the energy accumulated in the smoothing reactor 46 is supplied to the power source 40 and the current I( 46 ) decreases gradually.
  • the current I( 46 ) had been flowing on the path passing through the diode 211 (MOSFET 201 ) and secondary winding 32 for the period of time a 1
  • the current is also shunted to a path passing through the diode 212 (MOSFET 202 ) and secondary winding 33 for the period of time c 1 .
  • the more the circulating current is reduced the more the current is equivalently shunted to those two paths. Consequently, the conduction loss can be reduced.
  • a circulating current value necessary to enable the IGBT 102 to perform the zero voltage switching can be arithmetically operated from a voltage of the power source 10 (voltage sensor 51 ), an electrostatic capacitance of the resonance capacitor 22 , an inductance of the resonance reactor 20 , electrostatic capacitances of the snubber capacitors 121 to 124 , and a dead time of the IGBTs 101 and 102 (or a dead time of the IGBTs 103 and 104 ).
  • off timing for the MOSFET 204 has to be accurately decided.
  • Such timing can be arithmetically operated on the basis of an off time lag of the IGBT 104 and the MOSFET 204 from the voltage of the power source 40 (voltage sensor 52 ), the current I( 46 ) in the smoothing reactor 46 (current sensor 62 ), and a turn ratio of the transformer 30 in addition to the information which has been used in order to obtain the circulating current value necessary to enable the IGBT 102 to perform the zero voltage switching in the above process, or the off timing for the MOSFET 204 may be determined on the basis of a measurement value from the current sensor 61 showing the input current of a full bridge and a measurement value from the current sensor 63 showing the circulating current.
  • the IGBT 101 is turned off and, thereafter, the IGBT 102 is turned on and the MOSFET 201 is turned off.
  • the IGBT 101 is turned off, the circulating current which has been flowing in the IGBT 101 discharges the snubber capacitor 122 while charging the snubber capacitor 121 .
  • the voltage V( 102 ) reaches the zero voltage, the diode 112 is made conductive.
  • the IGBT 102 is turned on (zero voltage switching).
  • the MOSFET 201 is turned off before the period of time d 1 is finished.
  • the circulating current flows in the diode 112 , resonance capacitor 22 , resonance reactor 20 , primary winding 31 , and diode 113 and reaches the power source 10 .
  • the voltage of the power source 10 is applied to the resonance reactor 20 and the circulating current decreases.
  • the IGBTs 102 and 103 are in the ON state, after the circulating current reached zero, the circulating current increases in the reverse direction. In association with it, the current flowing through the diode 211 (MOSFET 201 ) and the secondary winding 32 decreases and the current flowing through the diode 212 (MOSFET 202 ) and the secondary winding 33 increases. The MOSFET 201 is turned off before the current flowing through the secondary winding 32 reaches zero (the current flowing through the secondary winding 33 reaches the current I( 46 )).
  • the MOSFET 203 is turned on.
  • the diode 211 is reversely made conductive and, thereafter, reversely recovered.
  • the current which had been flowing during the reverse conduction is commutated to the diode 213 after the diode 211 was reversely recovered.
  • the MOSFET 203 is turned on (zero voltage switching).
  • the MOSFET 203 can be also turned on by detecting such a voltage increase by the voltage sensor 53 .
  • the voltage of the power source 10 has been applied to the primary winding 31 of the transformer 30 through the IGBTs 102 and 103 , resonance capacitor 22 , and resonance reactor 20 .
  • the MOSFETs 201 and 204 are in the OFF state, the voltage developed in the secondary winding 33 is applied to the smoothing reactor 46 through the power source 40 and diode 212 , the current I( 46 ) increases gradually, and the energy is supplied to the power source 40 .
  • the voltages developed in the secondary windings 32 and 33 are applied to the clamping capacitor 48 through the diodes 213 and 212 , thereby charging the clamping capacitor 48 .
  • the operation for the period of time e 1 is symmetrical with the operation for the period of time a 1 . Subsequently, after the periods of time f 1 to h 1 , the operation cycle is returned to the period of time a 1 . Since the operations for the periods of time f 1 to h 1 are symmetrical with those for the periods of time b 1 to d 1 , their detailed description is omitted here.
  • the resonance capacitor 22 Since the voltage is developed in the resonance capacitor 22 in the direction adapted to raise the voltage which is applied to the primary winding 31 for the periods of time a 1 and e 1 mentioned above, the resonance capacitor 22 provides an effect of increasing the output electric power.
  • the voltage clamping circuit suppresses the generation of the surge voltage as mentioned above. Therefore, the voltage clamping circuit has such an effect that elements of a low withstanding voltage can be used as diodes 211 and 212 and MOSFETs 201 and 202 .
  • the zero voltage switching at the time of turning on the IGBTs 101 and 102 can be realized as mentioned above. Therefore, since the circulating current decreases at the time of the small load, it is difficult to perform the zero voltage switching. To solve such a problem, for example, if the MOSFET 204 is turned off and the MOSFET 204 is turned on for the periods of time b 1 and c 1 prior to turning off the IGBT 104 , since the circulating current increases, the zero voltage switching at the time of turning on the IGBT 101 can be realized even in the case of the small load. Order of the timing for turning on the MOSFET 202 and the timing for turning off the IGBT 104 is not limited.
  • the above operating method has an effect of improving the efficiency even at the time of the small load.
  • a time-dependent ratio of the period of time during which the IGBTs 101 and 104 are simultaneously in the ON state and a time-dependent ratio of the period of time during which the IGBTs 102 and 103 are simultaneously in the ON state are changed, thereby adjusting the output electric power.
  • Such a time-dependent ratio is called a duty.
  • the step-down operation 2 which will be described hereinbelow is applied.
  • FIG. 3 is a voltage/current waveform diagram for describing the step-down operation 2 .
  • the step-down operation 2 will be described in detail hereinbelow with reference to FIG. 3 .
  • a 2 to f 2 correspond to the periods of time a 2 to f 2 .
  • the operation for the period of time a 2 is similar to that for the period of time a 1 of the step-down operation 1 mentioned above and its detailed explanation is omitted here.
  • the ON/OFF states of the IGBTs 101 and 104 coincide in the step-down operation 2 .
  • the current I( 20 ) discharges the snubber capacitor 122 while charging the snubber capacitor 121 and discharges the snubber capacitor 123 while charging the snubber capacitor 124 .
  • the diode 112 is made conductive.
  • the diode 113 is made conductive.
  • the IGBTs 102 and 103 are turned on (zero voltage switching).
  • the current I( 20 ) flows through the diode 112 , resonance capacitor 22 , resonance reactor 20 , primary winding 31 , and diode 113 and reaches the power source 10 . Since the MOSFET 204 is in the ON state, the current I( 20 ) is negative, and voltage V( 201 ) is equal to the zero voltage, the voltage V( 48 ) in the clamping capacitor 48 is applied to the secondary windings 32 and 33 . The voltage of the power source 10 is applied to the resonance reactor 20 and the current I( 20 ) decreases.
  • the voltage developed in the primary winding 31 is also additionally applied to the resonance reactor 20 , a decreasing speed of the current I( 20 ) is higher than that for each of the periods of time b 1 and d 1 in the step-down operation 1 .
  • the voltage developed in the primary winding 31 is almost equal to the voltage of the power source 10 here.
  • the voltage has also been developed in the resonance capacitor 22 in the same direction as that of the voltage developed in the primary winding 31 .
  • the voltage which is twice or more times as high as the voltage of the power source 10 is applied to the resonance reactor 20 .
  • the energy accumulated in the smoothing reactor 46 is supplied to the power source 40 and the current I( 46 ) decreases gradually.
  • the MOSFET 201 is turned off before the current flowing in the secondary winding 32 reaches zero (the current flowing in the secondary winding 33 reaches the current I( 46 )).
  • the operation for the period of time d 2 is similar to that for the period of time e 1 of the step-down operation 1 and its detailed explanation is omitted here.
  • the operation for the period of time d 2 is symmetrical with that for the period of time a 2 .
  • the operation cycle is returned to the period of time a 2 . Since the operations for the periods of time e 2 and f 2 are symmetrical with those for the periods of time b 2 and c 2 , their detailed description is omitted here.
  • the resonance capacitor 22 has an effect of increasing the output electric power.
  • the voltage clamping circuit has such an effect that elements of a low withstanding voltage can be used as diodes 211 and 212 and MOSFETs 201 and 202 .
  • the positive voltage is applied to the smoothing reactor 46 and the voltage which is twice or more times as high as the voltage of the power source 10 is applied to the resonance reactor 20 and a change ratio of the current I( 20 ) is increased, thereby extending the periods of time a 2 and d 2 .
  • the voltage which is twice or more times as high as the voltage of the power source 10 is applied to the resonance reactor 20 .
  • the voltage which is applied to the resonance reactor 20 is equal to a voltage which is about twice as high as the voltage of the power source 10 .
  • a point that the electric power larger than the maximum output electric power in the step-down operation 1 can be outputted is a maximum advantage.
  • the ON/OFF states of the IGBTs 101 and 104 coincide, the ON/OFF states of the IGBTs 102 and 103 coincide as mentioned above, and the duty is maximum. Therefore, the output electric power is adjusted by changing the durations of the periods of time b 2 and e 2 , that is, by changing the off timing of the MOSFETs 203 and 204 .
  • the off timing can be delayed to a point before or after the timing when the diodes 211 and 212 are reversely recovered.
  • the off timing of the MOSFETs 203 and 204 can be also decided by detection signals from the voltage sensors 53 and 54 for detecting the reverse recovery of the diodes 211 and 212 . In this case, the periods of time c 2 and f 2 do not exist.
  • FIG. 4 is a voltage/current waveform diagram for describing the step-up operation 1 .
  • the step-up operation 1 will be explained in detail hereinbelow with reference to FIG. 4 .
  • a 1 to F 1 denote periods of time.
  • the MOSFETs 201 and 202 are in the ON state and the MOSFETs 202 and 203 are in the OFF state.
  • the voltage of the power source 40 is applied to the smoothing reactor 46 through the secondary windings 32 and 33 and MOSFETs 201 and 202 and the energy of the power source 40 is accumulated to the resonance reactor 46 .
  • the IGBT 103 is in the ON state, the IGBTs 101 , 102 , and 104 are in the OFF state, and the circulating current flows on a path passing through the IGBT 103 , primary winding 31 , resonance reactor 20 , resonance capacitor 22 , and diode 111 . Since the charges have been accumulated in the resonance capacitor 22 and the voltage has been developed in the direction adapted to increase the circulating current, the circulating current increases gradually.
  • the MOSFET 202 and IGBT 103 are turned off, the current which has been flowing in the MOSFET 202 flows in the diode 214 , thereby charging the clamping capacitor 48 . At this time, the MOSFET 204 is turned on (zero voltage switching). The circulating current which has been flowing in the IGBT 103 discharges the snubber capacitor 124 while charging the snubber capacitor 123 . When the voltage V( 104 ) reaches the zero voltage, the diode 114 is made conductive. At this time, the IGBT 104 is turned on (zero voltage switching).
  • the voltage V( 48 ) of the clamping capacitor 48 is applied to the secondary windings 32 and 33 .
  • the voltage obtained by subtracting the voltage of the power source 10 from the voltage developed in the primary winding 31 is applied to the resonance reactor 20 and the magnitude of the current I( 20 ) increases.
  • the current I( 20 ) flows through the diode 114 , primary winding 31 , resonance reactor 20 , resonance capacitor 22 , and diode 111 and reaches the power source 10 .
  • the energy is supplied to the power source 10 .
  • the energy accumulated in the smoothing reactor 46 is emitted and the current I( 46 ) decreases.
  • the MOSFET 204 When the MOSFET 204 is turned off, the discharge current of the clamping capacitor 48 which has been flowing in the MOSFET 204 makes the diode 212 conductive. At this time, the MOSFET 202 is turned on (zero voltage switching).
  • the voltage of the power source 40 is applied to the smoothing reactor 46 and the energy of the power source 40 is accumulated in the smoothing reactor 46 .
  • the diode 111 is reversely made conductive. After that, when the diode is reversely recovered, a snubber capacitor C 102 is discharged while the snubber capacitor 121 is charged. When the voltage V( 102 ) reaches the zero voltage, the diode 112 is made conductive.
  • the energy of the power source 10 accumulated in the resonance reactor 20 becomes the circulating current flowing on the path passing through the diode 112 , resonance capacitor 22 , resonance reactor 20 , primary winding 31 , and IGBT 104 . Since the charges have been accumulated in the resonance capacitor 22 and the voltage has been developed in the direction adapted to increase the circulating current, the circulating current increases gradually.
  • the voltage of the power source 40 is applied to the smoothing reactor 46 and the energy of the power source 40 is accumulated in the smoothing reactor 46 .
  • the operation for the period of time D 1 is symmetrical with that for the period of time A 1 . Subsequently, after the periods of time E 1 and F 1 , the operation cycle is returned to the period of time A 1 . Since the operations for the periods of time E 1 and F 1 are symmetrical with those for the periods of time B 1 and C 1 , their detailed description is omitted here.
  • the synchronous rectification is performed.
  • the direction of the circulating current flowing when the MOSFET 202 is turned off for the period of time B 1 is equal to the direction of the energy which is sent to the power source 10 for the periods of time B 1 and C 1 . Therefore, the larger the circulating current is, the larger output electric power is liable to be obtained.
  • the current I( 202 ) at the time of turning off the MOSFET 202 decreases and the energy which is lost when the MOSFET 202 is turned off can be reduced. However, by shutting off the current I( 202 ), the energy is accumulated in the clamping capacitor.
  • the MOSFET 202 is turned on for the period of time C 1 , the zero voltage switching can be performed by using such an energy. Therefore, the larger the circulating current is, the more it is difficult to perform the zero voltage switching at the time of turning on the MOSFET 202 . This is true of the period of time E 1 .
  • the zero voltage switching at the time of turning on the MOSFET 202 becomes difficult.
  • the IGBT 103 is turned off for the period of time B 1 prior to turning off the MOSFET 202 , since the circulating current is reduced or reversely flows, the current to be shut off at the time of turning off the MOSFET 202 increases.
  • the zero voltage switching can be realized when the MOSFET 202 is turned on. This is true of the period of time E 1 .
  • the above operating method has an effect of raising the efficiency even at the time of the small load.
  • the output electric power is adjusted by changing the time-dependent ratio (on duty) of the ON periods of time of the MOSFETs 201 and 202 . If the on duty is increased, the voltage V( 48 ) of the clamping capacitor 48 rises and the voltage developed in the primary winding 31 also rises, so that the output electric power increases. However, the voltages which are applied to the clamping capacitor 48 , MOSFETs 201 to 204 , and diodes 211 to 214 are determined by the voltage of the power source 40 , the on duty, the wiring inductance, and the breaking currents of the MOSFETs 201 to 204 .
  • FIG. 5 is a voltage/current waveform diagram for describing the step-up operation 2 .
  • the step-up operation 2 will be explained in detail hereinbelow with reference to FIG. 5 .
  • a 2 to H 2 denote periods of time.
  • the operation for the period of time A 2 is similar to that for the period of time A 1 of the step-up operation 1 and its detailed explanation is omitted here.
  • the MOSFET 202 When the MOSFET 202 is turned off, the current which has been flowing in the MOSFET 202 flows in the diode 214 and charges the clamping capacitor 48 . At this time, the MOSFET 204 is turned on (zero voltage switching).
  • the IGBT 103 is in the ON state. Therefore, since the voltage obtained without subtracting the voltage of the power source 10 from the voltage developed in the primary winding 31 is applied to the resonance reactor 20 , the magnitude of the current I( 20 ) increases at a speed higher than that for the period of time B 1 of the step-up operation 1 . At this time, the current I( 20 ) flows on the same path of the circulating current as that for the period of time A 2 .
  • the energy accumulated in the smoothing reactor 46 is emitted and the current I( 46 ) decreases.
  • the IGBT 103 When the IGBT 103 is turned off, the circulating current which has been flowing in the IGBT 103 discharges the snubber capacitor 124 while charging a snubber capacitor C 103 .
  • the voltage V( 104 ) reaches the zero voltage, the diode 114 is made conductive. At this time, the IGBT 104 is turned on (zero voltage switching).
  • the voltage has been developed in the primary winding 31 in a manner similar to that for the period of time B 2 .
  • the current I( 20 ) flows through the diode 114 , primary winding 31 , resonance reactor 20 , resonance capacitor 22 , and diode 111 and reaches the power source 10 .
  • the energy is supplied to the power source 10 .
  • the energy accumulated in the smoothing reactor 46 is emitted and the current I( 46 ) decreases.
  • the operation for the period of time D 2 is similar to that for the period of time C 1 of the step-up operation 1 and its detailed explanation is omitted here.
  • the operation for the period of time E 2 is similar to that for the period of time D 1 of the step-up operation 1 and its detailed explanation is omitted here.
  • the operation for the period of time E 2 is symmetrical with that for the period of time A 2 .
  • the operation cycle is returned to the period of time A 2 . Since the operations for the periods of time F 2 to H 2 are symmetrical with those for the periods of time B 2 to D 2 , their detailed description is omitted here.
  • the output electric power is adjusted by changing the off timing of the IGBTs 103 and 104 and by changing the duration of the periods of time B 2 and F 2 . If the off timing of the IGBTs 103 and 104 is delayed, a change ratio of the current I( 20 ) for each of the periods of time C 2 and G 2 decreases, a voltage drop due to a resonance reactor L is suppressed, and the output electric power increases. However, if the output electric power is increased by delaying the off timing of the IGBTs 103 and 104 , a peak value of the current I( 20 ) increases. The breaking currents of the MOSFETs 201 and 202 increase.
  • step-up operation 3 which will be described hereinbelow is applied.
  • FIG. 6 is a voltage/current waveform diagram for describing the step-up operation 3 .
  • the step-up operation 3 will be explained in detail hereinbelow with reference to FIG. 6 .
  • a 3 to H 3 denote periods of time.
  • the operation for the period of time B 3 is similar to that for the period of time B 2 of the step-up operation 2 and its detailed explanation is omitted here.
  • the operation for the period of time C 3 is similar to that for the period of time C 2 of the step-up operation 2 and its detailed explanation is omitted here.
  • the operation which is executed for a period of time until the magnitude of the current I( 20 ) decreases and reaches zero is similar to that for the period of time D 2 of the step-up operation 2 and its detailed explanation is omitted here.
  • the IGBTs 101 and 104 are in the ON state, the direction of the current I( 20 ) is reversed and the current I( 20 ) increases.
  • the energy of the power source 10 is accumulated in the resonance reactor 20 .
  • the current I( 20 ) discharges the snubber capacitor 122 while charging the snubber capacitor 121 .
  • the diode 112 is made conductive.
  • the zero voltage switching is performed.
  • Other operations are similar to those for the period of time E 2 of the step-up operation 2 and their detailed explanation is omitted here.
  • the operation for the period of time E 3 is symmetrical with that for the period of time A 3 . Subsequently, after the periods of time F 3 to H 3 , the operation cycle is returned to the period of time A 3 . Since the operations for the periods of time F 3 to H 3 are symmetrical with those for the periods of time B 3 to D 3 , their detailed description is omitted here.
  • the durations of the periods of time D 3 and H 3 are changed. That is, the output electric power is adjusted by changing the off timing of the IGBTs 101 and 102 . If the off timing of the IGBTs 101 and 102 is delayed, the circulating currents for the periods of time A 3 and E 3 increase and the output electric power increases.
  • the voltage clamping circuit accumulates the energy of the current which is shut off when the MOSFETs 201 and 202 are turned off, thereby suppressing that the surge voltage is generated in the drain voltage of each of the MOSFETs 201 and 202 . Therefore, the voltage clamping circuit has such an effect that elements of a low withstanding voltage can be used as diodes 211 and 212 and MOSFETs 201 and 202 .
  • the voltage has been developed in the resonance capacitor 22 in the direction adapted to increase the circulating current for the period of time during which the circulating current flows. Therefore, the resonance capacitor 22 has an effect of increasing the output electric power.
  • step-down operations 1 and 2 and the foregoing step-up operations 1 , 2 , and 3 can be also switched and executed, respectively.
  • the switching of the step-down operations and step-up operations of the bi-directional DC-DC converter 5 will be described hereinbelow.
  • the step-down operation 1 is applied at the small load.
  • the duty increases in association with an increase in load. A time difference between the off timing of the IGBT 103 and the off timing of the MOSFET 203 and a time difference between the off timing of the IGBT 104 and the off timing of the MOSFET 204 decrease and those time differences are soon eliminated.
  • the step-up operation 1 is applied at the small load.
  • the on duties of the MOSFETs 201 and 202 increase in association with the increase in load.
  • a time difference between the off timing of the MOSFET 202 and the off timing of the IGBT 103 and a time difference between the off timing of the MOSFET 201 and the off timing of the IGBT 104 decrease and those time differences are soon eliminated.
  • the on duty reaches an upper limit.
  • the reason why there is an upper limit in the on duty has already been described in the explanation of the step-up operation 1 .
  • the step-up operation 2 is applied. Specifically speaking, a time which is necessary until the IGBT 103 is turned off after the MOSFET 202 was turned off and a time which is necessary until the IGBT 104 is turned off after the MOSFET 201 was turned off are extended.
  • the duration of the time reaches an upper limit. The reason why there is an upper limit in the duration of the time has already been described in the explanation of the step-up operation 2 .
  • the step-up operation 3 is applied.
  • the bi-directional DC-DC converter 5 has such a feature that, in the cases of both of the step-down operation and the step-up operation, by switching a plurality of operations in accordance with a load state, the small insulating type bi-directional DC-DC converter in which the high efficiency and high output are obtained even in the case of the small load can be realized.
  • the invention is not limited to the foregoing embodiments. Naturally, the invention can be also applied to various circuit constructions in which, for example, a current doubler synchronous rectifying circuit may be used in place of the switching circuit on the low voltage side, the switching elements of the switching circuit on the low voltage side are constructed in a full bridge form, and the like.
US12/193,244 2007-08-28 2008-08-18 Bi-directional DC-DC converter and method for controlling the same Expired - Fee Related US7848118B2 (en)

Priority Applications (2)

Application Number Priority Date Filing Date Title
US12/896,176 US7911810B2 (en) 2007-08-28 2010-10-01 Bi-directional DC-DC converter and method for controlling the same
US13/024,434 US8130515B2 (en) 2007-08-28 2011-02-10 Bi-directional DC-DC converter and method for controlling the same

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
JP2007-221892 2007-08-28
JP2007221892A JP4378400B2 (ja) 2007-08-28 2007-08-28 双方向ｄｃ−ｄｃコンバータ及び双方向ｄｃ−ｄｃコンバータの制御方法

Related Child Applications (1)

Application Number Title Priority Date Filing Date
US12/896,176 Continuation US7911810B2 (en) 2007-08-28 2010-10-01 Bi-directional DC-DC converter and method for controlling the same

Publications (2)

Publication Number Publication Date
US20090059622A1 US20090059622A1 (en) 2009-03-05
US7848118B2 true US7848118B2 (en) 2010-12-07

Family

ID=40407188

Family Applications (3)

Application Number Title Priority Date Filing Date
US12/193,244 Expired - Fee Related US7848118B2 (en) 2007-08-28 2008-08-18 Bi-directional DC-DC converter and method for controlling the same
US12/896,176 Expired - Fee Related US7911810B2 (en) 2007-08-28 2010-10-01 Bi-directional DC-DC converter and method for controlling the same
US13/024,434 Expired - Fee Related US8130515B2 (en) 2007-08-28 2011-02-10 Bi-directional DC-DC converter and method for controlling the same

Family Applications After (2)

Application Number Title Priority Date Filing Date
US12/896,176 Expired - Fee Related US7911810B2 (en) 2007-08-28 2010-10-01 Bi-directional DC-DC converter and method for controlling the same
US13/024,434 Expired - Fee Related US8130515B2 (en) 2007-08-28 2011-02-10 Bi-directional DC-DC converter and method for controlling the same

Country Status (4)

Country Link
US (3) US7848118B2 (zh)
JP (1) JP4378400B2 (zh)
CN (2) CN101378232B (zh)
TW (1) TWI368384B (zh)

Cited By (18)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20100187909A1 (en) * 2008-07-29 2010-07-29 Odu Steckverbindungssysteme Gmbh & Co. Kg Method and System for Inductively Transmitting Energy and Information
US20100302824A1 (en) * 2009-05-29 2010-12-02 Sony Corporation Power source apparatus
US20110128759A1 (en) * 2007-08-28 2011-06-02 Takae Shimada Bi-directional dc-dc converter and method for controlling the same
US20110149611A1 (en) * 2009-12-21 2011-06-23 Intersil Americas Inc. Bidirectional signal conversion
US20110176335A1 (en) * 2010-01-21 2011-07-21 Li Zeng Resonant converters and burst mode control method thereof
US8593831B2 (en) * 2010-10-28 2013-11-26 Eltek Valere As Method for controlling a series resonant DC/DC converter
WO2014093038A1 (en) * 2012-11-28 2014-06-19 Guangdong Rui Ding Electrical Technology LTD. Super-efficient single-stage isolated switching power amplifier
US20150131337A1 (en) * 2013-09-09 2015-05-14 Aerovironment, Inc. Resonant Isolated Active PFC Rectifier
US20160072390A1 (en) * 2013-04-26 2016-03-10 Panasonic Intellectual Property Management Co., Ltd. Bidirectional dc/dc converter
US20160315557A1 (en) * 2015-04-23 2016-10-27 Panasonic Intellectual Property Management Co., Ltd. Power converter
US9762127B2 (en) * 2013-03-18 2017-09-12 Panasonic Intellectual Property Management Co., Ltd. Power converter and power conditioner
US9774269B2 (en) 2013-07-11 2017-09-26 Fuji Electric Co., Ltd. Bidirectional DC/DC converter
US9906150B1 (en) * 2016-11-09 2018-02-27 Astec International Limited Open loop bi-directional converters and corresponding control methods
CN109672344A (zh) * 2019-01-21 2019-04-23 中国科学院电工研究所 双向直流变换器及其控制方法
US20190280526A1 (en) * 2016-10-19 2019-09-12 Amosense Co., Ltd Magnetic induction power supply device
US11095219B2 (en) * 2017-08-22 2021-08-17 Diamond Electric Mfg. Co., Ltd. Converter with soft switching function
US11511637B2 (en) 2019-05-24 2022-11-29 Huawei Digital Power Technologies Co., Ltd. Integrated charger and motor control system
US20230246556A1 (en) * 2022-01-28 2023-08-03 Analog Devices, Inc. Active-clamp current-fed push-pull converter for bidirectional power transfer

Families Citing this family (108)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP4967588B2 (ja) * 2006-10-17 2012-07-04 トヨタ自動車株式会社 コンバータ制御装置
JP5282197B2 (ja) * 2007-09-01 2013-09-04 正敏 井森 共振の周波数依存性を利用して直流電圧を安定化する電源における搬送波の振幅の制御
JP5386751B2 (ja) * 2007-11-06 2014-01-15 国立大学法人 長崎大学 電力変換回路の制御装置
KR101832246B1 (ko) * 2008-03-25 2018-02-26 델타 일렉트로닉스, 인크. 부하 조건의 영역에 있어서 효율적으로 작동하는 전력 변환기 시스템
KR100940227B1 (ko) * 2008-07-04 2010-02-04 삼성전기주식회사 전류 스트레스를 개선한 위상 천이 풀 브릿지 컨버터
WO2010143304A1 (ja) * 2009-06-12 2010-12-16 日立コンピュータ機器株式会社 電源装置
JP5229139B2 (ja) * 2009-07-07 2013-07-03 株式会社豊田自動織機 双方向ｄｃｄｃコンバータ
KR101387279B1 (ko) 2009-07-31 2014-04-18 다이킨 고교 가부시키가이샤 전력변환장치
KR101031217B1 (ko) * 2009-10-21 2011-04-27 주식회사 오리엔트전자 고정 시비율로 동작하는 ｌｌｃ 공진 컨버터를 사용한 2단 방식 절연형 양방향 ｄｃ／ｄｃ 전력변환기
CN101741240B (zh) * 2009-12-30 2011-11-30 广东易事特电源股份有限公司 一种双向dc/dc变换器的拓扑结构及变换器
TWI415381B (zh) * 2010-02-01 2013-11-11 Chip Goal Electronics Corp 雙向驅動積體電路與雙向驅動的方法
US9276458B2 (en) * 2010-04-01 2016-03-01 Telefonaktiebolaget L M Ericsson (Publ) Switch control for controlling switch arrangements of a power converter based on efficiency
CN102917676A (zh) 2010-04-30 2013-02-06 塞罗斯医学有限责任公司 使用组合的方式治疗眼组织的方法和装置
WO2011161729A1 (ja) 2010-06-25 2011-12-29 株式会社 日立製作所 Ｄｃ－ｄｃコンバータ
JP5631675B2 (ja) * 2010-09-17 2014-11-26 新電元工業株式会社 Ｄｃ／ｄｃコンバータ
US8867986B1 (en) * 2010-10-26 2014-10-21 Pathfinder Digital, LLC Enhanced mobile satellite communication system
GB2484970B (en) * 2010-10-28 2012-11-07 Eltek Valere As Series resonant DC/DC converter
TWI406487B (zh) * 2010-11-17 2013-08-21 Univ Nat Cheng Kung 雙向式直流－直流電源轉換器
DE102010060957A1 (de) * 2010-12-02 2012-06-06 Sma Solar Technology Ag Verfahren zum Betreiben eines Gleichspannungswandlers
US8553691B2 (en) 2011-02-22 2013-10-08 Alcatel Lucent Efficient multicasting in a distributed system architecture
EP2495858A1 (de) * 2011-03-01 2012-09-05 Rheinisch-Westfälisch-Technische Hochschule Aachen Bidirektionaler Gleichspannungswandler
DE112012001746T5 (de) * 2011-04-18 2014-01-23 Mitsubishi Electric Corp. Energie-Umwandlungsvorrichtung und mit einer solchen ausgestattete Stromversorgungsvorrichtung in einem Fahrzeug
US8879279B2 (en) * 2011-05-13 2014-11-04 Texas Instruments Incorporated Systems and methods for constant current control in an LLC resonant power regulator
JP5530401B2 (ja) * 2011-05-26 2014-06-25 株式会社日立情報通信エンジニアリング 電源装置
JP5403438B2 (ja) * 2011-11-29 2014-01-29 Ｔｄｋ株式会社 Ｄｃｄｃコンバータ及びｄｃｄｃコンバータの制御方法
JP5382552B2 (ja) * 2011-11-18 2014-01-08 Ｔｄｋ株式会社 Ｄｃｄｃコンバータ及びｄｃｄｃコンバータの制御方法
JP5387628B2 (ja) 2011-07-29 2014-01-15 Ｔｄｋ株式会社 電流型絶縁コンバータ
JP5768886B2 (ja) * 2011-08-04 2015-08-26 株式会社村田製作所 スイッチング電源装置
DE102011086219A1 (de) * 2011-11-11 2013-05-16 Panasonic Corporation Getakteter leistungswandler mit aktiver begrenzung
US9602067B2 (en) * 2011-11-15 2017-03-21 Wen-Hsiung Hsieh Switching amplifier with pulsed current supply
CN102403905A (zh) * 2011-11-18 2012-04-04 江苏艾索新能源股份有限公司 双向dc/dc变换器
CN102570831B (zh) * 2012-01-05 2015-01-21 深圳市高斯宝电气技术有限公司 一种隔离直流双向变换器
WO2013121665A1 (ja) 2012-02-14 2013-08-22 三菱電機株式会社 Ｄｃ／ｄｃコンバータ
CN102611319B (zh) * 2012-03-14 2014-10-29 东南大学 非对称双向直流变换器
US9413246B2 (en) 2012-04-12 2016-08-09 On-Bright Electronics (Shanghai) Co., Ltd. Systems and methods for regulating power conversion systems with output detection and synchronized rectifying mechanisms
US9595874B2 (en) 2012-04-12 2017-03-14 On-Bright Electronics (Shanghai) Co., Ltd. Systems and methods for regulating power conversion systems with output detection and synchronized rectifying mechanisms
US10622902B2 (en) 2012-04-12 2020-04-14 On-Bright Electronics (Shanghai) Co., Ltd. Systems and methods for regulating power conversion systems with output detection and synchronized rectifying mechanisms
CN103378751B (zh) 2012-04-12 2015-04-01 昂宝电子（上海）有限公司 用于开关反激式电源变换系统的系统和方法
JP5556852B2 (ja) * 2012-06-01 2014-07-23 Ｔｄｋ株式会社 双方向ｄｃｄｃコンバータ
JP5987496B2 (ja) * 2012-06-27 2016-09-07 株式会社日立情報通信エンジニアリング Ｄｃ−ｄｃコンバータ
TWI540822B (zh) 2012-07-03 2016-07-01 盈正豫順電子股份有限公司 雙向直流／直流轉換器之控制方法
WO2014032156A1 (en) * 2012-08-27 2014-03-06 Bombardier Transportation Gmbh Adaptive soft switching control for power converter
JP5552149B2 (ja) * 2012-10-05 2014-07-16 オリジン電気株式会社 コンバータ及び双方向コンバータ
CN103715907B (zh) * 2012-10-09 2016-03-09 旭隼科技股份有限公司 双向直流/直流转换器
JP6103874B2 (ja) * 2012-10-12 2017-03-29 株式会社日立情報通信エンジニアリング 電源装置とその運転方法
TWI474599B (zh) * 2012-10-17 2015-02-21 Voltronic Power Technology Corp 雙向直流／直流轉換器
FR2997583B1 (fr) * 2012-10-31 2014-11-21 Valeo Equip Electr Moteur Systeme d'alimentation electrique a double stockeurs d'energie electrique d'un vehicule automobile ou hybride
US8929101B2 (en) * 2012-12-17 2015-01-06 Chung-Shan Institute Of Science And Technology DC to DC converting circuit
KR101397728B1 (ko) * 2012-12-21 2014-05-20 한국과학기술원 전원 공급 장치
US9413271B2 (en) * 2013-03-14 2016-08-09 Combined Energies, Llc Power conversion system with a DC to DC boost converter
EP2967710A4 (en) 2013-03-15 2016-11-02 Aleyegn Inc METHODS AND APPARATUS FOR ELASTO MODULATION OF SCLERAL TRANSLOCATION
WO2015004825A1 (ja) * 2013-07-11 2015-01-15 三菱電機株式会社 Ｄｃ／ｄｃコンバータ
CN103595287B (zh) * 2013-11-27 2016-09-07 东南大学 一种双向功率流高频隔离有源钳位逆变器的控制方法
CN103683941A (zh) * 2013-12-12 2014-03-26 南京航空航天大学 一种隔离型谐振高升压比电路
JP2015139258A (ja) 2014-01-21 2015-07-30 サンケン電気株式会社 スイッチング電源装置
JP6286793B2 (ja) * 2014-01-24 2018-03-07 株式会社日立情報通信エンジニアリング Ｄｃ−ｄｃコンバータ、二次電池充放電システム、およびｄｃ−ｄｃコンバータの制御方法
US9356521B2 (en) * 2014-01-30 2016-05-31 Sanken Electric Co., Ltd. Switching power-supply device having wide input voltage range
CN103840697B (zh) * 2014-04-01 2017-03-22 青岛理工大学 一种有源箝位高增益单级可升压逆变器
DE102014214542A1 (de) * 2014-07-24 2016-02-11 Rheinisch-Westfälisch-Technische Hochschule Aachen Gleichspannungswandler mit Transformator
JP5962717B2 (ja) * 2014-07-29 2016-08-03 株式会社日立製作所 Ｄｃ−ｄｃコンバータ
KR102344534B1 (ko) * 2014-11-11 2021-12-29 엘지이노텍 주식회사 전력변환부
EP3021475B1 (de) * 2014-11-13 2020-10-07 AEG Power Solutions GmbH Bidirektionale Stromrichterschaltungsanordnung mit Potentialtrennung und Verfahren zum Betreiben der bidirektionalen Stromrichterschaltungsanordnung
CN104333240A (zh) * 2014-11-21 2015-02-04 小米科技有限责任公司 一种谐振整流装置、谐振整流控制方法及装置
JP6217655B2 (ja) * 2015-01-15 2017-10-25 トヨタ自動車株式会社 電力変換装置
JP6349265B2 (ja) 2015-01-28 2018-06-27 オムロン株式会社 双方向ｄｃ−ｄｃコンバータ、パワーコンディショナ及び分散型電源システム
US9800071B2 (en) * 2015-02-24 2017-10-24 Green Cubes Technology Corporation Methods and system for add-on battery
JP6158241B2 (ja) 2015-03-19 2017-07-05 株式会社豊田中央研究所 電力変換回路システム
CN104753347B (zh) * 2015-03-31 2017-05-03 西安理工大学 一种用于双向dc/dc变换器中电感电流预测独立收敛控制方法
EP3104509A1 (en) * 2015-06-09 2016-12-14 Constructions Electroniques + Telecommunications Dual bridge dc/dc power converter
JP6442744B2 (ja) * 2015-10-08 2018-12-26 正敏 井森 周波数の可変な搬送波が共振回路を駆動する安定化電源
US9973099B2 (en) 2015-08-26 2018-05-15 Futurewei Technologies, Inc. AC/DC converters with wider voltage regulation range
CN106558993A (zh) * 2015-09-30 2017-04-05 株式会社村田制作所 Dc/dc转换装置
FR3042661B1 (fr) * 2015-10-16 2017-12-08 Valeo Systemes De Controle Moteur Convertisseur dc/dc isole
US10658932B2 (en) 2016-01-05 2020-05-19 Redisem Ltd. Power control circuit
FR3047855B1 (fr) * 2016-02-15 2018-03-16 Valeo Siemens Eautomotive France Sas Convertisseur dc/dc isole
TWI602388B (zh) 2016-03-10 2017-10-11 盈正豫順電子股份有限公司 雙向隔離式多階直流－直流電能轉換裝置及其方法
EP3430714B1 (en) * 2016-03-15 2021-06-23 ABB Schweiz AG Bidirectional dc-dc converter and control method therefor
JP6867616B2 (ja) * 2016-05-09 2021-04-28 株式会社アドバンスド・パワー・テクノロジー 電力変換装置
CN106026703B (zh) 2016-05-23 2018-07-13 昂宝电子（上海）有限公司 具有用于同步整流控制器的预测机制的系统和方法
US9960687B2 (en) * 2016-06-06 2018-05-01 General Electric Company System and method for a DC/DC converter
EP3499700A4 (en) * 2016-08-10 2019-07-31 Mitsubishi Electric Corporation POWER CONVERSION DEVICE
JP6848255B2 (ja) * 2016-08-10 2021-03-24 Tdk株式会社 スイッチング電源装置
KR101776770B1 (ko) * 2016-09-08 2017-09-08 현대자동차 주식회사 직류 변환 장치 및 직류 변환 장치의 제어 방법
KR20180059068A (ko) * 2016-11-25 2018-06-04 기아자동차주식회사 차량 및 차량용 직류-직류 변환기
CN106740233B (zh) * 2017-01-23 2019-02-15 苏州协鑫集成科技工业应用研究院有限公司 应用于汽车充电系统的双向拓扑结构
CN106817031B (zh) 2017-02-24 2019-05-28 昂宝电子（上海）有限公司 具有对于同步整流控制器的定时控制的系统和方法
CN108736725A (zh) 2017-04-20 2018-11-02 台达电子工业股份有限公司 电源转换器及其控制方法
TWI636655B (zh) * 2017-04-20 2018-09-21 台達電子工業股份有限公司 電源轉換器及其控制方法
JP6963930B2 (ja) * 2017-08-07 2021-11-10 株式会社Ｇｓユアサ インフラシステムズ Ｄｃ／ｄｃコンバータ
US10892678B2 (en) * 2017-08-09 2021-01-12 Infineon Technologies Austria Ag Method and apparatus for bidirectional operation of phase-shift full bridge converter using inductor pre-charging
DE102018200874A1 (de) * 2017-12-13 2019-01-31 Conti Temic Microelectronic Gmbh Bidirektionaler Gleichspannungswandler sowie Verfahren zum Betreiben eines bidirektionalen Gleichspannungswandlers
CN107911028B (zh) * 2017-12-22 2024-02-02 四川大学 一种加装缓冲器的dc-dc变换器及减小回流功率的方法
CN111713002B (zh) * 2018-02-15 2024-03-08 日本电产株式会社 电力转换装置、驱动装置以及助力转向装置
CN111869076B (zh) * 2018-03-20 2023-10-03 阿尔卑斯阿尔派株式会社 直流电压变换电路以及电源装置
JP7030254B2 (ja) * 2018-06-25 2022-03-07 ダイヤゼブラ電機株式会社 Ｄｃ－ｄｃコンバータ
US10819216B2 (en) 2018-07-26 2020-10-27 Infineon Technologies Austria Ag Power converter with low drain voltage overshoot in discontinuous conduction mode
CN111049378B (zh) * 2018-10-12 2021-08-27 广州汽车集团股份有限公司 一种dc/dc变换器及其控制方法和汽车
US10790754B2 (en) 2019-02-26 2020-09-29 Analog Devices International Unlimited Company Systems and methods for transferring power across an isolation barrier using an active resonator
US10797609B2 (en) 2019-02-26 2020-10-06 Analog Devices International Unlimited Company Systems and methods for transferring power across an isolation barrier using an active self synchronized rectifier
CN110932558B (zh) * 2019-11-29 2021-07-09 华为技术有限公司 电压转换电路、电源系统
CN111146961B (zh) 2020-01-20 2022-04-12 昂宝电子（上海）有限公司 用于控制同步整流系统的控制电路及方法
CN111697838B (zh) 2020-05-29 2023-09-26 昂宝电子（上海）有限公司 同步整流控制电路、方法和开关电源系统
JP7377168B2 (ja) 2020-06-10 2023-11-09 株式会社日立産機システム 共振形電源装置
JP2022002454A (ja) * 2020-06-22 2022-01-06 富士電機株式会社 電力変換装置
CN112491271B (zh) * 2020-11-09 2022-01-14 广州金升阳科技有限公司 一种升降压电路及其控制方法
JP2022121050A (ja) * 2021-02-08 2022-08-19 株式会社東芝 電力変換装置
CN113258807B (zh) * 2021-06-02 2021-10-22 华中科技大学 一种低输出阻抗dc-ac电源
US20220399821A1 (en) * 2021-06-15 2022-12-15 Texas Instruments Incorporated Llc converter and control

Citations (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5255174A (en) * 1991-10-18 1993-10-19 Allied-Signal Inc. Regulated bi-directional DC-to-DC voltage converter which maintains a continuous input current during step-up conversion
US6304461B1 (en) * 2000-06-15 2001-10-16 Supertex, Inc. DC power converter having bipolar output and bi-directional reactive current transfer
US6370050B1 (en) * 1999-09-20 2002-04-09 Ut-Batelle, Llc Isolated and soft-switched power converter
JP2002165448A (ja) 2000-11-20 2002-06-07 Denso Corp 双方向ｄｃ−ｄｃコンバータ
JP2006187147A (ja) 2004-12-28 2006-07-13 Hitachi Ltd 双方向絶縁型ｄｃ−ｄｃコンバータ
US7177163B2 (en) * 2003-03-13 2007-02-13 Honda Motor Co., Ltd Two-way DC-DC converter
US7692935B2 (en) * 2005-12-21 2010-04-06 Hitachi, Ltd. Bi-directional DC-DC converter and control method

Family Cites Families (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4399499A (en) * 1981-12-18 1983-08-16 Gte Automatic Electric Labs Inc. Bi-lateral four quadrant power converter
TWI314808B (en) * 2006-09-06 2009-09-11 Delta Electronics Inc Resonance converter and driving method for synchronous rectifier thereof
TWI326963B (en) * 2006-12-14 2010-07-01 Tungnan Inst Of Technology Resonant converter and synchronous rectification driving circuit thereof
JP4378400B2 (ja) * 2007-08-28 2009-12-02 日立コンピュータ機器株式会社 双方向ｄｃ−ｄｃコンバータ及び双方向ｄｃ−ｄｃコンバータの制御方法

Patent Citations (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5255174A (en) * 1991-10-18 1993-10-19 Allied-Signal Inc. Regulated bi-directional DC-to-DC voltage converter which maintains a continuous input current during step-up conversion
US6370050B1 (en) * 1999-09-20 2002-04-09 Ut-Batelle, Llc Isolated and soft-switched power converter
US6304461B1 (en) * 2000-06-15 2001-10-16 Supertex, Inc. DC power converter having bipolar output and bi-directional reactive current transfer
JP2002165448A (ja) 2000-11-20 2002-06-07 Denso Corp 双方向ｄｃ−ｄｃコンバータ
US7177163B2 (en) * 2003-03-13 2007-02-13 Honda Motor Co., Ltd Two-way DC-DC converter
JP2006187147A (ja) 2004-12-28 2006-07-13 Hitachi Ltd 双方向絶縁型ｄｃ−ｄｃコンバータ
US7692935B2 (en) * 2005-12-21 2010-04-06 Hitachi, Ltd. Bi-directional DC-DC converter and control method

Cited By (32)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US8130515B2 (en) * 2007-08-28 2012-03-06 Hitachi Computer Peripherals Co., Ltd. Bi-directional DC-DC converter and method for controlling the same
US20110128759A1 (en) * 2007-08-28 2011-06-02 Takae Shimada Bi-directional dc-dc converter and method for controlling the same
US20100187909A1 (en) * 2008-07-29 2010-07-29 Odu Steckverbindungssysteme Gmbh & Co. Kg Method and System for Inductively Transmitting Energy and Information
US8228691B2 (en) * 2008-07-29 2012-07-24 Odu Steckverbindungssysteme Gmbh & Co. Kg Method and system for inductively transmitting energy and information
US20100302824A1 (en) * 2009-05-29 2010-12-02 Sony Corporation Power source apparatus
US8305785B2 (en) * 2009-05-29 2012-11-06 Sony Corporation Power source apparatus
US20110149609A1 (en) * 2009-12-21 2011-06-23 Intersil Americas Inc. Bidirectional signal conversion
US20110149610A1 (en) * 2009-12-21 2011-06-23 Intersil Americas Inc. Bidirectional signal conversion
US20110149611A1 (en) * 2009-12-21 2011-06-23 Intersil Americas Inc. Bidirectional signal conversion
US8503194B2 (en) 2009-12-21 2013-08-06 Intersil Americas LLC Bidirectional signal conversion
US8570769B2 (en) * 2009-12-21 2013-10-29 Intersil Americas LLC Bidirectional signal conversion
US20110176335A1 (en) * 2010-01-21 2011-07-21 Li Zeng Resonant converters and burst mode control method thereof
US8582321B2 (en) * 2010-01-21 2013-11-12 Delta Electronics, Inc. Resonant converters and burst mode control method thereof
US8593831B2 (en) * 2010-10-28 2013-11-26 Eltek Valere As Method for controlling a series resonant DC/DC converter
WO2014093038A1 (en) * 2012-11-28 2014-06-19 Guangdong Rui Ding Electrical Technology LTD. Super-efficient single-stage isolated switching power amplifier
US9762127B2 (en) * 2013-03-18 2017-09-12 Panasonic Intellectual Property Management Co., Ltd. Power converter and power conditioner
US20160072390A1 (en) * 2013-04-26 2016-03-10 Panasonic Intellectual Property Management Co., Ltd. Bidirectional dc/dc converter
US9570991B2 (en) * 2013-04-26 2017-02-14 Panasonic Intellectual Property Management Co., Ltd. Bidirectional DC/DC converter
US9774269B2 (en) 2013-07-11 2017-09-26 Fuji Electric Co., Ltd. Bidirectional DC/DC converter
US20150131337A1 (en) * 2013-09-09 2015-05-14 Aerovironment, Inc. Resonant Isolated Active PFC Rectifier
US9479048B2 (en) * 2013-09-09 2016-10-25 Aerovironment, Inc. Resonant isolated active power factor correction (PFC) rectifier
US20160315557A1 (en) * 2015-04-23 2016-10-27 Panasonic Intellectual Property Management Co., Ltd. Power converter
US9673735B2 (en) * 2015-04-23 2017-06-06 Panasonic Intellectual Property Management Co., Ltd. Power converter
US20190280526A1 (en) * 2016-10-19 2019-09-12 Amosense Co., Ltd Magnetic induction power supply device
US10923951B2 (en) * 2016-10-19 2021-02-16 Amosense Co., Ltd Magnetic induction power supply device
US9906150B1 (en) * 2016-11-09 2018-02-27 Astec International Limited Open loop bi-directional converters and corresponding control methods
US11095219B2 (en) * 2017-08-22 2021-08-17 Diamond Electric Mfg. Co., Ltd. Converter with soft switching function
CN109672344B (zh) * 2019-01-21 2020-06-26 中国科学院电工研究所 双向直流变换器及其控制方法
CN109672344A (zh) * 2019-01-21 2019-04-23 中国科学院电工研究所 双向直流变换器及其控制方法
US11511637B2 (en) 2019-05-24 2022-11-29 Huawei Digital Power Technologies Co., Ltd. Integrated charger and motor control system
US20230246556A1 (en) * 2022-01-28 2023-08-03 Analog Devices, Inc. Active-clamp current-fed push-pull converter for bidirectional power transfer
US11804782B2 (en) * 2022-01-28 2023-10-31 Analog Devices, Inc. Active-clamp current-fed push-pull converter for bidirectional power transfer

Also Published As

Publication number Publication date
JP4378400B2 (ja) 2009-12-02
CN102624239B (zh) 2015-02-11
TW200922093A (en) 2009-05-16
CN102624239A (zh) 2012-08-01
TWI368384B (en) 2012-07-11
US20110128759A1 (en) 2011-06-02
US7911810B2 (en) 2011-03-22
CN101378232B (zh) 2012-05-30
US20090059622A1 (en) 2009-03-05
CN101378232A (zh) 2009-03-04
JP2009055747A (ja) 2009-03-12
US20110019440A1 (en) 2011-01-27
US8130515B2 (en) 2012-03-06

Similar Documents

Publication Publication Date Title
US7848118B2 (en) Bi-directional DC-DC converter and method for controlling the same
US9431917B2 (en) Switching power supply including a rectifier circuit having switching elements, and electric power converter
KR101395487B1 (ko) Dc-dc 컨버터
CN101755380B (zh) 具有续流mosfet的降压开关稳压器
US9487098B2 (en) Power conversion apparatus
US20100019807A1 (en) Current-source gate driver
US10581318B2 (en) Resonant converter including capacitance addition circuits
CN103595253A (zh) 一种降低mosfet开关损耗的新型控制方法
US20100220500A1 (en) Power converter and method for controlling power converter
US20140152276A1 (en) Dc/dc voltage converter and voltage conversion control method therefor
US20130207626A1 (en) Switching Power Supply Circuit
US10090752B2 (en) Power conversion device
KR20100115087A (ko) 양방향 ｄｃ－ｄｃ 컨버터 및 그의 제어방법
US10483853B2 (en) DC-DC converter
US20200366198A1 (en) Converter
US10305367B2 (en) Power conversion circuit
JP6132882B2 (ja) 電力変換装置
JP2019009848A (ja) Ｄｃ−ｄｃコンバータ、これを用いた電源システム及び当該電源システムを用いた自動車
US11799372B2 (en) Driving apparatus and driving method for semiconductor switch, power conversion apparatus, and vehicle
WO2011161728A1 (ja) スイッチング電源装置およびこれを用いた電源システム、電子装置
JP4328417B2 (ja) 電源回路
CN216794868U (zh) 自激式主动箝位电路
CN218473035U (zh) 反激变换电路、变换器、伺服电机及其驱动器
JP4544076B2 (ja) Ｄｃ−ｄｃコンバータおよびｄｃ−ｄｃコンバータの制御方法
JP2006101589A (ja) Ｄｃ−ｄｃコンバータ

Legal Events

Date Code Title Description
AS Assignment

Owner name: HITACHI COMPUTER PERIPHERALS CO., LTD., JAPAN

Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:SHIMADA, TAKAE;TAKAHASHI, YOSHIHIDE;TANIGUCHI, KIMIAKI;AND OTHERS;REEL/FRAME:021403/0079;SIGNING DATES FROM 20080725 TO 20080729

Owner name: HITACHI COMPUTER PERIPHERALS CO., LTD., JAPAN

Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:SHIMADA, TAKAE;TAKAHASHI, YOSHIHIDE;TANIGUCHI, KIMIAKI;AND OTHERS;SIGNING DATES FROM 20080725 TO 20080729;REEL/FRAME:021403/0079

STCF Information on status: patent grant

Free format text: PATENTED CASE

AS Assignment

Owner name: HITACHI INFORMATION & TELECOMMUNICATION ENGINEERIN

Free format text: MERGER;ASSIGNOR:HITACHI COMPUTER PERIPHERALS CO., LTD.;REEL/FRAME:031612/0272

Effective date: 20130401

FPAY Fee payment

Year of fee payment: 4

MAFP Maintenance fee payment

Free format text: PAYMENT OF MAINTENANCE FEE, 8TH YEAR, LARGE ENTITY (ORIGINAL EVENT CODE: M1552)

Year of fee payment: 8

FEPP Fee payment procedure

Free format text: MAINTENANCE FEE REMINDER MAILED (ORIGINAL EVENT CODE: REM.); ENTITY STATUS OF PATENT OWNER: LARGE ENTITY

LAPS Lapse for failure to pay maintenance fees

Free format text: PATENT EXPIRED FOR FAILURE TO PAY MAINTENANCE FEES (ORIGINAL EVENT CODE: EXP.); ENTITY STATUS OF PATENT OWNER: LARGE ENTITY

STCH Information on status: patent discontinuation

Free format text: PATENT EXPIRED DUE TO NONPAYMENT OF MAINTENANCE FEES UNDER 37 CFR 1.362

FP Lapsed due to failure to pay maintenance fee

Effective date: 20221207