US20210186553A1

US20210186553A1 - Medical device

Info

Publication number: US20210186553A1
Application number: US17/150,119
Authority: US
Inventors: Alan Edward Green; Paul Christopher Roberts; Mark David Tuckwell; Peter Ralph Bonham
Original assignee: Cilag GmbH International
Current assignee: Cilag GmbH International
Priority date: 2013-12-16
Filing date: 2021-01-15
Publication date: 2021-06-24
Also published as: GB2521228A; CN105992563A; WO2015094749A1; US10912580B2; EP3082627A1; CN105992563B; JP6483130B2; US20160324537A1; BR112016014030A2; BR112016014030B1; US20210186554A1; JP2017500940A; GB201322210D0

Abstract

A medical device is described having a handle, a shaft coupled to the handle and an end effector coupled to the shaft. In one embodiment, the device includes an ultrasonic transducer and is arranged so that ultrasonic or electrical energy can be delivered to a vessel or tissue to be treated. Various novel sensing circuits are described to allow a measure of the drive signal to be measured and fed back to a controller. An active fuse circuit is also described for protecting one or more batteries of the device from an over-current situation.

Description

The present invention relates to the field of medical devices and in particular, although not exclusively, to medical cauterization and cutting devices. The invention also relates to drive circuits and methods for driving such medical devices.
Many surgical procedures require cutting or ligating blood vessels or other internal tissue and many procedures are performed using minimally invasive techniques with a hand-held cauterization device to perform the cutting or ligating. Some existing hand-held cauterization devices use an ultrasonic transducer in the cauterization device to apply ultrasonic energy to the tissue to be cut or ligated. Other hand-held cauterization devices apply RF energy directly to the tissue/vessel being cauterized via forceps of the device.
The present invention aims to provide an alterative surgical device that is able to apply ultrasonic energy or RF energy to the vessel or tissue to be cauterized. Other aspects of the invention relate to the way in which control circuitry is provided to select between the different operating modes. Other aspects of the invention relate to the way in which voltage and current measurements can be made in the circuit design for reporting to a controller, such as a microprocessor; and to the way in which control circuitry can be provided to ensure that too much current is not drawn from the battery.
According to one aspect, the present invention provides a medical device comprising: an end effector for gripping a vessel/tissue; an ultrasonic transducer coupled to the end effector; a drive circuit coupled to the end effector and to the ultrasonic transducer and operable to generate a periodic drive signal and to provide the drive signal either to the ultrasonic transducer or to the end effector; and a controller operable to control the drive circuit so that the drive signal is applied to a desired one of the ultrasonic transducer and the end effector.
In one embodiment, the drive circuit comprises a first resonant circuit having a first resonant frequency and a second resonant circuit having a second resonant frequency that is different to the first resonant frequency, wherein the first resonant frequency corresponds to a resonant characteristic of the ultrasonic transducer and wherein the controller is operable to control the drive circuit so that the drive circuit generates a drive signal having a frequency corresponding to the first resonant frequency when the drive signal is to be applied to the ultrasonic transducer and so that the drive circuit generates a drive signal having a frequency corresponding to the second resonant frequency when the drive signal is to be applied to the end effector.
A signal generator may also be provided that is coupled between the controller and the drive circuit for generating a cyclically varying voltage from a DC voltage supply in dependence upon control signals from the controller and for supplying the cyclically varying voltage to the first and second resonant circuits of the drive circuit.
The controller may be arranged to vary the period of the drive signal about the first resonant frequency or the second resonant frequency to vary the energy supplied to the vessel or tissue gripped by the end effector. The controller may vary the period of the drive signal so that the frequency of the drive signal varies around the first resonant frequency within 0.1% to 1% of the first resonant frequency; or so that the frequency of the drive signal varies around the second resonant frequency within 40% to 60% of the second resonant frequency.
Typically, the resonant characteristics of the first and second resonant circuits vary with the tissue or vessel gripped by the forceps and in one embodiment, the controller is configured to vary the period of the drive signal to track changes in the respective resonant characteristic.
An ultrasonic waveguide may be provided that is coupled to the ultrasonic transducer for guiding ultrasonic energy generated by the ultrasonic transducer towards the end effector. The end effector may comprise first and second jaws and the second resonant circuit may be electrically coupled to the first and second jaws of the end effector. For example, the first jaw of the end effector may be electrically coupled to the waveguide and the second resonant circuit may be electrically coupled to the first jaw of the end effector via the ultrasonic waveguide. In some embodiments, the first resonant circuit is electrically coupled to the ultrasonic transducer and to the waveguide.
Sensing circuitry may be provided for sensing a drive signal applied to the ultrasonic transducer or to the end effector. In one embodiment, one or both of the first and second resonant circuits may comprise at least one of an inductor coil, a capacitor and a resistor and wherein the sensing circuitry may comprise an op-amp circuit for sensing the voltage across the inductor coil or the capacitor or the resistor and for converting the sensed voltage to a sensor signal suitable for inputting to the controller. In an alternative embodiment, one or both of the first and second resonant circuits may comprise an impedance element that is connected between the resonant circuit and a reference potential and wherein the sensing circuitry comprises a divider circuit for obtaining a measure of the voltage across the impedance element and a bias signal generator for applying a DC bias signal to the voltage measure. In this case, the impedance element may comprise a capacitor or a resistor. Typically, the sensing circuitry comprises DC blocking circuitry for preventing the DC bias signal from the bias signal generator from coupling with the drive circuit. The bias signal generator may comprises a voltage divider circuit connected between a reference voltage and a supply voltage of the controller.
The device is preferably a battery operated device and comprises one or more batteries for powering the device and further comprising an active fuse circuit for protecting the one or more batteries. The active fuse circuit may comprise a switch electrically coupled between a terminal of the one or more batteries and the drive circuit and control circuitry configured to open the switch to isolate the supply terminal from the drive circuit.
The present invention also provides a medical device comprising: an end effector for gripping a vessel/tissue; a drive circuit for generating a cyclically varying drive signal for driving energy into the vessel/tissue; sensing circuitry for sensing a drive signal generated by the drive circuit; and a controller responsive to the sensing circuitry and operable to control the drive circuit to control the energy delivered to the vessel/tissue; wherein the drive circuit comprises an impedance element that is coupled to a reference potential and wherein the sensing circuitry comprises a divider circuit for obtaining a measure of the voltage across the impedance element and a bias signal generator for applying a DC bias signal to the voltage measure.
The impedance element may be a capacitor or a resistor. The sensing circuitry may also comprise DC blocking circuitry for preventing the DC bias signal from the bias signal generator from coupling with the drive circuit. The bias signal generator may comprise a divider circuit connected between a reference voltage and a supply voltage of the controller. The divider circuit of the bias signal generator may be connected to the supply voltage of the controller via a switch and wherein the controller is configured to open the switch when the controller does not require signals from the sensing circuitry.
The present invention also provides a medical device comprising: an end effector for gripping a vessel/tissue; one or more batteries for providing a DC voltage supply; a signal generator coupled to the one or more batteries for generating a cyclically varying drive signal from the DC voltage supply for driving energy into the vessel/tissue; a controller operable to control the signal generator to control the energy delivered to the vessel/tissue; and an active fuse circuit coupled between the one or more batteries and the signal generator for protecting the one or more batteries.
The active fuse circuit may comprise a switch that is electrically coupled between a terminal of the one or more batteries and the signal generator; and control circuitry configured to switch the switch. The switch may be arranged to disconnect the signal generator from the one or more batteries or may connect a large impedance between the signal generator and the one or more batteries.
The control circuitry of the active fuse may comprise circuitry for sensing a measure of the current being drawn from the one or more batteries and is configured to switch the switch in the event that the current measure exceeds a threshold. The control circuitry of the active fuse may comprise a comparator for comparing the current measure with the threshold and wherein an output of the comparator controls the opening and closing of the switch.
The present invention also provides a method of operating a medical device comprising generating a periodic drive signal and applying the drive signal to an ultrasonic transducer or to an end effector of the medical device and controlling the drive circuit so that the drive signal is applied to a desired one of the ultrasonic transducer and the end effector.
The present invention also provides a method of cauterising or cutting a vessel or tissue, the method comprising: gripping the vessel or tissue with an end effector of a medical device; using a drive circuit to apply a periodic drive signal either to an ultrasonic transducer or to the end effector; and controlling the drive circuit so that the drive signal is applied to a desired one of the ultrasonic transducer and the end effector. The method may use the above described medical device.
The present invention also provides electronic apparatus for use in a medical device having an ultrasonic transducer and an end effector, the electronic apparatus comprising: a drive circuit for generating a periodic drive signal; and a controller operable to control the drive circuit so that the drive signal is applied to a desired one of the ultrasonic transducer and the end effector; wherein the drive circuit comprise a first resonant circuit having a first resonant frequency and a second resonant circuit having a second resonant frequency that is different to the first resonant frequency, and wherein the controller is operable to control the drive circuit so that the drive circuit generates a drive signal having a frequency corresponding to the first resonant frequency when the drive signal is to be applied to the ultrasonic transducer and so that the drive circuit generates a drive signal having a frequency corresponding to the second resonant frequency when the drive signal is to be applied to the end effector.
The present invention also provides a medical device comprising: an end effector for gripping a vessel/tissue; a drive circuit coupled to the end effector and operable to generate a drive signal and to provide the drive signal to the end effector; a controller operable to generate and output control signals to the drive circuit to control the drive signal generated by the drive circuit; wherein the drive circuit and a load formed by the vessel/tissue gripped by the end effector define a resonant circuit whose resonant frequency varies as the impedance of the load formed by the vessel/tissue gripped by the end effector changes; wherein the controller is arranged to generate control signals which cause the drive circuit to generate a drive signal having a frequency that tracks said resonant frequency as it changes; and wherein said controller is further arranged to reduce one or more of the power, current or voltage delivered to the load formed by the vessel/tissue gripped by the end effector.
Sensor circuitry may be provided for sensing signals applied to the load formed by the vessel/tissue gripped by the end effector and measurement circuitry for processing the signals from the sensor circuitry to determine a measure of the impedance of the load formed by the vessel/tissue gripped by the end effector. In this case, the controller can generate said control signals in dependence upon said measure of the impedance of the load formed by the vessel/tissue gripped by the end effector.
In one embodiment, the controller generates control signals having sequences of pulses and the controller skips one or more pulses from the control signals in order to reduce one or more of the power, current or voltage delivered to the load formed by the vessel/tissue gripped by the end effector.
Typically, in this case, the controller comprises a pulse signal generator that generates pulses at a desired frequency that depends on said resonant frequency and the controller skips pulses generated by said pulse signal generator by suppressing pulses generated by the pulse signal generator.

These and various other features and aspects of the invention will become apparent from the following detailed description of embodiments which are described with reference to the accompanying Figures in which:

FIG. 1 illustrates a hand-held cauterization device that has batteries and drive and control circuitry mounted in a handle portion of the device;

FIG. 2 is a part block diagram illustrating the main components of the cauterization device used in one embodiment of the invention;

FIG. 3 is a circuit diagram illustrating the main electrical components of the cauterization device shown in FIG. 2;

FIG. 4 schematically illustrates the way in which the ultrasonic transducer is coupled to a waveguide for delivering the generated ultrasonic energy to the forceps and illustrating the way in which the circuitry shown in FIG. 3 can deliver electrical energy to the forceps;

FIG. 5 is a block diagram that schematically illustrates processing modules that form part of the microprocessor shown in FIG. 2;

FIG. 6 illustrates the form of control signals generated by the microprocessor to control the drive circuit whilst minimising 3^rdharmonic content;

FIG. 7 is a contour plot illustrating the delivered power versus the load resistance and the drive frequency;

FIG. 8a is a circuit diagram illustrating one way in which a measure of the load current can be determined and supplied to the microprocessor;

FIG. 8b is a circuit diagram illustrating another way in which a measure of the load current can be determined and supplied to the microprocessor;

FIG. 8c is a circuit diagram illustrating one way in which a measure of the load voltage can be determined and supplied to the microprocessor;

FIG. 9a is a circuit diagram illustrating one way in which a measure of the load current can be determined and supplied to the microprocessor without using an op-amp circuit;

FIG. 9b is a circuit diagram illustrating one way in which a measure of the load current can be determined and supplied to the microprocessor and illustrating one way in which a measure of the load voltage can be determined and supplied to the microprocessor without using op-amp circuits; and

FIG. 10 is a circuit diagram illustrating an active fuse circuit used to protect the batteries shown in FIG. 2 from excessive current demand.

MEDICAL DEVICE

Many surgical procedures require cutting or ligating blood vessels or other vascular tissue. With minimally invasive surgery, surgeons perform surgical operations through a small incision in the patient's body. As a result of the limited space, surgeons often have difficulty controlling bleeding by clamping and/or tying-off transected blood vessels. By utilizing ultrasonic-surgical forceps or electro-surgical forceps, a surgeon can cauterize, coagulate/desiccate, and/or simply reduce bleeding by controlling the ultrasonic energy applied to the tissue/vessel by an ultrasonic transducer or by controlling the RF energy applied to the tissue/vessel via the forceps.
FIG. 1 illustrates the form of an ultrasonic/RF-surgical medical device 1 that is designed for minimally invasive medical procedures, according to one embodiment of the present invention. As shown, the device 1 is a self contained device, having an elongate shaft 3 that has a handle 5 connected to the proximal end of the shaft 3 and an end effector 7 connected to the distal end of the shaft 3. In this embodiment, the end effector 7 comprises medical forceps 9 that are controlled by the user manipulating control levers 11 and 13 of the handle 5.
During a surgical procedure, the shaft 3 is inserted through a trocar to gain access to the patient's interior and the operating site. The surgeon will manipulate the forceps 9 using the handle 5 and the control levers 11 and 13 until the forceps 9 are located around the vessel to be cut or cauterised. Electrical energy is then applied, in a controlled manner, either to the tissue directly via the forceps 9 (as RF energy) or to an ultrasonic transducer 8 that is mounted within the handle 5 and coupled to the forceps 9 via a waveguide (not shown) within the shaft 3, in order to perform the desired cutting/cauterisation using ultrasonic energy. As shown in FIG. 1, in this embodiment, the handle 5 also houses batteries 15 and control electronics 17 for generating and controlling the electrical energy required to perform the cauterisation. In this way, the device 1 is self contained in the sense that it does not need a separate control box and supply wire to provide the electrical energy to the forceps 9. However, such a separate control box may be provided if desired.

System Circuitry

FIG. 2 is a schematic block diagram illustrating the main electrical circuitry of the cauterization/cutting device 1 used in this embodiment to generate and control the electrical energy supplied to the ultrasonic transducer or to the forceps 9. As will be explained in more detail below, in this embodiment, the circuitry is designed to control the period of an electrical drive waveform that is generated in order to control the amount of power delivered to the tissue/vessel being cauterized.
As shown in FIG. 2, the cauterization/cutting device 1 comprises a user interface 21—via which the user is provided with information (such as an indication that energy is being applied to the gripped tissue/vessel by electrical energy or ultrasonic energy) and through which the user controls the operation of the cauterization/cutting device 1, including selection of ultrasonic operation or RF operation. As shown, the user interface 21 is coupled to a microprocessor 23 that controls the cutting/cauterisation procedure by generating control signals that it outputs to gate drive circuitry 25. In response to the control signals from the microprocessor 23, the gate drive circuitry 25 generates gate control signals that cause a bridge signal generator 27 to generate a desired drive waveform that is applied either to the ultrasonic transducer 8 or to the forceps 9 via a drive circuit 29. Voltage sensing circuitry 31 and current sensing circuitry 33 generate measures of the current and voltage applied to the ultrasonic transducer 8 or to the forceps 9, which they feed back to the microprocessor 23 for control purposes. FIG. 2 also shows the batteries 15 that provide the power for powering the electrical circuitry shown in FIG. 2. In this embodiment, the batteries 15 are arranged to supply 0V and 14V rails.
FIG. 3 illustrates in more detail the components of the gate drive circuitry 25, the bridge signal generator 27 and the drive circuit 29. FIG. 3 also shows an electrical equivalent circuit 30 of the piezo-electric ultrasonic transducer 8 and the load (R_load) formed by the tissue/vessel to be treated. As shown in FIG. 3, the gate drive circuitry 25 includes two FET gate drives 37—FET gate drive 37-1 and FET gate drive 37-2. A first set of control signals (CTRL₁) from the microprocessor 23 is supplied to FET gate drive 37-1 and a second set of control signals (CTRL₂) from the microprocessor 23 is supplied to FET gate drive 37-2. FET gate drive 37-1 uses the first set of control signals (CTRL₁) to generate two drive signals—one for driving each of the two FETs 41-1 and 41-2 of the bridge signal generator 27. The FET gate drive 37-1 generates drive signals that causes the upper FET (41-1) to be on when the lower FET (41-2) is off and vice versa. This causes the node A to be alternately connected to the 14V rail (when FET 41-1 is switched on) and the 0V rail (when the FET 41-2 is switched on). Similarly, FET gate drive 37-2 uses the second set of control signals (CTRL₂) to generate two drive signals—one for driving each of the two FETs 41-3 and 41-4 of the bridge signal generator 27. The FET gate drive 37-2 generates drive signals that causes the upper FET (41-3) to be on when the lower FET (41-4) is off and vice versa. This causes the node B to be alternately connected to the 14V rail (when FET 41-3 is switched on) and the 0V rail (when the FET 41-4 is switched on). Thus the two sets of control signals (CTRL₁and CTRL₂) output by the microprocessor 23 control the digital waveform that is generated and applied between nodes A and B. Each set of control signals (CTRL₁and CTRL₂) comprises of a pair of signal lines, one to indicate when the high side FET is on and the other to indicate when the low side FET is on. Thus the microprocessor 23, either through software or through a dedicated hardware function can ensure that the undesirable condition when both high and low side FETs are simultaneously turned on does not occur. In practice this requires leaving a dead time when both high and low side FETs are turned off to ensure that, even when allowing for variable switching delays, there is no possibility that both FETs can be simultaneously on. In the present embodiment a dead time of about 100 ns was used.
As shown in FIG. 3, the nodes A and B are connected to the drive circuit 29, thus the digital voltage generated by the bridge signal generator 27 is applied to the drive circuit 29. This applied voltage will cause current to flow in the drive circuit 29. As shown in FIG. 3, the drive circuit 29 includes two transformer circuits 42-1 and 42-2. The first transformer circuit 42-1 is designed for efficient driving of the ultrasonic transducer 8 and includes a capacitor-inductor-inductor resonant circuit 43-1 formed by capacitor C ^US _s 45, inductor L ^US _s 47 and inductor L ^US _m 49. When driving the ultrasonic transducer 8, the microprocessor 23 is arranged to generate control signals for the gate drive circuitry 25 so that the fundamental frequency (f_d) of the digital voltage applied across nodes A and B is around the resonant frequency of the resonant circuit 43-1, which in this embodiment is about 50 kHz. As a result of the resonant characteristic of the resonant circuit 43-1, the digital voltage applied across nodes A and B will cause a substantially sinusoidal current at the fundamental frequency (f_d) to flow within the resonant circuit 43-1. This is because higher harmonic content of the drive voltage will be attenuated by the resonant circuit 43-1 and the impedance of L_tand C_t1referred to the transformer primary.
As illustrated in FIG. 3, the inductor L ^US _m 49 forms the primary of the transformer circuit 42-1, the secondary of which is formed by inductor L ^US _sec 53. The transformer up-converts the drive voltage (V^US _d) across the inductor L _m 49 to a load voltage (V_L; typically about 120 volts) that is applied to the ultrasonic transducer 8. The electrical characteristics of the ultrasonic transducer 8 change with the impedance of the forceps' jaws and any tissue or vessel gripped by the forceps 9; and FIG. 3 models the ultrasonic transducer 8 and the impedance of the forceps' jaws and any tissue or vessel gripped by the forceps 9 by the inductor L_t 57, the parallel capacitors C _t1 59 and C _t2 61 and the resistance R_load.
The inductor L^US _sand capacitor C^US _sof the drive circuit 29 are designed to have a matching LC product to that of inductor L_tand capacitor C_t1of the ultrasonic transducer 8. Matching the LC product of a series LC network ensures that the resonant frequency of the network is maintained. Similarly, the magnetic reactance of the inductor L^US _mis chosen so that at resonance it matches with the capacitive reactance of the capacitor C_t2of the ultrasonic transducer 8. For example, if the transducer 8 is defined such that capacitor C_t2has a capacitance of about 3.3 nF, then the inductor L^US _mshould have an inductance of about 3 mH (at a resonant frequency of about 50 kHz). Designing the drive circuit 29 in this way provides for the optimum drive efficiency in terms of energy delivery to the tissue/vessel gripped by the forceps 9. The efficiency improvement is realised because the current flowing in C^US _sand consequently the FET bridge (27) is reduced, because the transformer magnetising current cancels out the current flowing in C_t2In addition, because of this current cancellation, the current flowing in C^US _sis proportional to the current flowing in Rload, which allows the load current to be determined by measuring the current flowing in C^US _s.
The second transformer circuit 42-2 is designed for efficient driving of electrical RF energy directly to the tissue/vessel via the forceps 9 and includes a capacitor-inductor-inductor resonant circuit 43-2 formed by capacitor C^F _s 46, inductor L ^F _s 48 and inductor L ^F _m 50. When driving the forceps 9 directly with electrical energy, the microprocessor 23 is arranged to generate control signals for the gate drive circuitry 25 so that the fundamental frequency (f_d) of the digital voltage applied across nodes A and B is around the resonant frequency of the resonant circuit 43-2, which in this embodiment is about 500 kHz. As a result of the resonant characteristic of the resonant circuit 43-2, the digital voltage applied across nodes A and B will cause a substantially sinusoidal current at the fundamental frequency (f_d) to flow within the resonant circuit 43-2. This is because higher harmonic content of the drive voltage will be attenuated by the resonant circuit 43-2.
As illustrated in FIG. 3, the inductor L ^F _m 50 forms the primary of the transformer circuit 42-1, the secondary of which is formed by inductor L ^US _sec 54. The transformer up-converts the drive voltage (V^F _d) across inductor L ^F _m 50 to the load voltage (V^F _L; typically about 120 volts) that is applied to the forceps 9. The tissue or vessel gripped by the jaws of the forceps 9 is represented as the resistive load R_loadin the box labelled 9 in FIG. 3. In practice, this will be the same resistive load that is illustrated in the electrical equivalent circuit 30 of the ultrasonic transducer 8.
FIG. 4 is a schematic diagram illustrating the way in which the ultrasonic transducer 8 couples to the tissue/vessel to be cauterized and the way in which the circuit components illustrated in FIG. 3 connect to the ultrasonic transducer 8 and to the forceps 9. In particular, FIG. 4 shows the shaft 3, the forceps 9 and the ultrasonic transducer 8. FIG. 4 also shows the waveguide 72 along which the ultrasonic signal that is generated by the ultrasonic transducer 8 is guided. The waveguide 72 is connected to the node “BB” show in FIG. 3, whilst the input supply to the ultrasonic transducer 8 is connected to node “AA” shown in FIG. 3. The output node “CC” of the second transformer circuit 42-2 is connected to a conductive inner wall of the sheath 3, which is electrically connected to the upper jaw 74 of the forceps 9. The return path is through the tissue/vessel to be cauterized and the lower jaw 76, which is electrically connected to the node “BB”.
When the drive signal has a drive frequency of about 50 kHz, very little current will flow within the second transformer circuit 42-2 because the drive frequency is far away from the resonant frequency of the resonant circuit 43-2 such that the input impedance of the second transformer circuit 42-2 will be very high for this drive signal. Therefore, the power will be delivered almost entirely via the first transformer circuit 42-1. Similarly, when the drive signal has a drive frequency of about 500 kHz, very little current will flow within the first transformer circuit 42-1 because the drive frequency is far away from the resonant frequency of the resonant circuit 43-1 such that the input impedance of the first transformer circuit 42-1 will be very high for this drive signal. Therefore, the power will be delivered almost entirely via the second transformer circuit 42-2. In this way, the two transformer circuits 42-1 and 42-2 can be driven by a common bridge signal generator 27; although it is also feasible to drive each transformer circuit with separate bridge signal generators.
It is not always desired to apply full power to the tissue/vessel to be treated. Therefore, in this embodiment in the ultrasonic mode of operation, the amount of ultrasonic energy supplied to the vessel/tissue is controlled by varying the period of the digital waveform applied across nodes A and B so that the drive frequency (f_d) moves away from the resonant frequency of the ultrasonic transducer 8. This works because the ultrasonic transducer 8 acts as a frequency dependent (lossless) attenuator. The closer the drive signal is to the resonant frequency of the ultrasonic transducer 8, the more ultrasonic energy the ultrasonic transducer 8 will generate. Similarly, as the frequency of the drive signal is moved away from the resonant frequency of the ultrasonic transducer 8, less and less ultrasonic energy is generated by the ultrasonic transducer 8. In addition or instead, the duration of the pulses of the drive signals may be varied to control the amount of ultrasonic energy delivered to the tissue/vessel.
Similarly, in the electrical mode of operation, the amount of electrical power supplied to the forceps 9 is controlled by varying the period of the digital waveform applied across nodes A and B so that the drive frequency (f_d) moves away from the resonant frequency of the resonant circuit 43-2. This works because the resonant circuit 43-1 acts as a frequency dependent (lossless) attenuator. The closer the drive signal is to the resonant frequency of the resonant circuit 43-1, the less the drive signal is attenuated. Conversely, as the frequency of the drive signal is moved away from the resonant frequency of the circuit 43-1, the more the drive signal is attenuated and so the electrical energy supplied to the tissue/vessel reduces. The drive frequency needs to move away from the resonant frequency by about 50% of the resonant frequency to achieve the desired range of power variation. An alternative approach to controlling the power, current or voltage applied during the electrical mode of operation is to continuously tune the frequency of the excitation signal to keep it matched with the resonant frequency of the drive circuit (as it changes with the changing load impedance) and thereby maintain efficient operation and to skip some of the pulses of the drive control signals until the average power, current and/or voltage is below the relevant limit.
A further alternative, which is most effective when using the pulse skipping technique, is to remove the inductor 48 shown in FIG. 3 and therefore the drive circuit for the electrical mode of operation becomes a substantially parallel LC resonant circuit (it is not a pure parallel LC resonant circuit because the transformer leakage inductance appears in series with inductor 48 and cannot be entirely removed). The advantage of removing the inductor 48 is that the overall efficiency can be increased, because there are no longer any losses in the inductor. A further advantage is that the physical size of the circuit can be reduced, because often the inductor is a physically large component relative to the FETs, microprocessor, capacitors and other system components.
The microprocessor 23 controls the power delivery based on a desired power to be delivered to the circuitry 30 (which models the ultrasonic transducer 8 and the tissue/vessel gripped by the forceps 9) or to the forceps 9 and based on measurements of the load voltage (V_L) and of the load current (i_L) obtained from the voltage sensing circuitry 31 and the current sensing circuitry 33. The microprocessor 23 also selects the frequency of the drive signal (around 50 kHz or around 500 kHz) based on a user input received via the user interface 21 that selects either electrical operation or ultrasonic operation.

Microprocessor

FIG. 5 is a block diagram illustrating the main components of the microprocessor 23 that is used in this embodiment. As shown, the microprocessor 23 includes synchronous I,Q sampling circuitry 81 that receives the sensed voltage and current signals from the sensing circuitry 31 and 33 and obtains corresponding samples which are passed to a measured voltage and current processing module 83. The measured voltage and current processing module 83 uses the received samples to calculate the impedance of, and the RMS voltage applied to and the RMS current flowing through, the ultrasonic transducer 8 and/or directly to the tissue/vessel gripped by the forceps 9; and from them the power that is presently being supplied to the circuitry 30 or directly to the tissue/vessel gripped by the forceps 9. The determined values are then passed to a power controller 85 for further processing. The measured voltage and current processing module 83 can also process the received I and Q samples to calculate the phase difference between the load voltage (V_L) and the load current (i_L). During the ultrasonic mode of operation, at resonance, this phase difference should be around zero and so this phase measure can be used as a feedback parameter for the power controller 85.
The power controller 85 uses the received impedance value and the delivered power value to determine, in accordance with a predefined algorithm and a power set point value and a mode indication signal (received from a medical device control module 89 and indicating ultrasonic operation or electrical operation), a desired period/frequency (Δt_new) of the control signals (CTRL₁and CTRL₂) that are used to control the gate drive circuit 25. This desired period/frequency is passed from the power controller 85 to the control signal generator 95, which changes the control signals CTRL₁and CTRL₂in order to change the waveform period to match the desired period. The CTRL control signals may comprise square wave signals having the desired period or they may comprise periodic pulses with the period corresponding to the desired period (Δt_new) and with the relative timing of the pulses of the control signals being set to minimise harmonic content of the waveform that is generated by the bridge signal generator 27 (such as to minimise the 3^rdorder harmonic). In this embodiment, the control signals CTRL₁are output to the FET gate drive 37-1 (shown in FIG. 2), which amplifies the control signals and then applies them to the FETs 41-1 and 41-2; and the control signals CTRL₂are output to the FET gate drive 37-2 (shown in FIG. 2), which amplifies the control signals and then applies them to the FETs 41-3 and 41-4, to thereby generate the desired waveform with the new period (Δt_new).
In order to drive the circuitry with the optimal RMS waveform, the MOSFETs 41 are driven as complementary, opposing pairs. Although the maximum output voltage is achieved when the MOSFET pairs are driven at a phase shift of 180 degrees, the resulting harmonic content of such a drive waveform, particularly the 3rd harmonic (which is poorly excluded by the output filter) is quite high. The inventors have found that the optimal phase shift between the control signals applied to the two pairs of MOSFETs 41, for 3rd harmonic reduction, is around 120°. This is illustrated in FIG. 6 which shows in the upper plot the output from the first MOSFET pair 41-1 and 41-2; in the middle plot the output from the second MOSFET pair 41-3 and 41-4 (shifted by 120° relative to the upper plot); and in the lower plot the resulting (normalised) output voltage applied across inputs A and B. The shape of this normalised output voltage has very low 3^rdorder harmonic content.

I & Q Signal Sampling

Both the load voltage and the load current will be substantially sinusoidal waveforms, although they may be out of phase, depending on the impedance of the load represented by the transducer 8 and/or the vessel/tissue gripped by the forceps 9. The load current and the load voltage will be at the same drive frequency (f_d) corresponding to the presently defined waveform period (Δt_new). Normally, when sampling a signal, the sampling circuitry operates asynchronously with respect to the frequency of the signal that is being sampled. However, as the microprocessor 23 knows the frequency and phase of the switching signals, the synchronous sampling circuit 81 can sample the measured voltage/current signal at predefined points in time during the drive period. In this embodiment, during the ultrasonic mode of operation, the synchronous sampling circuit 81 oversamples the measured signal eight times per period to obtain four I samples and four Q samples. Oversampling allows for a reduction of errors caused by harmonic distortion and therefore allows for the more accurate determination of the measured current and voltage values. However, oversampling is not essential and indeed under sampling (less than two samples per period) is performed when the device is operating in the electrical mode of operation and is possible due to the synchronous nature of the sampling operation. The timing that the synchronous sampling circuit 81 makes these samples is controlled, in this embodiment, by the control signals CTRL₁and CTRL₂. Thus when the period of these control signals is changed, the period of the sampling control signals CTRL₁and CTRL₂also changes (whilst their relative phases stay the same). In this way, the sampling circuitry 81 continuously changes the timing at which it samples the sensed voltage and current signals as the period of the drive waveform is changed so that the samples are always taken at the same time points within the period of the drive waveform. Therefore, the sampling circuit 81 is performing a “synchronous” sampling operation instead of a more conventional sampling operation that just samples the input signal at a fixed sampling rate defined by a fixed sampling clock. Of course, such a conventional sampling operation could be used instead.

Measurements

The samples obtained by the synchronous sampling circuitry 51 are passed to the measured voltage and current processing module 83 which can determine the magnitude and phase of the measured signal from just one “I” sample and one “Q” sample of the load current and load voltage. However, in this embodiment, to achieve some averaging, the processing module 83 averages consecutive “I” samples to provide an average “I” value and consecutive “Q” samples to provide an average “Q” value; and then uses the average I and Q values to determine the magnitude and phase of the measured signal. Of course, it should be recognised that some pre-processing of the data may be required to convert the actual measured I and Q samples into I and Q samples of the load voltage or the load current, for example, scaling, integration or differentiation of the sample values may be performed to convert the sampled values into true samples of the load voltage (V_L) and the load current (i_L). Where integration or differentiation is required, this can be achieved simply by swapping the order of the I and Q samples—as integrating/differentiating a sinusoidal signal simply involves a scaling and a 90 degree phase shift.
The RMS load voltage, the RMS load current and the delivered power, P_delivered, can then be determined from:
$V_{RMS} = \frac{1}{\sqrt{2}} \sqrt{(V_{I}^{2} + V_{Q}^{2})}$ $I_{RMS} = \frac{1}{\sqrt{2}} \sqrt{(I_{I}^{2} + I_{Q}^{2})}$ $Power = V \cdot I^{*} = \frac{1}{\sqrt{2}} (V_{I} + j V_{Q}) (I_{I} - {jI}_{Q}) = P_{delivered} + j P_{reactive}$ $P_{delivered} = \frac{1}{\sqrt{2}} (V_{I} I_{I} + V_{Q} I_{Q})$ $P_{reactive} = \frac{1}{\sqrt{2}} (V_{Q} I_{I} - V_{I} I_{Q})$ $\langle Power \rangle = V_{RMS} I_{RMS} = \langle P_{delivered} + j P_{reactive} \rangle$
The impedance of the load represented by the ultrasonic transducer 8 and the vessel/tissue gripped by the forceps 9 (or just the impedance of the forceps 9 and the vessel/tissue gripped by the forceps 9 if the electrical energy is directly applied to the forceps 9) can be determined from:
$Z_{Load} = \frac{(V_{I} + j V_{Q})}{(I_{I} + {jI}_{Q})} = \frac{(V_{I} + {jV}_{Q}) (I_{I} - {jI}_{Q})}{(I_{I} + {jI}_{Q}) (I_{I} - {jI}_{Q})} = \frac{(V_{I} I_{I} + V_{Q} I_{Q} + {jV}_{Q} I_{I} - {jjV}_{I} I_{Q})}{\sqrt{2} I_{RMS}^{2}} = R_{Load} + {jX}_{Load}$
An alternative way of computing R_Loadand X_Loadis as follows:
$R_{Load} = \frac{P_{delivered}}{\sqrt{2} I_{R M S}^{2}} X_{Load} = \frac{P_{reactive}}{\sqrt{2} I_{RMS}^{2}}$
and the phase difference between the load voltage and the load current can be determined from:
Phase_measured =a tan 2(P _reactive ,P _delivered)
A computationally efficient, approximation to the a tan 2 function can be made using look up tables and interpolation in fixed point arithmetic, or using a ‘CORDIC’ like algorithm,

Limits

As with any system, there are certain limits that can be placed on the power, current and voltage that can be delivered either to the ultrasonic transducer 8 or to the forceps 9. The limits used in this embodiment and how they are controlled will now be described.
In this embodiment, the drive circuitry 29 is designed to deliver ultrasonic energy into tissue or to deliver electrical energy into tissue with the following requirements:

1) Supplied with a nominally 14V DC supply
2) Substantially sinusoidal output waveform at approximately 50 kHz in the case of ultrasonic operation
3) Substantially sinusoidal output waveform at approximately 500 kHz in the case of RF electrical operation
4) Power limited output of 90 W in the case of ultrasonic operation
5) Power limited output of 100 W in the case of electrical operation
6) Current limited to 1.4A_rmsand voltage limited to 130V_rmsin the case of ultrasonic operation
7) Current limited to 1.4A_rmsand voltage limited to 100V_rmsin the case of electrical operation
8) In the case of ultrasonic operation, the measured phase is greater than a system defined phase limit

The power controller 85 maintains data defining these limits and uses them to control the decision about whether to increase or decrease the waveform period or whether to skip pulses of the control signals given the latest measured power, load impedance and/or measured phase. In this embodiment, when operating in the ultrasonic mode of operation, the phase limit that is used depends on the measured load impedance. In particular, the power controller 85 maintains a look up table (not shown) relating load impedance to the phase limit; and the values in this table limit the phase so that when the measured load impedance is low (indicating that the jaws of the forceps 9 are open and not gripping tissue or a vessel), the delivered power is reduced (preferably to zero).
As discussed above, one of the ways to control the operation of the device (when operating in the electrical mode of operation) is to maximise the drive efficiency. When controlling the device in this way, the power controller 85 tracks a maximum power delivery condition as the load changes. The way that this can be done will now be described.

Maximum Power Delivery Tracking Condition

The complex impedance of the circuitry shown in FIG. 3 (when operating in the electrical mode of operation and with inductor 48 removed) can be approximated by the following equation:
$Z = j 2 π {fL}_{S}^{F} + \frac{1}{J 2 π {fC}_{S}^{F}} + \frac{j 2 π {fL}_{M}^{F} R_{load_ref}}{j 2 π {fL}_{M}^{F} + R_{load_ref}} + R_{s}$

Where:

R_{load_ref}is the load resistance referred to the primary (by the square of the turns ratio); and R_srepresents the equivalent series resistance of the inductor, transformer capacitor and switching devices. This complex impedance may be rewritten as:
$Z = j 2 π {fL}_{S}^{F} + \frac{1}{J 2 π {fC}_{S}^{F}} + \frac{4 π^{2} f^{2} L_{M F}^{^{2}} R_{load_ref}}{4 π^{2} f^{2} L_{M F}^{^{2}} + R_{load_ref}^{2}} \frac{j 2 π {fL}_{M}^{F} R_{load_ref}^{2}}{4 π^{2} f^{2} L_{M F}^{^{2}} + R_{load_ref}^{2}} + R_{s}$
Therefore, the real part of this complex impedance is:
$(Z) = \frac{4 π^{2} f^{2} L_{M}^{F 2} R_{load_ref}}{4 π^{2} f^{2} L_{M}^{F 2} + R_{load_ref}^{2}} + R_{s}$
And the imaginary part of this complex impedance is:
$(Z) = 2 π {fL}_{S}^{F} - \frac{1}{2 π {fC}_{S}^{F}} + \frac{2 π {fL}_{M}^{F} R_{load_ref}^{2}}{4 π^{2} f^{2} L_{M}^{F 2} + R_{load_ref}^{2}}$
When the drive frequency (t) corresponds to the resonant frequency of this complex impedance, the imaginary part
(Z)=0. Therefore, the power controller 85 can vary the drive frequency (t) to keep the imaginary part
(Z) at or around zero using a phase locked loop. Indeed, it can be shown that when
(Z)=0 the maximum power (for a given supply voltage) is delivered to the load.
FIG. 7 is a contour plot showing the power contours than can be delivered to the load versus the drive frequency and the load resistance (Rload). As shown in FIG. 7, the power that can be delivered varies with the load resistance and the drive frequency. FIG. 7 also shows the line 92 of maximum power delivery that can be achieved as the load resistance and drive frequency change. Therefore, the power controller 85 can use the measured value of Rload together with stored data defining the line 92 shown in FIG. 7 (which may be a look-up-table) to determine the corresponding drive frequency to be used. In this way, the microprocessor 23 will track along the line 92 shown in FIG. 7 as the load resistance changes during the cutting/cauterisation process.
One of the advantages of this approach is that it enables a useful operating condition at low values of Rload, in particular for values of Rload less than the critical value (i.e. when in which a maximum power will be delivered even if this is below the R_{load_ref}<2π∫L^F _M), desired power level. However, operating along the line 92 of maximum power delivery can result in some of the above system limits being breached unless further control action is taken. In the preferred embodiment, this further control action is to use pulse skipping techniques until the average power, current and/or voltage is below the relevant limit. For example, as can be seen from the measurements described above, the measured voltage and current processing module 83 can determine the delivered power, the RMS voltage and the RMS current. The power controller 85 can therefore use these values to skip one or more pulses of the CTRL control signals until the measured voltage and current values are below the relevant system limits and the delivered power is at or below the power set-point defined by the medical device control module 89.
Pulses may be skipped, for example, by passing the pulses generated by the control signal generator 87 through a logic gate (not shown) and selectively suppressing pulses that are generated by the control signal generator 87 by controlling the logic level of another input to the logic gate. For example, the pulses of each control signal that is generated by the control signal generator 87 may be passed through an AND gate, with another input of the AND gate being generated by the power controller 85 and being a logic “1” when the pulses are to be output to the FET gate drives 37 as normal and being a logic “0” when the pulses are to be skipped or suppressed. Other pulse skipping techniques could of course be used.

Medical Device Control Module

As mentioned above, the medical device control module 89 controls the general operation of the cauterisation/cutting device 1. It receives user inputs via the user input module 91. These inputs may specify that the jaws of the forceps 9 are now gripping a vessel or tissue and that the user wishes to begin cutting/cauterisation and specify whether ultrasonic energy or electrical energy is to be applied to the vessel/tissue. In response, in this embodiment, the medical device control module 89 initiates a cutting/cauterisation control procedure. Initially, the medical device control module 89 sends an initiation signal to the power controller 85 and obtains the load impedance measurements determined by the measured voltage and current processing module 83. The medical device control module 89 then checks the obtained load impedance to make sure that the load is not open circuit or short circuit. If it is not, then the medical device control module 89 starts to vary the power set point to perform the desired cutting/cauterisation and sets the initial period/frequency of the drive signal to be generated. As discussed above, for ultrasonic operation, the initial frequency of the drive signal will be set around 50 kHz and for RF electrical operation, the initial frequency will be set around 500 kHz.

Voltage/Current Sensing Circuitry

As shown in FIG. 2, voltage sensing circuitry 31 is provided to sense the load voltage applied to the load and current sensing circuitry 33 is provided to sense the current applied to the load. The sensed signals are supplied to the microprocessor 23 for use in controlling the operation of the medical device. There are various ways of sensing the load voltage and the load current and some of these will now be described.
FIG. 8a illustrates the primary side of the first transformer circuit 42-1 and one way in which the current sensing circuitry 33 obtains a measure of the load current. As shown, the current sensing circuitry 33 comprises an additional inductor turn(s) 67 which link the flux present in inductor 47 (or inductor 49) and that consequently outputs a voltage across inductor 67 that varies with the rate of change of load current. The voltage across the inductor 67 is a bipolar voltage whose amplitude is directly proportional to the rate of change of load current and the number of turns in 67. This bipolar voltage is scaled and converted into a unipolar voltage suitable for input to the microprocessor 23 by the op-amp circuit 69-1 which outputs a measured voltage (V^meas). This measured voltage will also depend on the current flowing in the inductor 47 and so will also depend on the current flowing on the secondary side of the transformer circuit 42-1 and thus the current flowing through the load. As the ratio of the number of turns of inductor 47 to inductor 67 is known, the measured voltage and current processing module 83 can use V^measto determine the voltage across inductor 47. The voltage across inductor 47 is related to the current flowing through the inductor 47 by V=Ldi/dt. As the inductance of inductor 47 is known, the measured voltage and current processing module 83 can determine the current flowing in the primary side of the transformer circuit 42-1 by integrating the voltage across the inductor 47 and by scaling the result to account for the inductance of the inductor 47 (and the scaling of the op-amp-circuit 69-1). This current measure can then be converted into a suitable measure of the load current (i_L) by a further scaling to take into account the number of turns between inductor 49 and inductor 53. Of course, the measured voltage and current processing module 83 does not need to integrate the voltage across the inductor 47—as the measured signals are sinusoidal and so integration can be achieved by applying a suitable scaling factor and a 90 degree phase shift. Thus, the measured voltage and current processing module 83 can determine the load current by applying a suitable (pre-stored) scale factor to the measured voltage (V^meas) and by applying a suitable 90 degree phase shift (which can be achieved simply by swapping the order of the I and Q samples as discussed above).
FIG. 8b illustrates the primary side of the first transformer circuit 42-1 and another way in which the current sensing circuitry 33 can obtain a measure of the load current. As shown, in this case, the current sensing circuitry 33 measures the voltage across the capacitor 45. The voltage across the capacitor 45 is a bipolar voltage. This bipolar voltage is scaled and converted into a unipolar voltage suitable for input to the microprocessor 23 by the op-amp circuit 69-2 which outputs a measured voltage (V^meas). This voltage is related to the current flowing in the primary side of the transformer circuit 42 by I=CdV^meas/dt and thus the current flowing through the load. As the capacitance of the capacitor 45 is known, the measured voltage and current processing module 83 can determine the current flowing in the primary side of the transformer circuit 42-1 by differentiating the voltage across the capacitor 45 and by scaling the result to account for the capacitance of the capacitor 45 (and the scaling of the op-amp-circuit 69-2). This current measure can then be converted into a suitable measure of the load current (i_L) by a further scaling to take into account the number of turns between inductor 49 and inductor 53. Of course, the measured voltage and current processing module 83 does not need to differentiate the voltage across the capacitor 45—as the measured signals are sinusoidal and so differentiation can be achieved by applying a suitable scaling factor and a 90 degree phase shift. Thus, the measured voltage and current processing module 83 can determine the load current by applying a suitable (pre-stored) scale factor to the measured voltage (V^meas) and by applying a suitable 90 degree phase shift (which can be achieved simply by swapping the order of the I and Q samples as discussed above).
FIG. 8c schematically illustrates how a measure of the load voltage can be determined. FIG. 8c shows the secondary side of the first transformer circuit 42 and illustrates the use of a voltage divider circuit (in this case formed by resistors R1 and R2), with the voltage across resistor R2 being input to the op-amp circuit 69-3. Therefore, by applying an appropriate scale on the measured voltage from op-amp 69-3, the measured voltage and current processing module 83 can determine the load voltage.
The sensing circuits described above use op-amp circuits 69 to convert the bipolar drive signals into unipolar voltages that are suitable for input to the microprocessor 23. The use of such op-amp circuits has a number of disadvantages, including that they are costly, they consume power and they requires space within the electronics. These are important factors when the device is designed to be battery powered and the electronics are housed within the handle 5 of the device. FIG. 9 illustrates various sensing circuits that can be used without an op-amp. The circuit of FIG. 9a is suitable when the AC drive signal is unipolar. This may be achieved by replacing the full bridge signal generator 27 with a half bridge signal generator. This would mean removing, for example, the FETs 41-3 and 41-4 and connecting node B to ground. In this case, the microprocessor 23 only needs to generate one control signal (CTRL₁) to control FETs 41-1 and 41-2. The circuit of FIG. 9b is suitable for both unipolar and bipolar drive signals.
In FIG. 9a , the capacitor 45 has been moved to be between the inductor 49 that forms the primary of the transformer circuit 42-1 and ground (GND). The current sensing circuitry 33 is arranged to measure the voltage across the capacitor 45 via a potential divider formed by resistors R1//R2 and R3. The resistor R1 connects the output of a DC blocking capacitor C_Bto a supply voltage rail of the microprocessor 23 (in this case at 3.3V) through the switch 121; and resistor R2 connects the output of the DC blocking capacitor C_Bto a reference potential (in this case ground). The resistors R1 and R2 therefore provide a divider circuit that applies a DC bias to the measured AC signal. The DC blocking capacitor prevents this DC bias from coupling to the drive circuit. Typically, the resistors R1 and R2 are equal so that the DC bias will be at 1.65V. Thus, the voltage output from the sensing circuit 33 will be an AC voltage having a mid-rail value of about 1.65 V and whose peak voltage will be a proportion of the voltage across the capacitor 45. The potential divider formed by resistors R1//R2 and R3 is such that the peak to peak amplitude of the AC signal passed to the microprocessor 23 is less than the 3.3V input range of the microprocessor 23. If the microprocessor 23 operates at a different voltage rail (for example 5V), then the values of the resistors R1, R2 and R3 can be adjusted accordingly. To minimise current drawn either from the 3.3V rail or from the transformer circuit 42-1, the resistors R1, R2 and R3 can have relatively large values and typical values for these resistors are: R1=R2=200Ω and R3=1000Ω. The values of R1 and R2 should be chosen to meet the input impedance requirements of the Analogue to digital converter used to sample the signals. The switch 121 allows the microprocessor 23 to disconnect the sensing circuit 33 from the 3.3V rail—so that when sensing is not required, the circuit 33 does not consume any power.
FIG. 9b illustrates the way in which similar circuits can be provided on the secondary side of the transformer circuit 42-1. In particular, FIG. 9b shows the voltage sensing circuit 31 used to obtain a measure of the load voltage V_Lvia a voltage divider formed by capacitors C1 and C2. If the capacitors C1 and C2 were replaced with resistors, then a blocking capacitor C_Bshould be provided before the divider circuit formed by resistors R1 and R2 in the manner shown in FIG. 9a . FIG. 9b also shows the current sensing circuit 33, which senses the load current by sensing the voltage across capacitor C3 via a potential divider formed by resistors R4//R5 and R6. As shown, a DC blocking capacitor C_Bis provided to enable a DC bias signal to be added to the output to the microprocessor via the divider circuit formed from resistors R4 and R5. As before, the switches 121-1 and 121-2 allow the microprocessor to switch off the sensing circuits 31 and 33 when sensor signals are not required. As those skilled in the art will appreciate, the “op-ampless” sensing circuits 31 and 33 illustrated in FIG. 9 are cheaper to manufacture, can be made to consume less power and take up less space on the circuit board than the op-amp circuits illustrated in FIG. 8.
The circuits illustrated in FIGS. 8 and 9 were used to obtain measurements from the first transformer circuit 42-1 of the drive circuit 29. As those skilled in the art will appreciate, the same or similar sensing circuits would be provided for sensing signals in the second transformer circuit 42-2. Further, although the sensing circuits illustrated in FIG. 9 sense the voltage across a capacitor, the circuits could also sense the voltage across another impedance element, such as across a resistor of the transformer circuit 42.

Active Battery Protection

Typically, with battery operated devices such as the medical device 1 described above, a fuse is provided between the batteries and the electric circuitry, to protect the battery from damage caused by short circuits and the like. However, standard fuses have a resistance of about 10 mOhms. With such a standard fuse, when 10A is drawn from the batteries, approximately 1 W is dissipated through the fuse. An active fuse circuit 130 that is used in this embodiment will now be described with reference to FIG. 10 that reduces the power dissipation associated with such standard fuses.
FIG. 10 shows the batteries 15, which supply the 14V rail and the GND rail to the circuitry shown in FIG. 3. The active fuse circuit 130 comprises a differential amplifier 131 that measures the voltage drop across part of a PCB conductor trace 133 that is connected between the 14V rail and the positive terminal of the batteries (V_bat+). The conductor trace 133 has a resistance of approximately 1-2 mOhms and so the voltage drop is proportional to the current that is being drawn from the batteries 15. The measured voltage drop is low pass filtered by the low pass filter 135 to avoid transient spikes triggering the fuse circuit—so that it is just the voltage corresponding to the DC current drawn from the batteries 15 that will pass through the filter 135. In this embodiment, the low pass filter 135 has a cut-off frequency of about 10 Hz. The output from the low pass filter 135 is then compared with a reference voltage (V_ref) using a latching comparator 137. The reference voltage is set in advance to correspond to a desired limit on the current drawn from the batteries. In this embodiment, V_refis set to correspond to a current limit of 15A. When the voltage drop across the trace 133 is less than the reference voltage the output of the comparator 137 remains at a high value—which maintains the FET switch 139 on and so current can be drawn from the batteries 15 by the bridge signal generator 27. However, when the voltage drop across the trace 133 is greater than the reference voltage, then the output of the comparator 137 goes low and it is maintained low even if the current drawn from the batteries drops below the defined limit. When the comparator output is low the FET 139 is switched off, thereby disconnecting the batteries 15 from at least the bridge signal generator 27.
In this embodiment, the FET 139 is an N-channel enhanced mode switch having an on resistance of just 2 mOhms. This means that when the switch 139 is switched on and 10A is being drawn from the batteries, just 0.2 W is dissipated through the switch 139.
In this embodiment, when the comparator 137 is triggered and the switch 139 is switched off (open circuit), the batteries 15 have to be removed to reset the active fuse circuitry 130—as the opening of the switch 139 disconnects all the electronics except the circuit components of the active fuse 130 from the batteries. Alternatively, if the microprocessor 23 (or some other control circuitry) is powered directly from the batteries, then the comparator 137 could be reset either in response to a user input (for example in response to the user pressing a reset button or the like) or in response to some other trigger event (such as after a predetermined time-out period).

Modifications and Alternatives

A medical cauterisation/cutting device has been described above. As those skilled in the art will appreciate, various modifications can be made and some of these will now be described. Other modifications will be apparent to those skilled in the art.
In the above embodiment, various operating frequencies, currents, voltages etc were described. As those skilled in the art will appreciate, the exact currents, voltages, frequencies, capacitor values, inductor values etc. can all be varied depending on the application and any values described above should not be considered as limiting in any way. However, in general terms, the circuit described above has been designed to provide a drive signal to a medical device, where the delivered power is desired to be at least 10 W and preferably between 10 W and 200 W; the delivered voltage is desired to be at least 20 V_Rmsand preferably between 30 V_Rmsand 120 V_Rms; the delivered current is designed to be at least 0.5 A_RMSand preferably between 1 A_RMSand 2 A_RMS; and the drive frequency for ultrasonic operation is desired to be at least 20 kHz and preferably between 30 kHz and 80 kHz; and the drive frequency for RF operation is desired to be at least 100 kHz and preferably between 250 kHz and 1 MHz.
In the above embodiment, the resonant circuits 43-1 and 43-2 were formed from capacitor-inductor-inductor elements. As those skilled in the art will appreciate, other resonant circuit designs with multiple capacitors and inductors in various series and parallel configurations or simpler LC resonant circuits may also be used. Also, in some applications there is no need for a transformer to step-up the drive voltage, as the FETs can deliver the required drive voltage.
FIG. 1 illustrates one way in which the batteries and the control electronics can be mounted within the handle of the medical device. As those skilled in the art will appreciate, the form factor of the handle may take many different designs. Indeed, it is not essential for the device to be battery powered, although this is preferred for some applications to avoid the need for power cords and the like.
The embodiment described above included a description of various novel features, including the novel ability to selectively apply ultrasonic energy or RF energy to the tissue gripped by the forceps, the novel way in which the microprocessor controlled the operation of the device in the electrical mode of operation; the way in which load current/voltage is measured and the way in which the batteries are protected using an active fuse circuit. As those skilled in the art will appreciate, these novel features do not need to be employed together. For example, the current/voltage sensing techniques described above can be used with other devices as can the active fuse circuit. Similarly, the way in which the electrical mode of operation is controlled by tracking the maximum power delivery condition and by using pulse skipping techniques can be used in a device that does not have an ultrasonic transducer.
In the above embodiment, an exemplary control algorithm for performing the cutting/cauterisation of the vessel or tissue gripped by the forceps was described. As those skilled in the art will appreciate, various different procedures may be used and the reader is referred to the literature describing the operation of such cutting/cauterisation devices for further details.
In the above embodiment, four FET switches were used to convert the DC voltage provided by the batteries into an alternating signal at the desired frequency. As those skilled in the art will appreciate, it is not necessary to use four switches—two switches may be used instead (using a half bridge circuit). Additionally, although FET switches were used, other switching devices, such as bipolar transistor switches may be used instead. However, MOSFETs are preferred due to their superior performance in terms of low losses when operating at the above described frequencies and current levels.
In the above embodiment, the I & Q sampling circuitry 81 oversampled the sensed voltage/current signal in the ultrasonic mode of operation and undersampled the sensed voltage/current signal in the electrical mode of operation. As those skilled in the art will appreciate, this is not essential. Because of the synchronous nature of the sampling, samples may be taken more than once per period or once every n^thperiod if desired. The sampling rate used in the above embodiment was chosen to maximise the rate at which measurements were made available to the power controller 85 and the medical device control module 89 as this allows for better control of the applied power during the cauterisation process.
In the above embodiment, a 14V DC supply was provided. In other embodiments, lower (or higher) DC voltage sources may be provided. In this case, a larger (or smaller) transformer turns ratio may be provided to increase the load voltage to the desired level or lower operating voltages may be used.
In the above embodiment, the medical device was arranged to deliver a desired power (in the form of ultrasonic energy or electrical energy) to the tissue/vessel gripped by the forceps. In an alternative embodiment, the device may be arranged to deliver a desired current or a desired voltage level to the ultrasonic transducer or the forceps.
In the above embodiment the battery is shown integral to the medical device. In an alternative embodiment the battery may be packaged so as to clip on a belt on the surgeon or simply be placed on the Mayo stand. In this embodiment a relatively small two conductor cable would connect the battery pack to the medical device.
In the above embodiment, a microprocessor based control circuitry was provided. This is preferred due to the ease with which the microprocessor can be programmed to perform the above control actions using appropriate computer software. Such software can be provided on a tangible carrier, such as a CD-ROM or the like. Alternatively, hardware control circuitry can be used in place of the microprocessor based circuitry described above.
In the above embodiment, the user controlled whether the energy delivered to the vessel/tissue was ultrasonic energy or RF electrical energy. In alternative embodiments, the microprocessor may control the selection based on an internally generated control signal or in response to a control signal received from another device.
In the above embodiment, the active fuse circuit opened a switch that disconnected the bridge signal generator from the batteries. In an alternative embodiment, the active fuse circuit could instead switch in a large impedance between the batteries and the bridge signal generator to limit the current drawn from the batteries. Also, the switch could be used to disconnect the positive voltage supply from the signal generator instead of disconnecting the negative terminal of the batteries.

Claims

1.-17. (canceled)

18. A medical device comprising:

an end effector for gripping a vessel/tissue;

a drive circuit for generating a cyclically varying drive signal for driving energy into the vessel/tissue;

sensing circuitry for sensing a drive signal generated by the drive circuit; and

a controller responsive to the sensing circuitry and operable to control the drive circuit to control the energy delivered to the vessel/tissue;

wherein the drive circuit comprises an impedance element that is coupled to a reference potential and wherein the sensing circuitry comprises a divider circuit for obtaining a measure of the voltage across the impedance element and a bias signal generator for applying a DC bias signal to the voltage measure.

19. A device according to claim 18, wherein the impedance element comprises a capacitor or a resistor.

20. A device according to claim 18, wherein the sensing circuitry comprises DC blocking circuitry for preventing the DC bias signal from the bias signal generator from coupling with the drive circuit.

21. A device according to claim 18, wherein the bias signal generator comprises a divider circuit connected between a reference voltage and a supply voltage of the controller.

22. A device according to claim 21, wherein the divider circuit of the bias signal generator is connected to the supply voltage of the controller via a switch and wherein the controller is configured to open the switch when the controller does not require signals from the sensing circuitry.

23.-39. (canceled)