EP1278295A2 - Steuerung der gespeicherten magnetischen Energie in Stromwandlertransformatoren - Google Patents

Steuerung der gespeicherten magnetischen Energie in Stromwandlertransformatoren Download PDF

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Publication number
EP1278295A2
EP1278295A2 EP02078757A EP02078757A EP1278295A2 EP 1278295 A2 EP1278295 A2 EP 1278295A2 EP 02078757 A EP02078757 A EP 02078757A EP 02078757 A EP02078757 A EP 02078757A EP 1278295 A2 EP1278295 A2 EP 1278295A2
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EP
European Patent Office
Prior art keywords
transformer
reset
capacitor
switch
current
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Withdrawn
Application number
EP02078757A
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English (en)
French (fr)
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EP1278295A3 (de
Inventor
Patrizio Vinciarelli
Jay Prager
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VLT Corp
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VLT Corp
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Publication of EP1278295A3 publication Critical patent/EP1278295A3/de
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H02M1/34—Snubber circuits
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H02M1/34—Snubber circuits
    • H02M1/346—Passive non-dissipative snubbers
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • This invention relates to control of stored magnetic energy in power converter transformers.
  • a primary switching element 20 is used to repetitively connect an input source to the primary winding 329 of a power transformer 25 ( Figure 1) during a portion of each of a series of converter operating cycles.
  • a primary switching element 20 forward energy transfer from the input source toward the load occurs during the time that the switch is closed.
  • single-ended flyback converters energy is stored in the transformer during the time that the switch is closed and is transferred forward to the load after the switch is opened.
  • single-ended converters a portion of the energy may be transferred during both the on and off times of the switch.
  • circuitry for managing the effects of magnetic energy storage in the transformer.
  • circuitry is typically included to deal with the magnetizing energy which is stored in the transformer at the time that the switch is opened (e.g., to provide means for "resetting” the transformer); more generally, the effects of magnetic energy storage in the leakage inductance of the transformer must also be managed, e.g., to prevent the energy stored in the leakage field from overstressing the switch when the switch is turned off.
  • reset circuits also called core reset circuits because almost all switching power converter transformers include a magnetic core
  • These circuits differ in terms of their efficiency, cost and impact on converter power density.
  • One such circuit 15, shown in Figure 2 is described in Clemente, et al., "A Universal 100 KHz Power Supply Using a Single HEXFET,” International Rectifier Corporation Applications Note AN-939, December 1980.
  • One of the advantages of this circuit is that it can maintain the minimal value of peak switch voltage consistent with converter input voltage and duty cycle; one of its disadvantages is that it is dissipative and therefore compromises both converter efficiency and power density.
  • FIG. 3 Another circuit 30, of the kind described in Vinciarelli, "Optimal Resetting of the Transformer's Core in Single Ended Forward Converters," U. S. Patent No. 4,441,146, (incorporated by reference) is shown in Figure 3.
  • This circuit has several advantages, among which are the following: (1) it, too, minimizes the peak voltage seen by the switch, (2) it provides for bipolar transformer 25 core excitation, thereby maximizing utilization of the available flux swing in the transformer's core and allowing for a smaller transformer, and (3) it is essentially lossless.
  • circuits 15, 30 of the kinds shown in Figures 2 and 3 When used in single-ended forward converters, core reset circuits 15, 30 of the kinds shown in Figures 2 and 3 will also inherently control the effects of leakage energy by providing a capacitive sink which "snubs" (i.e., clamps) the effect of the release of the leakage energy on switch voltage.
  • circuits 15, 30 of the kinds shown in Figures 2 and 3 may be used solely as snubber circuits to manage and control the effects of stored leakage energy.
  • flyback converters all of the magnetizing energy stored in the transformer 25 during the time that the primary switch 20 is closed is magnetically coupled into the transformer secondary winding and released to the load after the switch opens.
  • Circuits 15, 30 of the kinds shown in Figures 2 and 3 can be modified to correspond to a single-ended flyback topology by simply reversing the polarity of one of the transformer 25 windings) can be used to clamp the voltage which might otherwise be caused by the stored leakage energy.
  • the invention features apparatus (and a method) useful in a switching power converter having a transformer and a primary switch for connecting a DC input source to a primary winding of the transformer during a portion of each of a succession of converter operating cycles, the apparatus includes a reset capacitor and a reset switch.
  • Reset circuitry cooperates with the reset switch to connect and disconnect the reset capacitor in a manner which provides for resetting the transformer and which allows a current having a non-zero average value to flow in the reset switch.
  • Embodiments may include one or more of the following features.
  • the reset circuitry may open and close the reset switch.
  • the reset circuitry may cause the connecting and disconnecting to occur at times based on switching of the main switch.
  • the reset switch may be connected in series with the capacitor.
  • the reset circuitry may include circuit elements which inhibit bidirectional energy flow between the capacitor and the transformer.
  • the apparatus may only allow transfer of magnetizing energy between the reset capacitor and the transformer if the polarity of the voltage across the reset capacitor is of a particular polarity.
  • the apparatus may be connected in parallel with a winding on the transformer.
  • the winding may be the primary winding, a secondary winding, or an auxiliary winding.
  • the reset switch may be a unidirectional switch (e.g., a MOSFET) in parallel with a unidirectional conducting element (e.g., the intrinsic body diode of the MOSFET), the unidirectional switch and the unidirectional conducting element being arranged to conduct in opposite directions.
  • the unidirectional switch may be a MOSFET in series with a series diode, the series diode and the MOSFET being poled to conduct in the same direction.
  • the circuit elements may include a reset diode connected in parallel with the capacitor. Bidirectional energy flow may be allowed only if the voltage across the reset capacitor is of a particular polarity.
  • the particular polarity may be that which will result in a reversal in the polarity of transformer magnetizing current during the time that the reset switch is closed.
  • the reset circuitry may include circuit elements which inhibit bidirectional energy flow between the capacitor and the transformer.
  • the circuit elements may include a reset diode connected across a series circuit comprising the reset capacitor and the series diode.
  • the switching power converter may be a forward power converter, a zero-current switching converter, or a PWM converter.
  • the reset circuitry may open the reset switch prior to the ON period of the primary switch, may close the reset switch during the OFF period of the primary switch, and may keep the reset switch open throughout the ON period of the primary switch.
  • the invention features a method for limiting the slew rate in a switching power converter which includes a transformer and a reset circuit of the kind which non-dissipatively recycles the magnetizing energy stored in a transformer during each of a succession of converter operating cycles.
  • the method includes sensing the magnetizing current which is flowing in said transformer, and allowing initiation of another converter operating cycle only if the magnetizing current meets a predefined criterion for safe converter operation.
  • the predefined criterion may be the magnetizing current passing through a predefined value, e.g., zero.
  • Resonance between the transformer and the reset capacitor can be inhibited. Very low dissipation may be achieved.
  • the selection of Cr can be made based solely on the operating requirements of the converter.
  • oscillation in the reset circuit 30 may generally result in undesirable coupling of energy into the converter output in the form of noise at the frequency fr, or, as seen in Figures 5B to 5D, at a higher frequency associated with resonant ringing of the transformer leakage inductance (Ll, 123, Fig. 6) with both the reset capacitor 34 and the secondary capacitor 60 during the time period that the voltage Vc is positive.
  • FIG. 7A Another characteristic of the circuit of Figure 3 is that the maximum allowable converter slew rate (e.g., the rate-of-change of duty cycle), and hence converter response time, are reduced as the value of Cr is increased.
  • the damped reset circuit 230 is identical to the circuit of Figure 3 except that a passive damper, consisting of a damper resistor 39 and damper capacitor 35, has been connected across the reset capacitor 34.
  • the damper resistor will dissipate energy at frequencies for which the impedance of Cd is low compared to Rd and for which the impedance of Cr is not low relative to Rd. Thus, at both very low and very high frequencies the damper creates little loss.
  • waveforms are shown for the converter of Figure 6 with the undamped reset circuit 130 replaced with the damped reset circuit 230 of Figure 8.
  • fr is chosen to be well below (e.g., approximately 10% of) the maximum operating frequency of the converter since this puts the "normal" operating frequency range of the converter well above fr and provides for minimum damper dissipation in this range.
  • the characteristic frequency, fr, of a ZCS converter operating at frequencies up to 1 MHz would be set to be near 100 KHz. This provides effective damping with relatively little power loss at converter operating frequencies well above fr.
  • Figure 10 shows a single-ended converter which includes an improved reset circuit 330.
  • the reset switch 332 is opened when the primary switch is closed and vice versa.
  • the improved reset circuit 330 differs from the circuits of Figures 3 and 8 by including a diode 350 connected across the reset capacitor 334. By poling the diode as shown, so that the reset capacitor can support positive values of the voltage Vc but cannot support negative values, the diode effectively prevents resonance (e.g., bidirectional energy transfer) from occurring between the reset capacitor 334 and the magnetizing inductance of the transformer 25.
  • FIG. 11B show sustained oscillations, either at the characteristic frequency, fr, or at higher frequencies; the absence of the higher frequency oscillation is also an indicator that forward power flow from the reset circuit to the load has been eliminated.
  • the decay in the current I in Figure 11B is due to the nonideal characteristics (e.g., parasitic resistances or finite voltage drops) of the diode 350, switch 332, and transformer primary winding 329 -- if all circuit elements were ideal, the current I would remain unchanged until the next operating cycle.
  • the parasitic resistances of the reset switches 332, 32 is 0.1 ohm; the parasitic resistance of the transformer primary winding (e.g., primary winding 329 is 0.5 ohm; the diode 350, where used, is a silicon device).
  • the average value of the capacitor voltage, Vc, in the circuit of Figure 10 will be approximately equal to Vin*D.
  • the diode 350 will not become conductive and the waveforms for the reset circuit of Figure 10 will revert to those shown in Figure 4 for the prior art circuit of Figure 3.
  • Preset represents the total amount of power dissipated in the reset circuit (e.g., circuits 230, 330) and Pd represents the loss in the "resonance-control" components used in the particular circuit (e.g., damper resistor 39 in circuit 230; diode 350 in circuit 330).
  • loss in the passive damper circuit is strongly influenced by the values of Cr, Cd and Rd, and reductions in Cr result in lower values of Cd and increased amounts of Pd and Preset.
  • the values of both Preset and Pd in the improved reset circuit 330 are seen to be much lower than the corresponding losses in the passively damped circuit 230, and the value of loss is only weakly a function of the value of Cr.
  • a fundamental and important difference between the prior art reset circuit topology 30 of Figure 3 and the improved topology 330 of Figure 10 is that the average value of the current, I, in the prior art topology must be zero, whereas in the improved topology the average value of I can assume values which are nonzero and positive in the direction of the arrow. This is because all of the current, I, in the topology of Figure 3 must flow in the reset capacitor 34, and, as a result, the voltage Vc can only stabilize at an average value which is consistent with zero average flow of current I.
  • the diode 350 provides a path for supporting a positive average value of I. This is how resonance is prevented.
  • the improved reset circuit 330 may be placed across a transformer winding other than the primary winding 329.
  • the circuit 330 is placed across the secondary winding 331 which delivers power to the load; in Figure 13B it is placed across an auxiliary winding 341.
  • a benefit of "direct-coupled" reset e.g., as in Figure 10, where the reset circuit 330 is directly across the winding which is connected to the primary switch 20) is that the reset circuit 330 automatically acts as a snubber for the primary switch when the switch is turned off.
  • a benefit of the use of an auxiliary winding 341 (Fig.
  • a single reset circuit embodiment e.g., same value of reset capacitor, Cr; the same switch and diode
  • a single reset circuit embodiment may be used within converters having widely varying values of both input and output voltage by simply adjusting the relative number of turns on the auxiliary winding to provide a predetermined value of transformer volts-per-turn.
  • the reset circuit 330 is connected in other than a direct-coupled configuration, the ability of the circuit to snub the primary switch 20 will be diminished by the leakage inductances between the windings. In such cases care must be used to ensure that the coupling between windings is sufficiently high to prevent excessive voltage spikes on the switch 20.
  • the reset switch 332 may, as shown in Figure 14, be embodied as a unipolar switch 339 (the direction of conduction of the current Isw being indicated by the arrow) in parallel with a diode 337. If, as shown in Figure 15A, the switch 332 is embodied as a MOSFET switching device 333, then the intrinsic body diode 337a of the MOSFET device 333 may, under appropriate circumstances, be exploited for use as the diode 337. Suitability of this approach will depend upon the reverse recovery characteristics of the intrinsic body diode (which, for a MOSFET switch, is a function of voltage rating) and the minimum required on-time of the reset switch during the converter operating cycle.
  • the reverse recovery time of the intrinsic body diode of an International Rectifier IRFZ15 MOSFET, having a breakdown voltage rating of 60 Volts is 140 nanoseconds, maximum
  • the reverse recovery time of the intrinsic body diode of an International Rectifier IRF710 MOSFET, having a breakdown voltage rating of 400 Volts is 520 nanoseconds, maximum.
  • the relatively long recovery time of the intrinsic body diode in the cited higher voltage MOSFET would make it unsuitable for use as the diode 337 (Fig. 14) in applications in which the minimum on-time of the reset switch falls below 520 nanoseconds (e.g., in a ZCS converter with a maximum operating frequency around 1 MHz; in most PWM converters at high duty cycles).
  • FIG. 15B shows a reset switch 332 embodiment using a MOSFET switch 333 and an external diode 342.
  • a series diode 344 is put in series with the MOSFET switch 333.
  • FIGS 16A to 16C Several embodiments of an improved reset circuit 330a, 330b, 330c are shown in Figures 16A to 16C, respectively.
  • the improved reset circuit 330a incorporates the MOSFET switch 333 of Figure 15A into the circuit topology 330 of Figure 10; in Figure 16B the improved reset circuit 330b incorporates the MOSFET switch 333 and diodes 342, 344 of Figure 15B into the circuit topology 330 of Figure 10.
  • the circuit of Figure 16B is modified by connecting the cathode of the reset diode 350 directly to the MOSFET switch 333, rather than to the series diode 344 as is done in Figure 16B. At times at which current is flowing in the reset diode 350, the circuit of Figure 16C will exhibit lower power loss than that of Figure 16B, since the current flow in the circuit of Figure 16C will bypass the series diode 344.
  • the diode 344 (Fig. 15B, 16B, 16C) should be a device having a relatively low breakdown voltage rating (e.g., a Schottky diode having a low avalanche voltage rating, Vav).
  • a relatively low breakdown voltage rating e.g., a Schottky diode having a low avalanche voltage rating, Vav.
  • One reason for this is that diodes having lower breakdown voltages also have lower forward voltage drops and therefore will result in less circuit loss.
  • Figures 17A and 17B show converter circuit models 10a, 10b which include, respectively, reset circuits 330a, 330c of the kind shown in Figures 16A and 16C, and inductances 127 which emulate the magnetizing inductance of a transformer.
  • the drain-to-source capacitance, Cd 344, and the gate-to-drain capacitance, Cg 348, of the MOSFET 333 (the intrinsic body diode is not shown) and, in Figure 17B, the parasitic junction capacitance, Cj 346, of the external diode 342.
  • Vin the converter input voltage
  • Im magnetizing current
  • the diode 344 has a low breakdown voltage (e.g., 20 Volts) then the diode 344 will avalanche when Im begins to flow in the reset circuit 330c and the capacitances 344, 348 will be discharged until the voltage across the MOSFET (e.g., the voltage across capacitor 344) falls below the breakdown voltage, Vav, of the diode 344, at which point the diode will come out of avalanche and cease conducting. Since Vav is much lower than Vin +Vc the amount of switching loss is substantially reduced. The same reasoning applies to the circuit embodiment of Figure 16B, which will also benefit from use of a series diode 344 having a low breakdown voltage rating.
  • a low breakdown voltage e.g. 20 Volts
  • the breakdown voltage rating, Vav, of the series diode 344 should not be made too low, however.
  • Vs drain-source voltage
  • Cd 344 equivalent gate-to-drain capacitance
  • the flow of Ig can disrupt control circuitry 16 connected to the FET gate. By keeping Vav at a suitable value (e.g., 20 V) this problem can be avoided with minimal increase in switching losses.
  • a suitable value e.g. 20 V
  • the flow of Ig may be bypassed around the control circuitry 16 by adding a high-speed diode 347 (e.g., a Schottky diode) across the gate-to-source terminals of the MOSFET.
  • Controlling the reset switch 332 (Fig. 10) in the improved reset circuit 330 is essentially the same as controlling the switch 32 in the prior art converter of Fig. 3.
  • the reset switch 332 must be turned on after the primary switch 20 opens and must be off when the primary switch closes. If a reset switch of the kind shown in Figures 14 and 15 is used, the timing of the closure of the reset switch is not critical since the diode (337, Fig. 14) will begin to conduct magnetizing current Im as soon as the current commutates into the reset circuit 330.
  • closure of the reset switch must not be allowed to occur while the primary switch is conductive as this can cause catastrophic failure of the converter. Therefore it is beneficial to provide means for sensing that the primary switch 20 has opened and enable closure of the reset switch only when this signal is present.
  • Figures 18A and 18B Ways to sense the state of the primary switch are shown in Figures 18A and 18B.
  • a reset circuit of the kind shown in Figure 16B is shown connected to a transformer 25 and a primary switch 20.
  • a sense resistor 402 of a relatively small value Rs, is connected in series with the diode 342.
  • the primary switch 20 opens and magnetizing current begins flowing, as a negative flow of the current I, in the diode 342 the voltage, Vm, across the resistor will go negative and this may be used to sense that the primary switch is no longer conductive.
  • One drawback of the method of Figure 18A is that the use of the resistor will cause power loss.
  • FIG. 18B A preferred method for sensing the state of the primary switch as a means of safely enabling the reset switch is shown in Figure 18B.
  • a diode 410 a large value resistor 412 and bias source 414, of voltage Vb ⁇ Vin, are used as sensing elements.
  • Vs a small voltage value
  • the switch 333 begins to carry the current and the voltage Vs increases to a value Vk, which is more positive than Vz and approximately equal to the voltage drop across the switch, Vsw.
  • Vk the voltage drop across the switch
  • the current which flows in the transformer primary, Ip will, as shown in Figures 21A through 21C, consist of a magnetizing current component, Im, and a forward current component, Io (e.g., in a ZCS converter, as shown in Fig. 21B, Io is the approximately sinusoidal portion of the transformer primary current which is reflected into the secondary winding).
  • Io a forward current component
  • the primary current, Ip is negative, and this flow of negative current will discharge the parasitic capacitance 343 of the primary switch 20 if the switch is off.
  • FIG. 20 shows a reset switch controller 520 which incorporates some of the features described above.
  • a primary switch controller 530 delivers Primary Switch Enable (MS) pulses 511 indicating when the primary switch 20 is to be turned on. Assuming that the reset switch 332 is turned on (e.g., the Reset Switch signal output of flip-flop 502 is high), delivery of a PSE pulse will cause both flip-flops 502, 504 to be reset via their reset inputs. This will cause the Reset Switch signal 501 to go low, initiating turn-off of the reset switch, and will also cause the Disable signal 515 to go high.
  • MS Primary Switch Enable
  • the Primary Switch Controller 530 will not deliver another PSE pulse unless the Disable signal 515 is low.
  • the result is that the primary switch 20 is turned on a short time after the reset switch is turned off.
  • the dead time is determined by circuit propagation times and the delay introduced by delay element 510 -- in a typical 1 Mhz ZCS converter the delay might be a few tens of nanoseconds. Provided that the delay is relatively short, its value is non-critical since the transformer magnetizing current which flows during the dead time will beneficially charge parasitic capacitances associated with both switches.
  • This causes the output of the comparator 500 to go low, bringing the Disable signal low. This, in turn, signals the primary switch controller 530 that it is safe for it to deliver a subsequent PSE pulse 511. Some time later (e.g., at approximately time t tf) another PSE pulse is generated and the cycle is repeated.
  • topologies and circuits described above may be adapted for use in other than single-ended converters (e.g.; full bridge or half-bridge converters).
EP02078757A 1995-01-17 1996-01-09 Steuerung der gespeicherten magnetischen Energie in Stromwandlertransformatoren Withdrawn EP1278295A3 (de)

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Application Number Priority Date Filing Date Title
US37311295A 1995-01-17 1995-01-17
US373112 1995-01-17
EP96300150A EP0723331B1 (de) 1995-01-17 1996-01-09 Regelung der in den Transformatoren von Schaltnetzteilen gepseicherten Energie

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EP1278295A2 true EP1278295A2 (de) 2003-01-22
EP1278295A3 EP1278295A3 (de) 2004-12-29

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EP02078757A Withdrawn EP1278295A3 (de) 1995-01-17 1996-01-09 Steuerung der gespeicherten magnetischen Energie in Stromwandlertransformatoren

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US5805434A (en) 1998-09-08
EP1278295A3 (de) 2004-12-29
DE69627909T2 (de) 2003-11-13
EP0723331B1 (de) 2003-05-07
USRE38196E1 (en) 2003-07-22
JPH0937548A (ja) 1997-02-07
DE69627909D1 (de) 2003-06-12
EP0723331A3 (de) 1997-04-23
EP0723331A2 (de) 1996-07-24
JP2781768B2 (ja) 1998-07-30

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