WO2020248472A1 - 不对称半桥变换器及控制方法 - Google Patents

不对称半桥变换器及控制方法 Download PDF

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Publication number
WO2020248472A1
WO2020248472A1 PCT/CN2019/113679 CN2019113679W WO2020248472A1 WO 2020248472 A1 WO2020248472 A1 WO 2020248472A1 CN 2019113679 W CN2019113679 W CN 2019113679W WO 2020248472 A1 WO2020248472 A1 WO 2020248472A1
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Prior art keywords
switch
network
anode
winding
capacitor
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PCT/CN2019/113679
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English (en)
French (fr)
Inventor
任鹏程
杜波
王志燊
李璐
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广州金升阳科技有限公司
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Publication of WO2020248472A1 publication Critical patent/WO2020248472A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to a switching converter, in particular to an asymmetric half-bridge converter and its control method.
  • Switching converters that have been developed and applied since the 1960s mostly use hard switching technology. However, with the development of science and technology, all walks of life have put forward higher requirements for power supplies, and high efficiency, high power density, and miniaturization have become the main topics of power supply industry research. Switching converters using hard switching technology have disadvantages such as large switching loss, low efficiency, low switching frequency, and poor EMI. For this reason, soft switching technology has emerged.
  • the so-called soft switching refers to: Zero-Voltage-Switching , Referred to as ZVS; Zero-Current-Switching, referred to as ZCS.
  • the soft switching technology mainly uses the principle of resonance to make the current (or voltage) in the switching device of the switching converter change according to the sine or quasi-sine law.
  • the switch When the current of the switching device naturally crosses zero, the switch is turned off, or the voltage of the switching device is zero , The switch is turned on, so that the loss of the switching device is zero. Therefore, it is possible to increase the switching frequency of the switching converter to the megahertz level, which provides the possibility for the high efficiency, high power density and miniaturization of the switching power supply.
  • a clamp circuit 330 is added at both ends of the transformer primary winding of the forward circuit, and the auxiliary switch 332 and the main switch 20 drive signals Complementaryly, when the main switch 20 is turned off, the clamping capacitor 334 absorbs the leakage inductance energy and reverses the current.
  • the diode 350 provides a freewheeling loop for the reverse current to achieve near lossless freewheeling, and suppress the clamping capacitor 334 and the excitation inductance and leakage. Sense resonance.
  • the main purpose of the clamp circuit 330 is to recover the leakage inductance energy, suppress resonance, and optimize EMI.
  • the clamp circuit does not achieve the effect of the zero voltage turn-on of the main switch.
  • Vicor Company also proposed a double clamp ZVS BUCK-BOOST topology, patent number US7561446, as shown in Figure 2-1 and Figure 2-2.
  • switch Q3 turns off and switch Q4 turns on.
  • the residual current of the transformer continues to flow in the clamping circuit, and the switch Q2 is turned off before the switch Q1 is turned on.
  • the residual current of the transformer makes the switch Q1 turn on at zero voltage.
  • This technology emphasizes that the secondary side current reaches zero, the switch Q3 is turned off and the switch Q4 is turned on, and the circuit connection mode is quite different from that of the asymmetric half-bridge flyback converter.
  • the asymmetric half-bridge flyback converter can use the magnetizing inductance and leakage inductance of the transformer to realize the ZVS and ZCS of the switching device in the full input/full load range. This improves the efficiency of switching power supply products, reduces the size, and simplifies the manufacturing process. Improvements in EMI have brought possibilities.
  • Figure 4 shows a circuit diagram of the conventional asymmetrical half-bridge flyback converter comprising an input capacitor Cin, a main switch and an auxiliary switch S M S A, resonant capacitor Cr, transformer Tr, the rectifier switch S D, the output filter capacitor Co.
  • the transformer Tr includes a primary winding and a secondary winding.
  • the anode of the primary winding of the transformer Tr and one end of the secondary winding have the same name for each other, and the anode of the primary winding of the transformer Tr and the other end of the secondary winding have the same name for each other; resonance; Cr the capacitor comprising a resonant transformer Tr energy transmission network, said network comprising a resonant energy transfer two input terminals and two output terminals; one ends of the main switch S M S a and is connected to the auxiliary switch, half-bridge structure consisting of, connected referred to as the switching node SW node, another end of the main switch S M and the input capacitance Cin, the input n + Vin is connected to other ends of the switches S a to the secondary input capacitance Cin, the input connected to the negative -Vin; resonant capacitor One end of Cr is connected to the anode of the primary winding of the transformer Tr, the other end of the resonant capacitor Cr is used as an input end of the resonant energy transmission network and is connected to the switch no
  • the equivalent circuit principle diagram of the existing asymmetric half-bridge flyback converter can be obtained, as shown in Figure 5.
  • the difference from Figure 4 is that the transformer Tr is equivalent to leakage
  • Leakage inductance Lr is used as the resonant inductance of the resonant energy transmission network (persons skilled in the art also often use external inductance and resonant capacitor Cr to resonate), which is recorded as resonant inductance Lr.
  • One end of the inductance Lr is connected to one end of the excitation inductance Lm and one end of the primary winding of the ideal transformer T.
  • the other end of the resonant inductance Lr serves as one end of the primary winding of the transformer Tr.
  • the other end of the excitation inductance Lm is connected to the primary winding of the ideal transformer T.
  • the other end of the winding is connected as the other end of the primary winding of the transformer Tr; the two ends of the main switch S M are connected in parallel with a diode D1 and a capacitor C1.
  • FIG. 6 is a typical working waveform diagram of an existing asymmetric half-bridge flyback converter working in CCM mode.
  • the existing asymmetric half-bridge flyback converter works in CCM mode.
  • Each cycle includes four phases: the excitation phase, The auxiliary switch zero voltage turn-on stage, the demagnetization stage, and the main switch zero voltage turn-on stage.
  • the four stages are described with reference to Figure 5, which are specifically as follows:
  • the zero voltage turn-on phase of the auxiliary switch starts from time t1 to time t2.
  • the main switch S M is turned off.
  • the capacitor C1, the capacitor C2, the resonant inductor Lr and the magnetizing inductance Lm form a series resonance, and the resonant inductor current I Lr feeds the capacitor C1 charging, the capacitor C2 is discharged, so that the voltage V C1 across the capacitor C1 rises, the voltage V C2 across the capacitor C2 drops to V C2 and V Cr reduced to the same voltage, the voltage across the magnetizing inductance Lm is zero, the voltage across the capacitor C2 V C2 continues to decrease and the polarity of the magnetizing inductance Lm changes, causing the resonant inductor Lr to bear a negative voltage.
  • the resonant inductor current I Lr is still positive, but the resonant inductor current I Lr begins to increase negatively, and the resonant inductor current I Lr Continue to charge the capacitor C1 and discharge the capacitor C2 until the capacitor C2 is discharged, V C2 drops to zero, the diode D2 naturally turns on, and the resonant inductor current I Lr flows through the diode D2.
  • the magnetizing inductance voltage across Lm is clamped to NV o (N is the transformer turns ratio), the secondary side S D rectifier switch is turned on, the resonant inductor Lr and the resonant capacitor Cr resonate, the electric energy stored in the transformer Tr began to shift to the secondary side, to time t2, Vgs2 of high level, the switch S A auxiliary control turned on, the auxiliary switch to achieve ZVS S A;
  • the rectifier switch S D in the switch current I D S a secondary off time could be decreased to zero, when the auxiliary switch S a turn-off instant of the rectifier switch current I D S D has been reduced to zero, compared with the DCM, if the auxiliary switching off S a time of the rectifier switch current I D S D is not zero, for the CCM mode.
  • the working principle of the existing asymmetric half-bridge flyback converter working in the DCM mode is not repeated here, and relevant engineering and technical personnel can derive it by themselves.
  • Asymmetric half-bridge flyback converter if the design meets the requirement that the full-load main switch realizes zero-voltage turn-on, the main switch will be easier to achieve zero-voltage turn-on at light load and no-load, and when the load is reduced, the duty cycle remains unchanged, the excitation There will be a large negative current in the inductor current I Lm . This negative current far exceeds the requirement for the main switch of the converter to achieve zero voltage turn-on. Excessive current will flow in the resonant cavity, resulting in large losses, resulting in Low light load efficiency and large no-load power consumption;
  • Asymmetrical half-bridge flyback converter if the design meets the requirement of low voltage full load, the main switch realizes zero voltage turn-on, then when high voltage full load, the main switch will inevitably realize zero voltage turn-on, and the high-voltage full-load excitation inductor current I Lm will have a large negative The negative current far exceeds the requirement of the main switch of the converter to achieve zero voltage turn-on.
  • the present invention proposes an asymmetric half-bridge converter, which can solve the problems of low light load efficiency, large no-load power consumption, and difficulty in balancing low-voltage input and high-voltage input in the existing technical solutions.
  • the control method of asymmetric half-bridge converter is simpler and more efficient.
  • the inventive concept of this application is: on the basis of the existing asymmetric half-bridge flyback converter, add a unidirectional clamping network electrically connected to both ends of the transformer primary winding or secondary winding or tertiary winding. a particular time of the switching cycle, when the magnetizing inductor current reaches the set value, the auxiliary control switch S a is turned off, current flows through the unidirectional clamp clamping network, one-way clamping network remains clamped substantially constant current to before opening the main switch S M release the clamp current period, the main switch S M to achieve zero-voltage.
  • the present invention proposes a control method for an asymmetric half-bridge flyback converter.
  • Each cycle includes five phases: excitation phase, auxiliary switch zero voltage turn-on phase, demagnetization phase, current clamping phase, and main switch Zero voltage turn-on stage; the duration of excitation stage and demagnetization stage are related to the input voltage and load.
  • the duration of the auxiliary switch zero-voltage turn-on stage and the main switch zero-voltage turn-on stage are related to the design of the main power device.
  • Current clamping stage The duration is determined by the duration of the cycle period and the duration of the excitation phase, the duration of the demagnetization phase, the duration of the auxiliary switch's zero-voltage turn-on phase, and the duration of the main switch's zero-voltage turn-on phase to ensure a fixed duration of each cycle period. That is, fixed frequency pulse width modulation (PWM) control.
  • PWM pulse width modulation
  • An asymmetric half-bridge converter comprising a primary circuit and secondary circuit of the transformer Tr: the primary circuit includes an input capacitor Cin, a main switch and an auxiliary switch S M S A, the resonant capacitor of Cr; comprising primary winding of transformer Tr and The secondary winding, the anode of the primary winding of the transformer Tr and one end of the secondary winding have the same name each other, the anode of the primary winding of the transformer Tr and the other end of the secondary winding have the same name each other; the secondary circuit includes a rectifier switch SD after the original resonant capacitor Cr and the transformer Tr windings connected in series, in series; and an output capacitance of Co; input capacitor Cin one end connected to input n, and the other end connected to the input negative; after the main switch S M and auxiliary switch S a in series with the input capacitor Cin in parallel end connection point of the main switch S M and auxiliary switches S a, and the other end of the series connected input positive or negative input; the transformer Tr of the secondary winding and the rectifier switch
  • the anode of the unidirectional clamp network is electrically connected with the anode of the transformer primary winding, and the cathode of the unidirectional clamp network is electrically connected with the cathode of the transformer primary winding;
  • the anode of the unidirectional clamp network is electrically connected with the end of the transformer secondary winding with the same name, and the cathode of the unidirectional clamp network is electrically connected with the different end of the transformer secondary winding;
  • the asymmetric half-bridge converter also includes a third winding.
  • the anode of the primary winding of the transformer and one end of the third winding have the same name for each other, and the anode of the transformer primary winding and the other end of the third winding have different names for each other.
  • the anode of the unidirectional clamp network is electrically connected with the end of the third winding with the same name, and the cathode of the unidirectional clamp network is electrically connected with the end of the third winding with the same name.
  • the asymmetric half-bridge converter is an asymmetric half-bridge flyback converter.
  • the asymmetric half-bridge converter is an asymmetric half-bridge forward converter.
  • the third winding is an independent winding.
  • the asymmetric half-bridge converter includes the third winding
  • the third winding and the auxiliary winding are the same winding.
  • the unidirectional clamp network As the first specific implementation of the unidirectional clamp network, it is characterized in that it includes a diode and a switch tube, and the connection relationship is one of the following two types:
  • the anode of the diode is the anode of the unidirectional clamp network
  • the cathode of the diode is connected to the drain of the switch tube
  • the source of the switch tube is the cathode of the unidirectional clamp network
  • the drain of the switch tube is the anode of the unidirectional clamp network
  • the source of the switch tube is connected to the anode of the diode
  • the cathode of the diode is the cathode of the unidirectional clamp network
  • the parasitic capacitance of the anode and cathode of the diode is C Dow
  • the parasitic capacitance of the drain and source of the switch tube is C Qow
  • the input voltage is V in
  • the output voltage is V o
  • the turns ratio of the primary winding and the secondary winding or the third winding is N, and each parameter satisfies the following relationship:
  • the one-way clamping network includes two switching tubes, the sources of the two switching tubes are connected, and the drain of one switching tube is a one-way clamping
  • the anode of the network and the drain of the other switch tube are the cathode of the unidirectional clamp network.
  • the unidirectional clamping network includes a diode, a switch tube and a capacitor, and the connection relationship is one of the following two:
  • the anode of the diode is connected to one end of the capacitor, which is the anode of the unidirectional clamp network, the cathode of the diode is connected to the other end of the capacitor and the drain of the switch tube, and the source of the switch tube is the cathode of the unidirectional clamp network;
  • the cathode of the diode is connected to one end of the capacitor, which is the cathode of the unidirectional clamp network, the anode of the diode is connected to the other end of the capacitor and the source of the switch tube, and the drain of the switch tube is the anode of the unidirectional clamp network.
  • the parasitic capacitance of the anode and cathode of the diode is C Dow
  • the parasitic capacitance of the drain and source of the switch tube is C Qow
  • the input voltage is V in
  • the output voltage is V o
  • the turns ratio of the primary winding and the secondary winding or the third winding is N, and each parameter satisfies the following relationship:
  • the unidirectional clamping network includes a diode, a switch tube, a voltage regulator tube and a capacitor, and the connection relationship is one of the following two One:
  • the anode of the diode is connected to the anode of the zener tube, which is the anode of the unidirectional clamp network.
  • the cathode of the diode is connected to one end of the capacitor and the drain of the switch tube, and the cathode of the zener tube is connected to the other end of the capacitor.
  • the source of the tube is the cathode of the unidirectional clamp network;
  • the cathode of the diode is connected to the cathode of the regulator tube, which is the cathode of the unidirectional clamp network.
  • the anode of the diode is connected to one end of the capacitor and the source of the switch tube, and the anode of the regulator tube is connected to the other end of the capacitor.
  • the drain of the tube is the anode of the unidirectional clamp network.
  • the present invention also provides the above-mentioned control method of the asymmetric half-bridge converter, and the technical solution is as follows:
  • a control method of an asymmetric half-bridge converter characterized in that: each cycle includes five stages: excitation stage, auxiliary switch zero voltage turn-on stage, demagnetization stage, current clamp stage, and main switch zero voltage turn-on stage ;
  • the unidirectional clamp network is turned off
  • the auxiliary switch In the demagnetization phase, the auxiliary switch is turned on, and the unidirectional clamp network can be turned on or off. No current flows through the unidirectional clamp network; at the end of this phase, the excitation inductor current reaches the set value and the auxiliary switch is turned off , The one-way clamp network is in the conducting state, and the clamp current flows through the one-way clamp network;
  • the one-way clamping network In the current clamping stage, the one-way clamping network is turned on, the clamping current flows through the one-way clamping network, and the one-way clamping network keeps the clamping current basically unchanged. At the end of this stage, the one-way clamping network is turned off ;
  • the one-way clamp network has been turned off, and the clamp current in the one-way clamp network is released, so that the main switch voltage is reduced to zero or close to zero.
  • the main switch is controlled to turn on. Realize the zero voltage turn-on of the main switch.
  • the setting value of the magnetizing inductor current is negative or zero, and is related to the input voltage.
  • the second preferred solution is that the duration of each switching cycle is the same.
  • control uses a pulse width modulation control method.
  • the center switching frequency of the switching cycle period is fixed, and the actual switching frequency is centered on the center switching frequency and periodically changes between the set lower limit frequency and the set upper limit frequency.
  • the anode of the unidirectional clamp network the end of the DC current flowing in from the unidirectional clamp network is the anode;
  • the cathode of the unidirectional clamp network the end of the DC current flowing out of the unidirectional clamp network is the cathode;
  • Clamp current unidirectional current flowing through the clamping network, specifically refers to, in the demagnetization phase, magnetizing the inductor current reaches the set value, the auxiliary control switch S A is turned off, current flows through the magnetizing inductance of the one-way clamping network Either the current coupled to the secondary winding through the transformer or the current coupled to the third winding through the transformer;
  • the anode of the primary winding During the excitation phase, the main pipe SM is turned on, and the end of the DC current flowing in from the primary winding is the anode of the primary winding;
  • the cathode of the primary winding during the excitation phase, the main pipe SM is turned on, and the end of the DC current flowing out of the primary winding is the cathode of the primary winding;
  • Positive direction of excitation inductance current flows in from the anode of the primary winding of the transformer, flows through the excitation inductance, and flows out from the cathode of the primary winding;
  • End with the same name refers to the ends of the two windings of the transformer with the same potential polarity at any time under the action of the same alternating magnetic flux;
  • Synonymous end refers to the ends of the two windings of the transformer with opposite potential polarity at any time under the action of the same alternating magnetic flux;
  • Direct connection In addition to direct connection, it also includes indirect connection (that is, other components can be connected between two electrical connection objects), and includes inductive coupling.
  • the technical scheme of the present invention can realize the effective control of the negative peak value of the excitation inductance current on the primary side of the transformer, thereby achieving the goal of controlling the negative current of the excitation inductance required to realize zero voltage turn-on, reducing the current flowing in the resonant cavity and reducing the converter
  • the effective value of the current of the power device under light no-load so as to greatly improve the light-load efficiency of the converter and reduce the no-load loss while retaining the advantages of the existing technical solution that can achieve zero voltage turn-on, making the asymmetric half-bridge flyback
  • the converter can better take into account the low-voltage input and the high-voltage input, and realize the requirement of wide input.
  • the inventive concept can also be used in an asymmetric half-bridge forward converter.
  • the fixed frequency pulse width modulation (PWM) control mode makes the control realization simpler and more efficient.
  • Figure 1 is attached drawing 10 of Vicor's patent US5805434 specification
  • FIG 2-1 is the attached drawing 1 of Vicor's patent US7561446 specification
  • Figure 2-2 is the attached drawing 3 of Vicor's patent US7561446 specification
  • FIG 3 is attached drawing 2 of Astec's patent US9973098 specification
  • Figure 4 is a circuit diagram of an existing asymmetric half-bridge flyback converter
  • Figure 5 is a schematic diagram of the equivalent circuit of the existing asymmetric half-bridge flyback converter
  • Figure 6 is a working waveform diagram of the existing asymmetric half-bridge flyback converter in CCM mode
  • FIG. 7 is a circuit diagram of an asymmetric half-bridge flyback converter according to the first embodiment of the present invention.
  • FIG. 8 is a schematic diagram of the equivalent circuit of the asymmetric half-bridge flyback converter according to the first embodiment of the present invention.
  • Fig. 9 is a CCM mode operating waveform of the asymmetric half-bridge flyback converter according to the first embodiment of the present invention.
  • FIG. 10 is a DCM mode operating waveform of the asymmetric half-bridge flyback converter according to the first embodiment of the present invention.
  • Figure 11-1 is an efficiency curve diagram of the existing scheme and the scheme of the first embodiment of the present invention under 120V input;
  • Figure 11-2 is an efficiency curve diagram of the existing scheme and the scheme of the first embodiment of the present invention under 160V input;
  • Figure 11-3 is an efficiency curve diagram of the existing scheme and the scheme of the first embodiment of the present invention under 320V input;
  • Figure 11-4 is an efficiency curve diagram of the existing scheme and the scheme of the first embodiment of the present invention at 370V input;
  • FIG. 12 is a circuit diagram of an asymmetric half-bridge flyback converter according to a second embodiment of the present invention.
  • FIG. 13 is a circuit diagram of an asymmetric half-bridge flyback converter according to a third embodiment of the present invention.
  • FIG. 14 is a circuit diagram of an asymmetric half-bridge flyback converter according to a fourth embodiment of the present invention.
  • FIG. 15 is a circuit diagram of an asymmetric half-bridge forward converter according to a fifth embodiment of the present invention.
  • 16 is a schematic diagram of an equivalent circuit of an asymmetric half-bridge forward converter according to a fifth embodiment of the present invention.
  • Figure 17 is a typical working waveform of the asymmetric half-bridge forward converter according to the fifth embodiment of the present invention.
  • FIG. 18 is a circuit diagram of an asymmetric half-bridge forward converter according to a sixth embodiment of the present invention.
  • FIG. 19 is a circuit diagram of an asymmetric half-bridge forward converter according to a seventh embodiment of the present invention.
  • 20 is a circuit diagram of an asymmetric half-bridge forward converter according to an eighth embodiment of the present invention.
  • 21-1 to 21-7 are specific implementation circuits of the unidirectional clamp network according to the first to eighth embodiments of the present invention.
  • Fig. 7 is a circuit diagram of the first embodiment of the present invention
  • Fig. 8 is a schematic diagram of the equivalent circuit of the asymmetric half-bridge flyback converter of the first embodiment of the present invention
  • Fig. 7 is different from Fig. 4, Fig. 8 and Fig. 5
  • the point is: add a unidirectional clamping network Sow to the primary side of the transformer, the anode of the unidirectional clamping network Sow is electrically connected to the anode of the transformer primary winding, and the cathode of the unidirectional clamping network Sow is electrically connected to the cathode of the transformer primary winding.
  • FIG. 9 is a typical working waveform diagram of the asymmetric half-bridge flyback converter working in CCM mode according to the first embodiment of the present invention.
  • Each cycle includes five phases: excitation phase, auxiliary switch zero voltage turn-on phase, and demagnetization phase , Current clamping stage, the zero voltage turn-on stage of the main switch.
  • the five stages described in each cycle (from time t0 to time t5, denoted as T) are now described in conjunction with Fig. 8. The details are as follows:
  • the control signal Vgs1, the control signal Vgs2, and the resonant inductor current I Lr, the main switch S M is the drain-source voltage Vds1, the same waveform and waveform of the current I D prior art rectifier switch S D in FIG. 6;
  • Auxiliary zero-voltage switching stages from time t1 until time t2 stopper (referred to as Tl), main switch S M is off, the capacitor C1, the capacitor C2, the resonant inductor Lr and magnetizing inductance Lm form a series resonant, resonant inductor current I Lr charges the capacitor C1 and discharges the capacitor C2, causing the voltage V C1 across the capacitor C1 to rise, and the voltage V C2 across the capacitor C2 to drop.
  • auxiliary control switches S A is turned on, the main switch S M is still turned off, the rectifier switch S D turned on, the rectifier switch current I D S D is increased , the voltage across the magnetizing inductance Lm is clamped at the negative voltage on the positive, the I Lm magnetizing inductor current decreases linearly, transformer demagnetization, t3 time, magnetizing the inductor current reaches the set value, the auxiliary control switch S a oFF.
  • the control signal Vgs3 is at a high level at this stage, the added one-way clamping network Sow is turned on, and the opening time of the one-way clamping network Sow can be any time between t2 and t3 ( That is, the unidirectional clamping network Sow between t2 and t3 can be turned on or off). Since the unidirectional clamping network Sow only allows current to flow from its anode to the cathode, there is no unidirectional clamping network Sow in the process.
  • this phase control signal Vgs1, Vgs2 of the control signal, the resonant inductor current I Lr, drain-source voltage Vds1 of the main switch S M, the waveform of the current I D waveform and prior art rectifier switch S D in FIG. 6 is the same ;
  • the resonant inductor current I Lr is negative and rapidly increases positively, discharging the capacitor C1 and charging the capacitor C2, so that the voltage V C1 across the capacitor C1 drops and the voltage V C2 across the capacitor C2 rises to V C2 rise with the same voltage V Cr, unidirectional clamp network Sow anode voltage is zero, is equal to current I Lm magnetizing inductance and the resonant inductor current I Lr, S D rectifier switch is turned off, the inductor magnetizing current I Lm (also known as clamp Bit current) naturally flows from the anode to the cathode through the unidirectional clamping network Sow.
  • the unidirectional clamping network Sow keeps the clamping current basically unchanged.
  • the control signal Vgs3 becomes low and the unidirectional clamping network Sow is turned off. ;
  • the excitation phase duration T0 and the demagnetization phase duration T2 are related to the input voltage and load, and the auxiliary switch zero voltage turn-on phase duration T1
  • the duration of the zero-voltage turn-on phase of the main switch T4 is related to the design of the main power device.
  • the duration of the current clamping phase T3 is determined by T and T0, T2, T1, and T4. That is, T3 changes with the input voltage and load. Ensure that the duration of each cycle period T is fixed, that is, realize the fixed frequency pulse width modulation (PWM) control.
  • PWM pulse width modulation
  • FIG. 10 shows the first embodiment of the present invention, a typical working waveform diagram of an asymmetric half-bridge flyback converter working in DCM mode.
  • Each cycle of the asymmetric half-bridge flyback converter includes five Two stages: excitation stage, auxiliary switch zero voltage turn-on stage, demagnetization stage, current clamping stage, and main switch zero voltage turn-on stage.
  • output rectifier switch current I D S D of the switch S A in the secondary off time could be decreased to zero, if the current is turned off auxiliary switching time S A S D output of the rectifier switch I D has been reduced to zero, compared with the DCM, if the auxiliary switch S a turn-off instant of the output rectifier switch current I D S D is not zero, for the CCM mode.
  • the working principle of the first embodiment of the present invention working in the DCM mode will not be repeated here, and relevant engineering and technical personnel can derive it by themselves.
  • the duty cycle D hardly changes.
  • the peak current I Lm _peak of the magnetizing inductance will have a large negative peak value.
  • the load change of the scheme of the present invention In the process, the duty cycle D decreases as the load decreases, the positive and negative peak values of the magnetizing inductance peak current I Lm _peak are reduced to a certain extent, and the effective value of the magnetizing inductance current I Lm _rms is also greatly reduced;
  • the peak value and effective value of the excitation inductance are large, resulting in low light load efficiency, and this situation becomes more serious with the increase of the input voltage, resulting in serious degradation of the high voltage input light load efficiency. The degree has improved this problem.
  • Figures 11-1, 11-2, 11-3, and 11-4 show the efficiency curves of the existing scheme and the scheme of the first embodiment of the present invention under 120V, 160V, 320V, and 370V input respectively. It can be clearly seen that, Under different input voltages, the light-load efficiency of the technical solution of the present invention is significantly improved compared with the existing solutions.
  • the frequency jitter function can be added on the basis of the technical solution of the present invention, that is: the center switching frequency of the switching cycle is fixed, the actual switching frequency is centered on the center switching frequency, and the lower limit frequency is set at The upper frequency changes periodically.
  • Figure 12 is a circuit diagram of the second embodiment of the present invention, an asymmetric half-bridge flyback converter.
  • the main difference between the second embodiment of the present invention and the first embodiment is the difference in the connection mode of the unidirectional clamping network Sow: first
  • a one-way clamping network Sow is added to the primary side of the transformer.
  • the anode of the one-way clamping network Sow is electrically connected to the anode of the transformer primary winding
  • the cathode of the one-way clamping network Sow is connected to the cathode of the transformer primary winding.
  • a one-way clamping network Sow is added to the secondary side of the transformer, the anode of the one-way clamping network Sow is electrically connected to the same-name end of the transformer secondary winding, and the cathode of the one-way clamping network Sow is connected to the transformer The opposite ends of the secondary winding are electrically connected.
  • the connection methods are different, according to the principle of coupling between the primary winding and the secondary winding of the transformer, it can be known that the secondary winding is connected through the unidirectional clamping network Sow, and the clamping current flows through the unidirectional clamping network Sow, which can also be realized. The effect of clamping and maintaining the magnetizing inductance current I Lm .
  • an asymmetric half-bridge flyback converter includes five phases in each cycle: excitation phase, auxiliary switch zero voltage turn-on phase, demagnetization phase, current clamping phase, and main switch zero voltage turn-on stage.
  • the main difference from the working process of the asymmetric half-bridge flyback converter described in the first embodiment is: the current clamping phase, the time when the unidirectional clamping network Sow is turned on is different.
  • the one-way clamping network Sow can be turned on at any time between t2 and t3.
  • the one-way clamping network Sow can only be turned on at the time t3.
  • Fig. 13 is a circuit diagram of the third embodiment of the present invention, an asymmetric half-bridge flyback converter.
  • the transformer Tr also includes a third winding Np_ow; While adding a unidirectional clamping network Sow, the anode of the unidirectional clamping network Sow is electrically connected with the anode of the transformer primary winding, and the cathode of the unidirectional clamping network Sow is electrically connected with the cathode of the transformer primary winding; the third embodiment In the third winding Np_ow of the transformer, a one-way clamping network Sow is added. The anode of the one-way clamping network Sow is electrically connected to the end of the third winding Np_ow with the same name.
  • the cathode of the one-way clamping network Sow has the same name as the third winding Np_ow. Terminal electrical connection. Although the connection methods are different, according to the principle of coupling between the primary winding of the transformer and the third winding: the third winding Np_ow is connected through the unidirectional clamping network Sow, and the clamping current flows through the unidirectional clamping network Sow. The effect of clamping and maintaining the magnetizing inductance current I Lm is achieved.
  • an asymmetric half-bridge flyback converter includes five phases in each cycle: excitation phase, auxiliary switch zero voltage turn-on phase, demagnetization phase, current clamp phase, and main switch zero voltage turn-on stage.
  • the main difference from the working process of the asymmetric half-bridge flyback converter described in the first embodiment is: the current clamping phase, the time when the unidirectional clamping network Sow is turned on is different.
  • the one-way clamping network Sow can be turned on at any time between t2 and t3.
  • the one-way clamping network Sow can only be turned on at the time t3.
  • the third winding Np_ow in the third embodiment of the present invention can be used independently, that is, as an independent winding; at the same time, the third winding Np_ow can also be used as an auxiliary winding, that is, the third winding Np_ow and the auxiliary winding are the same winding.
  • FIG. 14 is a circuit diagram of the fourth embodiment of the present invention.
  • the main difference between the fourth embodiment and the first embodiment lies in the connection mode of the resonant energy transmission network.
  • an input terminal of the resonant energy transmission network and the switch The node SW is connected, the other input terminal of the resonance energy transmission network is connected with the input negative -Vin, one input terminal of the resonance energy transmission network is connected with the switch node SW in the fourth embodiment, and the other input terminal of the resonance energy transmission network is connected with the input Positive +Vin is connected.
  • an asymmetric half-bridge flyback converter includes five phases in each cycle: excitation phase, auxiliary switch zero voltage turn-on phase, demagnetization phase, current clamp phase, and main switch zero voltage turn-on stage. This is the main difference between the half-bridge asymmetric flyback converter is also in the first embodiment: the different main switch and an auxiliary switch S M S A vertical position on the half-bridge structure.
  • FIG. 15 is a circuit diagram of an asymmetric half-bridge forward converter according to a fifth embodiment of the present invention
  • FIG. 16 is a schematic diagram of an equivalent circuit of an asymmetric half-bridge forward converter according to a fifth embodiment of the present invention.
  • the idea of the technical solution of one embodiment is applied to the asymmetric half-bridge forward converter, and the technical solution of the fifth embodiment of the present invention can be obtained through deduction. Therefore, the difference between FIG. 15 and FIG. 7, and FIG. 16 and FIG. The position of the black dot of the end relationship is different.
  • Fig. 17 is a fifth embodiment of the present invention, a typical working waveform diagram of an asymmetric half-bridge forward converter.
  • Each cycle of the asymmetric half-bridge forward converter includes five phases: the excitation phase, Auxiliary switch zero voltage turn-on stage, demagnetization stage, current clamping stage, main switch zero voltage turn-on stage.
  • the details of the specific working principle of the fifth embodiment of the present invention will not be repeated here, and those skilled in the art can deduce it according to the working principle of the first embodiment in conjunction with FIG. 16.
  • Figure 18 is a circuit diagram of the sixth embodiment of the present invention.
  • the difference from Figure 15 is that a one-way clamping network Sow is added to the secondary side of the transformer.
  • the anode of the one-way clamping network Sow is electrically connected to the same-named end of the transformer secondary winding.
  • the cathode of the clamping network Sow is electrically connected with the alias end of the secondary winding of the transformer.
  • the difference between Fig. 18 and Fig. 12 is that the positions of the black dots indicating the relationship of the end with the same name are different.
  • an asymmetric half-bridge forward converter includes five phases per cycle: excitation phase, auxiliary switch zero voltage turn-on phase, demagnetization phase, current clamp phase, and main switch zero voltage turn-on stage.
  • excitation phase auxiliary switch zero voltage turn-on phase
  • demagnetization phase demagnetization phase
  • current clamp phase current clamp phase
  • main switch zero voltage turn-on stage main switch zero voltage turn-on stage
  • Figure 19 is a circuit diagram of the seventh embodiment of the present invention.
  • the transformer Tr also includes a third winding Np_ow.
  • the third winding Np_ow of the transformer adds a unidirectional clamping network Sow.
  • the anode of the unidirectional clamping network Sow is connected to
  • the third winding Np_ow of the transformer is electrically connected with the same name end, and the cathode of the unidirectional clamping network Sow is electrically connected with the different end of the third winding Np_ow of the transformer.
  • the difference between Fig. 19 and Fig. 13 is that the positions of the black dots indicating the relationship of the end with the same name are different.
  • an asymmetric half-bridge forward converter includes five phases in each cycle: excitation phase, auxiliary switch zero voltage turn-on phase, demagnetization phase, current clamp phase, and main switch zero voltage turn-on stage.
  • excitation phase auxiliary switch zero voltage turn-on phase
  • demagnetization phase demagnetization phase
  • current clamp phase current clamp phase
  • main switch zero voltage turn-on stage main switch zero voltage turn-on stage
  • FIG. 20 is a circuit diagram of the eighth embodiment of the present invention.
  • the difference from FIG. 15 is that the other input terminal of the resonance energy transmission network is connected to the input +Vin.
  • the difference between FIG. 20 and FIG. 14 is that the positions of the black dots indicating the relationship of the end with the same name are different.
  • an asymmetric half-bridge forward converter includes five phases in each cycle cycle: excitation phase, auxiliary switch zero voltage turn-on phase, demagnetization phase, current clamp phase, and main switch zero voltage turn-on stage.
  • excitation phase auxiliary switch zero voltage turn-on phase
  • demagnetization phase demagnetization phase
  • current clamp phase current clamp phase
  • main switch zero voltage turn-on stage main switch zero voltage turn-on stage
  • FIGS 21-1 to 21-7 show the specific implementation circuits of the one-way clamping network Sow in the first to eighth embodiments, through the diode Dow, the voltage regulator tube Zow, the capacitor Cow, the switching tube Qow, the switching tube Qow1 and the switch Different combinations of tube Qow2 can achieve the effect of unidirectional clamping network Sow.
  • the one-way clamp network Sow As a specific implementation of the one-way clamp network Sow, as shown in Figure 21-1, it includes a diode Dow and a switch Qow.
  • the cathode of the diode Dow is connected to the drain of the switch Qow, and the anode of the diode Dow is used as a one-way clamp.
  • the anode of the bit network Sow, the source of the switch Qow are used as the cathode of the unidirectional clamp network Sow, and the gate of the switch Qow is used to receive the control signal Vgs3.
  • the one-way clamp network Sow contains a diode Dow and a switch Qow.
  • the anode of the diode Dow is connected to the source of the switch Qow, and the cathode of the diode Dow is used as a one-way clamp.
  • the cathode of the bit network Sow, the drain of the switch Qow is used as the anode of the unidirectional clamping network Sow, and the gate of the switch Qow is used to receive the control signal Vgs3;
  • the circuit two specific embodiments, in consideration of the parasitic capacitance C, and Dow switch QOW Dow drain and source of the diode anode and cathode electrodes of the parasitic capacitance C Qow, and some rules of conservation of charge by charge transfer, if When C Dow and C Qow meet a certain relationship, the switch Qow can realize zero-voltage turn-on. Specifically, C Dow and C Qow meet:
  • circuit 3 of the unidirectional clamping network Sow includes a switch tube Qow1 and a switch tube Qow2.
  • the source of the switch tube Qow1 is connected to the source of the switch tube Qow2, and the drain of the switch tube Qow1
  • the pole is used as the anode of the one-way clamping network Sow
  • the drain of the switch Qow2 is used as the cathode of the one-way clamping network Sow
  • the grid of the switch Qow1 and the grid of the switch Qow2 are used to receive the control signal Vgs3.
  • the specific implementation circuit 4 of the unidirectional clamping network Sow includes a diode Dow, a switch Qow and a capacitor Cow.
  • the cathode of the diode Dow is connected to one end of the capacitor Cow and the drain of the switch Qow.
  • the anode of the diode Dow is connected to the other end of the capacitor Cow as the anode of the one-way clamping network Sow
  • the source of the switch Qow is the cathode of the one-way clamping network Sow
  • the gate of the switch Qow is used to receive the control signal Vgs3 .
  • circuit 5 of the unidirectional clamping network Sow contains a diode Dow, a switch Qow and a capacitor Cow.
  • the anode of the diode Dow is connected to one end of the capacitor Cow and the source of the switch Qow.
  • the cathode of the diode Dow is connected to the other end of the capacitor Cow as the cathode of the unidirectional clamping network Sow, the drain of the switch Qow is used as the anode of the unidirectional clamping network Sow, and the gate of the switch Qow is used to receive the control signal Vgs3 .
  • the specific implementation circuit 6 of the unidirectional clamping network Sow includes the diode Dow, the switch Qow, the capacitor Cow, the voltage regulator Zow, the cathode of the diode Dow, one end of the capacitor Cow and the switch Qow
  • the anode of the diode Dow is connected to the anode of the Zener tube Zow, as the anode of the unidirectional clamp network Sow
  • the cathode of the Zener tube Zow is connected to the other end of the capacitor Cow
  • the source of the switch tube Qow is used as The cathode of the unidirectional clamp network Sow and the gate of the switch Qow are used to receive the control signal Vgs3.
  • the seventh specific implementation circuit of the unidirectional clamping network Sow includes the diode Dow, the switch Qow, the capacitor Cow, the voltage regulator Zow, the anode of the diode Dow, one end of the capacitor Cow and the switch Qow
  • the cathode of the diode Dow is connected to the cathode of the Zener tube Zow, as the cathode of the unidirectional clamp network Sow
  • the anode of the Zener tube Zow is connected to the other end of the capacitor Cow
  • the drain of the switch tube Qow is used as The anode of the unidirectional clamp network Sow and the gate of the switch Qow are used to receive the control signal Vgs3.

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Abstract

本发明公开了一种不对称半桥变换器及控制方法，通过增加一个与变压器原边、副边或第三绕组并联的单向钳位网络，在励磁电感电流达到设定值时，控制辅开关关断、单向钳位网络导通，钳位电流流过单向钳位网络，单向钳位网络钳位并维持钳位电流保持基本不变，在主开关导通前的一段时间，控制单向钳位网络关断，释放此钳位电流，使主开关两端电压降低至零或接近零，实现主开关的零电压开通。本发明能实现对励磁电感电流负向峰值的有效控制，降低变换器轻空载下功率器件的电流有效值，在保留现有技术方案能够实现零电压开通的优势的情况下，大幅提升变换器轻载效率，降低空载损耗，且控制实现上简单高效。

Description

不对称半桥变换器及控制方法 技术领域
本发明涉及开关变换器，特别涉及不对称半桥变换器及其控制方法。
背景技术
从20世纪60年代开始得到发展和应用的开关变换器多使用硬开关技术。但是随着科学技术的发展，各行各业对电源提出了更高地要求，高效率、高功率密度、小型化等成为电源行业研究的主要课题。采用硬开关技术的开关变换器存在开关损耗大、效率低、开关频率不高、EMI差等缺点，为此出现了软开关技术，所谓软开关是指：零电压开关(Zero-Voltage-Switching)，简称ZVS；零电流开关(Zero-Current-Switching)，简称ZCS。软开关技术主要利用谐振原理，使开关变换器的开关器件中的电流(或电压)按正弦或准正弦规律变化，当开关器件电流自然过零时，开关关断，或开关器件电压为零时，开关开通，从而使开关器件损耗为零。因而有可能使得开关变换器的开关频率提高到兆赫兹级水平，为开关电源的高效率、高功率密度及小型化提供了可能。
Vicor公司提出了一种有源钳位正激拓扑，专利号US5805434，如图1所示，在正激电路的变压器原边绕组两端增加钳位电路330，辅开关332与主开关20驱动信号互补，主开关20关断时，钳位电容334吸收漏感能量并使电流反向，二极管350为反向电流提供续流回路，实现近似无损续流，抑制钳位电容334与励磁电感、漏感谐振。钳位电路330的主要目的是漏感能量回收与抑制谐振、并优化EMI，钳位电路并没有实现主开关零电压开通的效果。
Vicor公司还提出了一种双钳位ZVS BUCK-BOOST拓扑，专利号US7561446，如图2-1、图2-2所示，副边电流到零后开关Q3关断、开关Q4导通，变压器被钳位至零电压，变压器剩余电流在钳位电路续流，在开关Q1导通前关断开关Q2，变压器剩余电流使开关Q1实现零电压开通。此技术强调副边电流到零开关Q3关断、开关Q4导通，且电路连接方式上与不对称半桥反激变换器存在较大差别。
Astec公司提出了一种有源钳位反激拓扑，专利号US9973098，如图3所示，钳位阶段钳位电容C2能量全部释放至变压器，变压器被二极管D2钳位至零电压，漏感电流通过二极管D2续流，在主开关管Q1导通前关断钳位开关管Q2，漏感电流实现主开关管Q1的零电压开通。钳位电路回收漏感能量，在主开关管Q1导 通前释放回收的漏感能量，实现主管零电压开通。此技术强调回收漏感能量并实现零电压开通，且电路连接方式上与不对称半桥反激变换器也存在较大差别。
不对称半桥反激变换器可利用变压器的激磁电感、漏感，实现全输入/全负载范围内开关器件的ZVS、ZCS，这为开关电源产品的效率提升、体积减小、制造工艺简化、EMI改善等带来了可能。
图4所示为现有不对称半桥反激变换器电路图，包含输入电容Cin、主开关S M和辅开关S A、谐振电容Cr、变压器Tr、整流开关S D、输出滤波电容Co。变压器Tr包含原边绕组和副边绕组，变压器Tr原边绕组的阳极与副边绕组的一端互为同名端，变压器Tr原边绕组的阳极与副边绕组的另一端互为异名端；谐振电容Cr与变压器Tr组成谐振能量传输网络，所述谐振能量传输网络包含两个输入端和两个输出端；主开关S M的一端和辅开关S A的一端连接，组成半桥结构，相连的节点记为开关节点SW，主开关S M的另一端与输入电容Cin的一端、输入正+Vin相连，辅开关S A的另一端与输入电容Cin的另一端、输入负-Vin相连；谐振电容Cr的一端与变压器Tr原边绕组的阳极相连，谐振电容Cr的另一端作为谐振能量传输网络的一个输入端，接于开关节点SW处，变压器Tr原边绕组的阴极作为谐振能量传输网络的另一个输入端，与输入负-Vin相连；变压器Tr副边绕组的异名端作为谐振能量传输网络的一个输出端，与整流开关S D的一端相连，整流开关S D的另一端与输出滤波电容Co的一端相连，作为输出正+Vo，变压器Tr副边绕组的同名端作为谐振能量传输网络的另一个输出端，与输出滤波电容Co的另一端相连，作为输出负-Vo。
考虑开关器件寄生参数、变压器寄生参数，即可得出现有不对称半桥反激变换器的等效电路原理图，如图5所示，与图4不同之处在于：变压器Tr等效为漏感Lr、励磁电感Lm以及理想变压器T的组合，漏感Lr作为谐振能量传输网络的谐振电感(本领域的技术人员也常用外置电感与谐振电容Cr进行谐振)，记为谐振电感Lr，谐振电感Lr的一端与励磁电感Lm一端、理想变压器T的原边绕组的一端相连，谐振电感Lr的另一端作为变压器Tr的原边绕组的一端，励磁电感Lm的另一端与理想变压器T的原边绕组的另一端相连，作为变压器Tr的原边绕组的另一端；主开关S M的两端并联有二极管D1和电容C1，具体地，二极管D1的阴极与电容C1的一端、主开关S M的一端相连，接于输入正+Vin，二极管D1的阳极与电容C1的另一端、主开关S M的的另一端相连，接于开关节点SW处；辅开关 S A的两端并联有二极管D2和电容C2，具体地，二极管D2的阴极与电容C2的一端、辅开关S A的一端相连，接于开关节点SW处，二极管D2的阳极与电容C2的另一端、辅开关S A的另一端相连，接于输入负-Vin。
图6为现有不对称半桥反激变换器工作于CCM模式的典型工作波形图，现有不对称半桥反激变换器工作于CCM模式，每个循环周期包含四个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，主开关零电压开通阶段。现结合图5对所述四个阶段进行说明，具体如下：
励磁阶段，从t0时刻起至t1时刻止，t0时刻主开关S M受控制信号Vgs1作用导通，辅开关S A受控制信号Vgs2作用处于关断状态(Vgs1为主开关S M的控制信号、Vgs2为辅开关S A的控制信号)，输入电压Vin向谐振电容Cr、谐振电感Lr和励磁电感Lm充电，励磁电感电流I Lm和谐振电感电流I Lr线性上升，至t1时刻，Vgs1由高电平变为低电平，控制主开关S M关断，此过程中整流开关S D处于关断状态，变压器Tr不对外传递能量，输入电能存储在变压器Tr中；
辅开关零电压开通阶段，从t1时刻起至t2时刻止，t1时刻主开关S M关断，电容C1、电容C2、谐振电感Lr和励磁电感Lm形成串联谐振，谐振电感电流I Lr给电容C1充电、电容C2放电，使得电容C1两端的电压V C1上升、电容C2两端的电压V C2下降，至V C2降至与V Cr电压相同，励磁电感Lm两端电压为零，电容C2两端的电压V C2继续下降，励磁电感Lm极性发生变化，导致谐振电感Lr两端承受负向电压，谐振电感电流I Lr依然为正，但谐振电感电流I Lr开始负向增大，谐振电感电流I Lr持续给电容C1充电、电容C2放电，至电容C2放电完毕，V C2降为零，二极管D2自然导通，谐振电感电流I Lr流过二极管D2，与此同时，由于输出电压的反射，励磁电感Lm两端的电压被钳位至NV o(N为变压器匝比)，副边整流开关S D导通，谐振电感Lr与谐振电容Cr发生谐振，存储在变压器Tr中的电能开始向副边转移，至t2时刻，Vgs2为高电平，控制辅开关S A导通，辅开关S A实现零电压开通；
去磁阶段，从t2时刻起至t3时刻止，t2时刻辅开关S A导通，主开关S M处于关断状态，副边整流开关S D导通，整流开关S D的电流I D增加，励磁电感Lm两端的电压被钳位，电压上负下正，励磁电感电流I Lm线性减小，同时，谐振电感Lr与谐振电容Cr发生谐振，谐振电感电流I Lr快速负向增大，谐振电感电流I Lr由正 变为零，并很快从零变为负，整流开关S D仍然导通，变压器Tr继续释放电能，至t3时刻，Vgs2由高电平变为低电平，控制辅开关S A关断；
主开关零电压开通阶段，从t3时刻起至t4时刻止，辅开关S A关断，电容C1、电容C2、谐振电容Cr和谐振电感Lr形成串联谐振，谐振电感电流I Lr为负，并迅速正向增大，给电容C1放电、电容C2充电，使得电容C1两端的电压V C1下降、电容C2两端的电压V C2上升，至V C2上升与V Cr电压相同时，励磁电感电流I Lm和谐振电感电流I Lr相等，整流开关S D关断，励磁电感电流I Lm和谐振电感电流I Lr被释放，并给电容C1继续放电、给电容C2继续充电，电容C1两端的电压V C1继续下降，电容C2两端电压V C2继续上升，至电容C1两端电压下降到零，谐振电感电流I Lr开始流过二极管D1，t4时刻，Vgs1为高电平，控制主开关S M导通，主开关S M实现零电压开通；至此，一个循环周期结束。
以上是对现有不对称半桥反激变换器工作于CCM模式的工作过程的说明，现有不对称半桥反激变换器工作于DCM模式与CCM模式的主要差别在：整流开关S D的电流I D在辅开关S A关断时刻能否降到零，若辅开关S A关断时刻整流开关S D的电流I D已经降到零，则为DCM模式，若辅开关S A关断时刻整流开关S D的电流I D不为零，则为CCM模式。现有不对称半桥反激变换器工作于DCM模式的工作原理，此处不再赘述，相关工程技术人员可自行推演得出。
所述现有不对称半桥反激变换器，主开关S M和辅开关S A互补导通，即：主开关S M导通、辅开关S A关断，主开关S M关断、辅开关S A导通；所述现有不对称半桥反激变换器，其每个循环周期的时间相同，即开关频率固定。
需要说明的是对于本领域的技术人员而言，图4中还可以有多种等同变换，包括但不限于如下几种情况及其交换组合：
(1)将谐振电容Cr置于输入负-Vin与变压器Tr原边绕组的另一端之间；
(2)将变压器Tr原边绕组的另一端连接至输入正+Vin；
(3)将整流开关S D置于变压器Tr副边绕组与输出负-Vo连接的线路中。
根据行业目前研究现状，不对称半桥反激变换器存在的主要问题是：
1、轻负载效率低、空载功耗大；
不对称半桥反激变换器，设计若满足满载主开关实现零电压开通，则轻负载和空载时，主开关更容易实现零电压开通，且当负载减轻，由于占空比不变，励磁电感电流I Lm会存在较大的负向电流，此负向电流远远超出变换器主开关实现零 电压开通的需求，过多的电流将在谐振腔内流动，产生较大的损耗，从而导致轻负载效率低、空载功耗大；
2、不适合宽压输入场合；
不对称半桥反激变换器，设计若满足低压满载时，主开关实现零电压开通，则高压满载时，主开关必然实现零电压开通，且高压满载励磁电感电流I Lm会存在较大的负向电流，此负向电流远远超出变换器主开关实现零电压开通的需求，过多的电流将在谐振腔内流动，产生较大的损耗，不利于效率的优化；设计若满足高压满载时，主开关恰好实现零电压开通，则低压满载时，主开关必然无法实现零电压开通，从而会使主开关产生较大的开通损耗。即：不对称半桥反激变换器设计上很难兼顾低压输入与高压输入，对于85VAC～264VAC输入范围的开关电源应用(母线电压变化范围约为120VDC-370VDC)，电路设计将变得十分困难，很难兼顾低压输入与高压输入。
发明内容
有鉴于此，本发明提出一种不对称半桥变换器，能解决现有技术方案轻负载效率低、空载功耗大，以及难以兼顾低压输入与高压输入的问题，本发明同时提出一种不对称半桥变换器控制方法，控制实现上更加简单高效。
本申请的发明构思为：在现有的不对称半桥反激变换器的基础上，增加一个与变压器原边绕组或者副边绕组或者第三绕组两端电联接的单向钳位网络，在一个开关循环周期的特定时刻，励磁电感电流达到设定值时，控制辅开关S A关断，钳位电流流过单向钳位网络，单向钳位网络保持钳位电流基本不变，至主开关S M开通前一段时间释放钳位电流，实现主开关S M零电压开通。具体地，本发明提出的一种不对称半桥反激变换器控制方法，每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段；励磁阶段持续时间、去磁阶段持续时间与输入电压和负载相关，辅开关零电压开通阶段持续时间、主开关零电压开通阶段持续时间与主功率器件设计有关，电流钳位阶段持续时间由循环周期持续时间和励磁阶段持续时间、去磁阶段持续时间、辅开关零电压开通阶段持续时间、主开关零电压开通阶段持续时间共同决定，从而保证每个循环周期持续时间的固定，即定频脉冲宽度调制(PWM)控制。
本发明的不对称半桥变换器技术方案如下：
一种不对称半桥变换器，包含原边电路、变压器Tr和副边电路：原边电路包含输入电容Cin、主开关S M和辅开关S A、谐振电容Cr；变压器Tr包含原边绕组和副边绕组，变压器Tr原边绕组的阳极与副边绕组的一端互为同名端，变压器Tr原边绕组的阳极与副边绕组的另一端互为异名端；副边电路包含整流开关S D和输出电容Co；输入电容Cin一端连接输入正、另一端连接输入负；主开关S M与辅开关S A串联后与输入电容Cin并联；谐振电容Cr与变压器Tr的原边绕组串联，串联后的一端连接主开关S M与辅开关S A的连接点，串联后的另一端连接输入正或输入负；变压器Tr的副边绕组与整流开关S D串联后与输出电容Co并联，输出电容Co一端连接输出正、另一端连接输出负；其特征在于：还包括单向钳位网络，单向钳位网络用于控制励磁电感电流负向峰值，其连接关系为如下情况之一：
(1)单向钳位网络的阳极与变压器原边绕组的阳极电联接，单向钳位网络的阴极与变压器原边绕组阴极电联接；
(2)单向钳位网络的阳极与变压器副边绕组的同名端电联接，单向钳位网络的阴极与变压器副边绕组的异名端电联接；
(3)不对称半桥变换器还包括第三绕组，变压器原边绕组的阳极与第三绕组的一端互为同名端，变压器原边绕组的阳极与第三绕组的另一端互为异名端，单向钳位网络的阳极与第三绕组的同名端电联接，单向钳位网络的阴极与第三绕组的异名端电联接。
优选地，变压器副边绕组的异名端与整流开关S D的一端电联接，整流开关S D的另一端与输出电容Co的一端电联接，作为输出正，变压器副边绕组的同名端与输出电容Co的另一端电联接，作为输出负。此时不对称半桥变换器为不对称半桥反激变换器。
优选地，变压器副边绕组的同名端与整流开关S D的一端电联接，整流开关S D的另一端与输出电容Co的一端电联接，作为输出正，变压器副边绕组的异名端与输出电容Co的另一端电联接，作为输出负。此时不对称半桥变换器为不对称半桥正激变换器。
当不对称半桥变换器包括第三绕组时，第三绕组为一个独立绕组。
当不对称半桥变换器包括第三绕组时，第三绕组与辅助绕组为同一绕组。
作为单向钳位网络的第一种具体实施方式，其特征在于：包括一只二极管和一只开关管，连接关系为以下两种之一：
(1)二极管的阳极为单向钳位网络的阳极，二极管的阴极连接开关管的漏极，开关管的源极为单向钳位网络的阴极；
(2)开关管的漏极为单向钳位网络的阳极，开关管的源极连接二极管的阳极，二极管的阴极为单向钳位网络的阴极。
优选地，上述单向钳位网络的第一种具体实施方式中，二极管阳极与阴极的寄生电容容值为C Dow、开关管漏极与源极的寄生电容容值为C Qow、输入电压为V in、输出电压为V o、原边绕组和副边绕组或者第三绕组的匝比为N，各参数满足如下关系式：
Figure PCTCN2019113679-appb-000001
作为单向钳位网络的第二种具体实施方式，其特征在于：单向钳位网络包括两只开关管，两只开关管的源极连接，其中一只开关管的漏极为单向钳位网络的阳极，另一只开关管的漏极为单向钳位网络的阴极。
作为单向钳位网络的第三种具体实施方式，其特征在于：单向钳位网络包括一只二极管、一只开关管和一只电容，连接关系为以下两种之一：
(1)二极管的阳极与电容一端相连，为单向钳位网络的阳极，二极管的阴极与电容另一端、开关管的漏极相连，开关管的源极为单向钳位网络的阴极；
(2)二极管的阴极与电容一端相连，为单向钳位网络的阴极，二极管的阳极与电容另一端、开关管的源极相连，开关管的漏极为单向钳位网络的阳极。
优选地，上述单向钳位网络的第三种具体实施方式中，二极管阳极与阴极的寄生电容容值为C Dow、开关管漏极与源极的寄生电容容值为C Qow、电容容值为C ow、输入电压为V in、输出电压为V o、原边绕组和副边绕组或者第三绕组的匝比为N，各参数满足如下关系式：
Figure PCTCN2019113679-appb-000002
作为单向钳位网络的第四种具体实施方式，其特征在于：单向钳位网络包括一只二极管、一只开关管、一只稳压管和一只电容，连接关系为以下两种之一：
(1)二极管的阳极与稳压管的阳极相连，为单向钳位网络的阳极，二极管的阴极与电容一端、开关管的漏极相连，稳压管的阴极与电容的另一端相连，开关管的源极为单向钳位网络的阴极；
(2)二极管的阴极与稳压管的阴极相连，为单向钳位网络的阴极，二极管的阳极与电容一端、开关管的源极相连，稳压管的阳极与电容的另一端相连，开关管的漏极为单向钳位网络的阳极。
对应地，本发明还提供上述不对称半桥变换器的控制方法，技术方案如下：
一种不对称半桥变换器的控制方法，其特征在于：每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段；
在励磁阶段、辅开关零电压开通阶段，单向钳位网络关断；
在去磁阶段，辅开关导通，单向钳位网络导通或者关断均可，单向钳位网络没有电流流过；至此阶段结束时刻，励磁电感电流达到设定值，辅开关关断，单向钳位网络处于导通状态，钳位电流流过单向钳位网络；
在电流钳位阶段，单向钳位网络导通，钳位电流流过单向钳位网络，单向钳位网络保持钳位电流基本不变，至此阶段结束时刻，单向钳位网络关断；
在主开关零电压开通阶段，单向钳位网络已被关断，单向钳位网络中的钳位电流被释放，使主开关电压降低至零或接近零，此时控制主开关导通，实现主开关零电压开通。
优选方案之一，励磁电感电流设定值为负值或零、且与输入电压相关。
优选方案之二，每个开关循环周期的持续时间相同。
作为上述优选方案之二，进一步地，控制使用脉冲宽度调制控制方式。
作为上述优选方案之二，进一步地，开关循环周期的中心开关频率固定，实际开关频率以中心开关频率为中心，在设定的下限频率、设定的上限频率之间周期性变化。
术语解释：
单向钳位网络的阳极：直流电流从单向钳位网络向内流入的一端为阳极；
单向钳位网络的阴极：直流电流从单向钳位网络向外流出的一端为阴极；
钳位电流：单向钳位网络中流过的电流，具体指，在去磁阶段，励磁电感电流达到设定值时，控制辅开关S A关断，流过单向钳位网络的励磁电感电流或者经过变压器耦合到副边绕组的电流或者经过变压器耦合到第三绕组的电流；
原边绕组的阳极：励磁阶段，主管S M导通，直流电流从原边绕组向内流入的一端为原边绕组的阳极；
原边绕组的阴极：励磁阶段，主管S M导通，直流电流从原边绕组向外流出的一端为原边绕组的阴极；
励磁电感电流正方向：自变压器原边绕组的阳极向内流入，流经励磁电感，从原边绕组的阴极向外流出；
同名端：指在同一交变磁通的作用下任一时刻变压器两个绕组中都具有相同电势极性的端头彼此互为同名端；
异名端：指在同一交变磁通的作用下任一时刻变压器两个绕组中都具有相反电势极性的端头彼此互为异名端；
电联接：代表的含义除了直接连接，还包括间接连接(即两个电联接对象之间还可以连接其它的元器件)，并且包括通过感应耦合等方式。
本发明技术方案能够实现对变压器原边励磁电感电流负向峰值的有效控制，进而达到对实现零电压开通所需的励磁电感负向电流控制的目的，降低谐振腔内流动的电流，降低变换器轻空载下功率器件的电流有效值，从而在保留现有技术方案能够实现零电压开通的优势的情况下，大幅提升变换器轻载效率，并降低空载损耗，使得不对称半桥反激变换器能够更好地兼顾低压输入与高压输入，实现宽输入的要求。并且该发明构思也可以用于不对称半桥正激变换器。
本发明的优点显而易见，有益效果如下：
1、增加单向钳位网络，实现对励磁电感电流负向峰值的有效控制，降低变换器轻空载下功率器件的电流有效值，提高轻载效率、降低空载功耗；
2、增加单向钳位网络，解决随着输入电压的升高，轻负载效率严重劣化的问题，使得电路设计上，更容易实现兼顾低压输入与高压输入；
3、定频脉冲宽度调制(PWM)控制方式，控制实现上更加简单高效。
附图说明
图1为Vicor公司专利US5805434说明书附图10；
图2-1为Vicor公司专利US7561446说明书附图1；
图2-2为Vicor公司专利US7561446说明书附图3；
图3为Astec公司专利US9973098说明书附图2；
图4为现有不对称半桥反激变换器电路图；
图5为现有不对称半桥反激变换器等效电路原理图；
图6为现有不对称半桥反激变换器CCM模式工作波形图；
图7为本发明第一实施例不对称半桥反激变换器电路图；
图8为本发明第一实施例不对称半桥反激变换器等效电路原理图；
图9为本发明第一实施例不对称半桥反激变换器CCM模式工作波形；
图10为本发明第一实施例不对称半桥反激变换器DCM模式工作波形；
图11-1为现有方案与本发明第一实施例方案在120V输入下效率曲线图；
图11-2为现有方案与本发明第一实施例方案在160V输入下效率曲线图；
图11-3为现有方案与本发明第一实施例方案在320V输入下效率曲线图；
图11-4为现有方案与本发明第一实施例方案在370V输入下效率曲线图；
图12为本发明第二实施例不对称半桥反激变换器电路图；
图13为本发明第三实施例不对称半桥反激变换器电路图；
图14为本发明第四实施例不对称半桥反激变换器电路图；
图15为本发明第五实施例不对称半桥正激变换器电路图；
图16为本发明第五实施例不对称半桥正激变换器等效电路原理图；
图17为本发明第五实施例不对称半桥正激变换器典型工作波形；
图18为本发明第六实施例不对称半桥正激变换器电路图；
图19为本发明第七实施例不对称半桥正激变换器电路图；
图20为本发明第八实施例不对称半桥正激变换器电路图；
图21-1至21-7为本发明第一至第八实施例单向钳位网络的具体实施电路。
具体实施方式
为了使本发明更加清楚明白，以下将结合附图及具体实施例，对现有技术方案及本发明技术方案进行更加清楚、完整地描述。
第一实施例
图7所示为本发明第一实施例电路图、图8为本发明第一实施例不对称半桥反激变换器等效电路原理图，图7与图4、图8与图5的不同之处在于：在变压 器的原边增加一个单向钳位网络Sow，单向钳位网络Sow的阳极与变压器原边绕组的阳极电联接，单向钳位网络Sow的阴极与变压器原边绕组阴极电联接；
图9为本发明第一实施例的不对称半桥反激变换器工作于CCM模式的典型工作波形图，每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段。现结合图8对每个循环周期(从t0时刻起至t5时刻止，记为T)所述的五个阶段进行说明，具体如下：
励磁阶段，从t0时刻起至t1时刻止(记为T0)，控制主开关S M导通，输入电压Vin向谐振电容Cr、谐振电感Lr和励磁电感Lm充电，励磁电感电流I Lm和谐振电感电流I Lr线性上升，即输入电压Vin给变压器励磁。此阶段控制信号Vgs3为低电平，单向钳位网络Sow关断，不起作用，电路工作情况与现有技术相同，在此不赘述，因此控制信号Vgs1、控制信号Vgs2、谐振电感电流I Lr、主开关S M的漏源极电压Vds1、整流开关S D的电流I D波形与现有技术图6的波形相同；
辅开关零电压开通阶段，从t1时刻起至t2时刻止(记为T1)，控制主开关S M关断，电容C1、电容C2、谐振电感Lr和励磁电感Lm形成串联谐振，谐振电感电流I Lr给电容C1充电、电容C2放电，使得电容C1两端的电压V C1上升、电容C2两端的电压V C2下降，至电容C2放电完毕，V C2降为零，二极管D2自然导通，谐振电感电流I Lr流过二极管D2，t2时刻，控制辅开关S A导通，辅开关S A实现零电压开通。此阶段控制信号Vgs3仍然为低电平，单向钳位网络Sow关断，不起作用，电路工作情况还是与现有技术相同，在此同样不赘述，因此控制信号Vgs1、控制信号Vgs2、谐振电感电流I Lr、主开关S M的漏源极电压Vds1、整流开关S D的电流I D波形与现有技术图6的波形也相同；
去磁阶段，从t2时刻起至t3时刻止(记为T2)，控制辅开关S A导通，主开关S M继续关断，整流开关S D导通，整流开关S D的电流I D增加，励磁电感Lm两端的电压被钳位，电压上负下正，励磁电感电流I Lm线性减小，变压器去磁，t3时刻，励磁电感电流达到设定值时，控制辅开关S A关断。与现有技术不同之处在于，此阶段控制信号Vgs3为高电平，增加的单向钳位网络Sow导通，单向钳位网络Sow的开通时刻可以是t2至t3之间的任意时刻(即t2至t3之间单向钳位网络Sow导通与关断均可)，由于单向钳位网络Sow只允许电流从其阳极流到阴极，故此过程中单向钳位网络Sow中并没有电流流过，此阶段控制信号Vgs1、控制信 号Vgs2、谐振电感电流I Lr、主开关S M的漏源极电压Vds1、整流开关S D的电流I D波形与现有技术图6的波形也相同；
电流钳位阶段，从t3时刻起至t4时刻止(记为T3)，t3时刻，辅开关S A关断，单向钳位网络Sow继续导通，电容C1、电容C2、谐振电容Cr和谐振电感Lr形成串联谐振，谐振电感电流I Lr为负，并迅速正向增大，给电容C1放电、电容C2充电，使得电容C1两端的电压V C1下降、电容C2两端的电压V C2上升，至V C2上升与V Cr电压相同时，单向钳位网络Sow阳极电压为零，励磁电感电流I Lm和谐振电感电流I Lr相等，整流开关S D关断，励磁电感电流I Lm(或称钳位电流)自然通过单向钳位网络Sow阳极流向阴极，单向钳位网络Sow保持钳位电流基本不变，至t4时刻，控制信号Vgs3变为低电平，单向钳位网络Sow关断；
主开关零电压开通阶段，从t4时刻起至t5时刻止(记为T4)，t4时刻单向钳位网络Sow关断，主开关S M和辅开关S A保持关断状态，单向钳位网络Sow钳位并维持的钳位电流被释放，并给电容C1继续放电、给电容C2继续充电，电容C1两端的电压V C1继续下降，电容C2两端电压V C2继续上升，至电容C1两端电压下降到零，钳位电流开始流过二极管D1，t5时刻，控制信号Vgs1变为高电平，主开关S M导通，主开关S M实现零电压开通；
至此，本发明第一实施例一个循环周期结束。
在主功率器件设计确定的情况下，所述每个循环周期的五个阶段中，励磁阶段持续时间T0、去磁阶段持续时间T2与输入电压和负载相关，辅开关零电压开通阶段持续时间T1、主开关零电压开通阶段持续时间T4与主功率器件设计有关，电流钳位阶段持续时间T3由T和T0、T2、T1以及T4共同决定，即T3随着输入电压和负载变化而变化，从而保证每个循环周期持续时间T的固定，即实现定频脉冲宽度调制(PWM)控制。
图10所示为本发明第一实施例，一种不对称半桥反激变换器工作于DCM模式的典型工作波形图，所述一种不对称半桥反激变换器每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段。与上述CCM模式的工作过程的主要差别在：输出整流开关S D的电流I D在辅开关S A关断时刻能否降到零，若辅开关S A关断时刻输出整流开关S D的电流I D已经降到零，则为DCM模式，若辅开关S A关断时刻输出整流开关S D的电 流I D不为零，则为CCM模式。对于本发明第一实施例工作于DCM模式的工作原理，此处不再赘述，相关工程技术人员可自行推演得出。
对本发明第一实施例技术方案与现有技术方案对比，按照表1所列输入输出规格，进行合理设计及优化，通过计算机仿真手段，得到本发明第一实施例技术方案与现有技术方案在不同输入、不同负载状态下的占空比D、励磁电感峰值电流I Lm_peak正峰值(+Max)/负峰值(-Max)、主开关S M和辅开关S A电流有效值Is_rms、励磁电感电流有效值I Lm_rms的对比数据，如表2所列。
表1
输入电压范围 85VAC-264VAC(母线电压范围约为120VDC-370VDC)
输出规格 Vo＝12V、Io＝5A、Po＝60W
开关频率 f＝100KHz
表2
Figure PCTCN2019113679-appb-000003
Figure PCTCN2019113679-appb-000004
从表2数据可以看出：现有方案负载变化过程中，占空比D几乎不发生变化，其结果就是励磁电感峰值电流I Lm_peak会有一个较大的负向峰值，本发明方案负载变化过程中，占空比D随负载减小而减小，励磁电感峰值电流I Lm_peak的正向峰值和负向峰值都有一定程度的降低，励磁电感电流有效值I Lm_rms也大幅减小；现有方案轻负载下，励磁电感峰值、有效值较大，导致轻负载效率低，且随着输入电压的升高这种情况愈加严重，导致高压输入轻负载效率严重劣化，本发明方案在一定程度改善了这个问题。
图11-1、11-2、11-3、11-4所示为现有方案与本发明第一实施例方案分别在120V、160V、320V、370V输入下效率曲线图，可以明显看出，在不同输入电压下，本发明技术方案的轻载效率相较于现有方案都有非常明显的提高。
为了进一步优化EMI，可在本发明技术方案的基础上增加抖频功能，即：开关循环周期的中心开关频率固定，实际开关频率以中心开关频率为中心，在设定的下限频率、设定的上限频率之间周期性变化。
第二实施例
图12为本发明第二实施例电路图，一种不对称半桥反激变换器，本发明第二实施例与第一实施例的主要差别在于单向钳位网络Sow连接方式的不同：第一实施例中在变压器的原边增加一个单向钳位网络Sow，单向钳位网络Sow的阳极与变压器原边绕组的阳极电联接，单向钳位网络Sow的阴极与变压器原边绕组的阴极电联接；第二实施例中在变压器的副边增加一个单向钳位网络Sow，单向钳位网络Sow的阳极与变压器副边绕组同名端电联接，单向钳位网络Sow的阴极与变压器副边绕组异名端电联接。虽然连接方式不同，但是根据变压器原边绕组与 副边绕组彼此耦合的原理可知：通过单向钳位网络Sow将副边绕组接通，单向钳位网络Sow流过钳位电流，同样可以实现将励磁电感电流I Lm钳位和维持的效果。
本发明第二实施例，一种不对称半桥反激变换器每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段。与第一实施例所述不对称半桥反激变换器工作过程的主要差别在：电流钳位阶段，单向钳位网络Sow导通的时刻不同。第一实施例中单向钳位网络Sow可以在t2至t3之间的任意时刻导通，对于第二实施例，单向钳位网络Sow只能在t3时刻导通。本发明第二实施例具体工作原理细节，此处不再赘述，本邻域技术人员可根据第一实施例工作原理推演得出。
第三实施例
图13为本发明第三实施例电路图，一种不对称半桥反激变换器，与第一实施例不同之处在于，变压器Tr还包含第三绕组Np_ow；第一实施例中在变压器的原边增加一个单向钳位网络Sow，单向钳位网络Sow的阳极与变压器原边绕组的阳极电联接，单向钳位网络Sow的阴极与变压器原边绕组的阴极电联接；第三实施例中在变压器的第三绕组Np_ow增加一个单向钳位网络Sow，单向钳位网络Sow的阳极与第三绕组Np_ow同名端电联接，单向钳位网络Sow的阴极与第三绕组Np_ow异名端电联接。虽然连接方式不同，但是根据变压器原边绕组与第三绕组彼此耦合的原理可知：通过单向钳位网络Sow将第三绕组Np_ow接通，单向钳位网络Sow流过钳位电流，同样可以实现将励磁电感电流I Lm钳位和维持的效果。
本发明第三实施例，一种不对称半桥反激变换器每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段。与第一实施例所述不对称半桥反激变换器工作过程的主要差别在：电流钳位阶段，单向钳位网络Sow导通的时刻不同。第一实施例中单向钳位网络Sow可以在t2至t3之间的任意时刻导通，对于第三实施例，单向钳位网络Sow只能在t3时刻导通。本发明第三实施例具体工作原理细节，此处不再赘述，本邻域技术人员可根据第一实施例工作原理推演得出。
实际变换器在应用中往往需要增加相应的辅助供电电路，行业通行的做法是给变压器增加辅助绕组，将辅助绕组的输出经过整流滤波后输出，用于给控制、 驱动等供电。应当明确，本发明第三实施例所述第三绕组Np_ow除了与单向钳位网络电联接，用于单向钳位网络钳位并维持钳位电流的同时，也可以作为辅助供电电路的辅助绕组使用。本发明第三实施例所述第三绕组Np_ow可以独立使用，即作为一个独立绕组；同时，第三绕组Np_ow也可作为辅助绕组使用，即第三绕组Np_ow与辅助绕组为同一绕组。
第四实施例
图14所示为本发明第四实施例电路图，第四实施例与第一实施例的主要差别在于谐振能量传输网络连接方式的不同，第一实施例中谐振能量传输网络的一个输入端与开关节点SW相连，谐振能量传输网络的另一个输入端与输入负-Vin相连，第四实施例中谐振能量传输网络的一个输入端与开关节点SW相连，谐振能量传输网络的另一个输入端与输入正+Vin相连。
本发明第四实施例，一种不对称半桥反激变换器每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段。这与第一实施例所述不对称半桥反激变换器的主要差别还在在：主开关S M和辅开关S A在半桥结构上的上下位置的不同。本发明第四实施例具体工作原理细节，此处不再赘述，本邻域技术人员可根据第一实施例工作原理推演得出。
第五实施例
图15为本发明第五实施例不对称半桥正激变换器电路图；图16为本发明第五实施例不对称半桥正激变换器等效电路原理图，本实施例在于将本发明第一实施例技术方案的思想应用于不对称半桥正激变换器，即可经过推演得出本发明第五实施例技术方案，因此图15与图7、图16与图8的区别在于表示同名端关系的黑点位置有所不同。
图17为本发明第五实施例，一种不对称半桥正激变换器的典型工作波形图，所述一种不对称半桥正激变换器每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段。本发明第五实施例具体工作原理细节，此处不再赘述，本领域技术人员可根据第一实施例工作原理结合图16推演得出。
第六实施例
图18为本发明第六实施例电路图，与图15的区别在于与变压器的副边增加一个单向钳位网络Sow，单向钳位网络Sow的阳极与变压器副边绕组同名端电联接，单向钳位网络Sow的阴极与变压器副边绕组异名端电联接。图18与图12的区别在于表示同名端关系的黑点位置有所不同。
本发明第六实施例，一种不对称半桥正激变换器每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段。本发明第六实施例具体工作原理细节，此处不再赘述，本邻域技术人员可根据第五实施例和第二实施例工作原理推演得出。
第七实施例
图19为本发明第七实施例电路图，与图15的区别在于变压器Tr还包含第三绕组Np_ow，变压器的第三绕组Np_ow增加一个单向钳位网络Sow，单向钳位网络Sow的阳极与变压器第三绕组Np_ow同名端电联接，单向钳位网络Sow的阴极与变压器第三绕组Np_ow异名端电联接。图19与图13的区别在于表示同名端关系的黑点位置有所不同。
本发明第七实施例，一种不对称半桥正激变换器每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段。本发明第七实施例具体工作原理细节，此处不再赘述，本邻域技术人员可根据第五实施例和第三实施例工作原理推演得出。
第八实施例
图20所示为本发明第八实施例电路图，与图15的区别在于谐振能量传输网络的另一个输入端与输入正+Vin相连。图20与图14的区别在于表示同名端关系的黑点位置有所不同。
本发明第八实施例，一种不对称半桥正激变换器每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段。本发明第八实施例具体工作原理细节，此处不再赘述，本邻域技术人员可根据第五实施例和第四实施例工作原理推演得出。
图21-1至21-7所示为第一至第八实施例单向钳位网络Sow的具体实施电路，通过二极管Dow、稳压管Zow、电容Cow、开关管Qow、开关管Qow1和开关管Qow2的不同组合均可实现单向钳位网络Sow的效果。
作为单向钳位网络Sow的具体实施电路一，如图21-1所示，包含二极管Dow、开关管Qow，二极管Dow的阴极与开关管Qow的漏极相连，二极管Dow的阳极作为单向钳位网络Sow的阳极，开关管Qow的源极作为单向钳位网络Sow的阴极，开关管Qow的栅极用于接收控制信号Vgs3。
作为单向钳位网络Sow的具体实施电路二，如图21-2所示，包含二极管Dow、开关管Qow，二极管Dow的阳极与开关管Qow的源极相连，二极管Dow的阴极作为单向钳位网络Sow的阴极，开关管Qow的漏极作为单向钳位网络Sow的阳极，开关管Qow的栅极用于接收控制信号Vgs3；
在具体实施电路一、具体实施电路二中，考虑二极管Dow阳极与阴极的寄生电容C Dow和开关管Qow漏极与源极的寄生电容C Qow，通过电荷转移的一些规律及电荷守恒定律，若C Dow与C Qow满足一定关系时，开关管Qow可以实现零电压开通，具体的，C Dow与C Qow满足：
C Dow×V Cr>C Qow×(V in-V Cr)
考虑输出电压V o与V Cr和匝比N(N＝N P/N S)的关系，C Dow与C Qow满足：
Figure PCTCN2019113679-appb-000005
作为单向钳位网络Sow的具体实施电路三，如图21-3所示，包含开关管Qow1、开关管Qow2，开关管Qow1的源极与开关管Qow2的源极相连，开关管Qow1的漏极作为单向钳位网络Sow的阳极，开关管Qow2的漏极作为单向钳位网络Sow的阴极，开关管Qow1的栅极、开关管Qow2的栅极用于接收控制信号Vgs3。
作为单向钳位网络Sow的具体实施电路四，如图21-4所示，包含二极管Dow、开关管Qow和电容Cow，二极管Dow的阴极与电容Cow的一端、开关管Qow的漏极相连，二极管Dow的阳极与电容Cow的另一端相连，作为单向钳位网络Sow的阳极，开关管Qow的源极作为单向钳位网络Sow的阴极，开关管Qow的栅极用于接收控制信号Vgs3。
作为单向钳位网络Sow的具体实施电路五，如图21-5所示，包含二极管Dow、开关管Qow和电容Cow，二极管Dow的阳极与电容Cow的一端、开关管Qow的源极相连，二极管Dow的阴极与电容Cow的另一端相连，作为单向钳位网络 Sow的阴极，开关管Qow的漏极作为单向钳位网络Sow的阳极，开关管Qow的栅极用于接收控制信号Vgs3。
在具体实施电路四、具体实施电路五中，考虑二极管Dow阳极与阴极的寄生二极管C Dow和开关管Qow漏极与源极的寄生电容C Qow，通过电荷转移的一些规律及电荷守恒定律，若C ow与C Dow、C Qow满足一定关系时，开关管Qow可以实现零电压开通，具体的，考虑输出电压V o与匝比N(N＝N P/N S，N P表示原边绕组或第三绕组匝数)的关系，C ow满足：
Figure PCTCN2019113679-appb-000006
作为单向钳位网络Sow的具体实施电路六，如图21-6所示，包含二极管Dow、开关管Qow、电容Cow、稳压管Zow，二极管Dow的阴极、电容Cow的一端与开关管Qow的漏极相连，二极管Dow的阳极与稳压管Zow的阳极相连，作为单向钳位网络Sow的阳极，稳压管Zow的阴极与与电容Cow的另一端相连，开关管Qow的源极作为单向钳位网络Sow的阴极，开关管Qow的栅极用于接收控制信号Vgs3。
作为单向钳位网络Sow的具体实施电路七，如图21-7所示，包含二极管Dow、开关管Qow、电容Cow、稳压管Zow，二极管Dow的阳极、电容Cow的一端与开关管Qow的源极相连，二极管Dow的阴极与稳压管Zow的阴极相连，作为单向钳位网络Sow的阴极，稳压管Zow的阳极与与电容Cow的另一端相连，开关管Qow的漏极作为单向钳位网络Sow的阳极，开关管Qow的栅极用于接收控制信号Vgs3。
以上仅是本发明的优选实施方式，应当指出的是，上述优选实施方式及单向钳位网络Sow优选的具体实施电路不应视为对本发明的限制，对于本技术领域的普通技术人员来说，在不脱离本发明的精神和范围内，还可以做出若干改进和润饰，如将谐振电容Cr位置改变、通过简单的串联或者并联、改变单向钳位网络Sow电路组合方式等，这些改进和润饰也应视为本发明的保护范围，这里不再用实施例赘述，本发明的保护范围应当以权利要求所限定的范围为准。

Claims (16)

  1. 一种不对称半桥变换器，包含原边电路、变压器Tr和副边电路：原边电路包含输入电容Cin、主开关S M和辅开关S A、谐振电容Cr；变压器Tr包含原边绕组和副边绕组，变压器Tr原边绕组的阳极与副边绕组的一端互为同名端，变压器Tr原边绕组的阳极与副边绕组的另一端互为异名端；副边电路包含整流开关S D和输出电容Co；输入电容Cin一端连接输入正、另一端连接输入负；主开关S M与辅开关S A串联后与输入电容Cin并联；谐振电容Cr与变压器Tr的原边绕组串联，串联后的一端连接主开关S M与辅开关S A的连接点，串联后的另一端连接输入正或输入负；变压器Tr的副边绕组与整流开关S D串联后与输出电容Co并联，输出电容Co一端连接输出正、另一端连接输出负；其特征在于：还包括单向钳位网络，单向钳位网络用于控制励磁电感电流负向峰值，其连接关系为如下情况之一：
    (1)单向钳位网络的阳极与变压器原边绕组的阳极电联接，单向钳位网络的阴极与变压器原边绕组阴极电联接；
    (2)单向钳位网络的阳极与变压器副边绕组的同名端电联接，单向钳位网络的阴极与变压器副边绕组的异名端电联接；
    (3)不对称半桥变换器还包括第三绕组，变压器原边绕组的阳极与第三绕组的一端互为同名端，变压器原边绕组的阳极与第三绕组的另一端互为异名端，单向钳位网络的阳极与第三绕组的同名端电联接，单向钳位网络的阴极与第三绕组的异名端电联接。
  2. 根据权利要求1所述的不对称半桥变换器，其特征在于：变压器副边绕组的异名端与整流开关S D的一端电联接，整流开关S D的另一端与输出电容Co的一端电联接，作为输出正，变压器副边绕组的同名端与输出电容Co的另一端电联接，作为输出负。
  3. 根据权利要求1所述的不对称半桥变换器，其特征在于：变压器副边绕组的同名端与整流开关S D的一端电联接，整流开关S D的另一端与输出电容Co的一端电联接，作为输出正，变压器副边绕组的异名端与输出电容Co的另一端电联接，作为输出负。
  4. 根据权利要求1所述的不对称半桥变换器，其特征在于：当不对称半桥变换器包括第三绕组时，第三绕组为一个独立绕组。
  5. 根据权利要求1所述的不对称半桥变换器，其特征在于：当不对称半桥变换器包括第三绕组时，第三绕组与辅助绕组为同一绕组。
  6. 根据权利要求1至3任一项所述的不对称半桥变换器，其特征在于：单向钳位网络包括一只二极管和一只开关管，连接关系为以下两种之一：
    (1)二极管的阳极为单向钳位网络的阳极，二极管的阴极连接开关管的漏极，开关管的源极为单向钳位网络的阴极；
    (2)开关管的漏极为单向钳位网络的阳极，开关管的源极连接二极管的阳极，二极管的阴极为单向钳位网络的阴极。
  7. 根据权利要求6所述的不对称半桥变换器，其特征在于：二极管阳极与阴极的寄生电容容值为C Dow、开关管漏极与源极的寄生电容容值为C Qow、输入电压为V in、输出电压为V o、原边绕组和副边绕组或者第三绕组的匝比为N，各参数满足如下关系式：
    Figure PCTCN2019113679-appb-100001
  8. 根据权利要求1至3任一项所述的不对称半桥变换器，其特征在于：单向钳位网络包括两只开关管，两只开关管的源极连接，其中一只开关管的漏极为单向钳位网络的阳极，另一只开关管的漏极为单向钳位网络的阴极。
  9. 根据权利要求1至3任一项所述的不对称半桥变换器，其特征在于：单向钳位网络包括一只二极管、一只开关管和一只电容，连接关系为以下两种之一：
    (1)二极管的阳极与电容一端相连，为单向钳位网络的阳极，二极管的阴极与电容另一端、开关管的漏极相连，开关管的源极为单向钳位网络的阴极；
    (2)二极管的阴极与电容一端相连，为单向钳位网络的阴极，二极管的阳极与电容另一端、开关管的源极相连，开关管的漏极为单向钳位网络的阳极。
  10. 根据权利要求9所述的不对称半桥变换器，其特征在于：二极管阳极与阴极的寄生电容容值为C Dow、开关管漏极与源极的寄生电容容值为C Qow、电容容值为C ow、输入电压为V in、输出电压为V o、原边绕组和副边绕组或者第三绕组的匝比为N，各参数满足如下关系式：
    Figure PCTCN2019113679-appb-100002
  11. 根据权利要求1至3任一项所述的不对称半桥变换器，其特征在于：单向钳位网络包括一只二极管、一只开关管、一只稳压管和一只电容，连接关系为以下两种之一：
    (1)二极管的阳极与稳压管的阳极相连，为单向钳位网络的阳极，二极管的阴极与电容一端、开关管的漏极相连，稳压管的阴极与电容的另一端相连，开关管的源极为单向钳位网络的阴极；
    (2)二极管的阴极与稳压管的阴极相连，为单向钳位网络的阴极，二极管的阳极与电容一端、开关管的源极相连，稳压管的阳极与电容的另一端相连，开关管的漏极为单向钳位网络的阳极。
  12. 一种不对称半桥变换器的控制方法，其特征在于：每个循环周期包含五个阶段：励磁阶段，辅开关零电压开通阶段，去磁阶段，电流钳位阶段，主开关零电压开通阶段；
    在励磁阶段、辅开关零电压开通阶段，单向钳位网络关断；
    在去磁阶段，辅开关导通，单向钳位网络导通或者关断均可，单向钳位网络没有电流流过；至此阶段结束时刻，励磁电感电流达到设定值，辅开关关断，单向钳位网络处于导通状态，钳位电流流过单向钳位网络；
    在电流钳位阶段，单向钳位网络导通，钳位电流流过单向钳位网络，单向钳位网络保持钳位电流基本不变，至此阶段结束时刻，单向钳位网络关断；
    在主开关零电压开通阶段，单向钳位网络已被关断，单向钳位网络中的钳位电流被释放，使主开关电压降低至零或接近零，此时控制主开关导通，实现主开关零电压开通。
  13. 根据权利要求12所述的一种不对称半桥变换器的控制方法，其特征在于：励磁电感电流设定值为负值或零、且与输入电压相关。
  14. 根据权利要求12所述的一种不对称半桥变换器的控制方法，其特征在于：每个开关循环周期的持续时间相同。
  15. 根据权利要求14所述一种不对称半桥变换器的控制方法，其特征在于：控制使用脉冲宽度调制控制方式。
  16. 根据权利要求14所述的一种不对称半桥变换器的控制方法，其特征在于：开关循环周期的中心开关频率固定，实际开关频率以中心开关频率为中心，在设定的下限频率、设定的上限频率之间周期性变化。
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