WO2012100740A1 - 准谐振推挽变换器及其控制方法 - Google Patents

准谐振推挽变换器及其控制方法 Download PDF

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Publication number
WO2012100740A1
WO2012100740A1 PCT/CN2012/070699 CN2012070699W WO2012100740A1 WO 2012100740 A1 WO2012100740 A1 WO 2012100740A1 CN 2012070699 W CN2012070699 W CN 2012070699W WO 2012100740 A1 WO2012100740 A1 WO 2012100740A1
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WIPO (PCT)
Prior art keywords
resonant
power
secondary winding
input
switch
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PCT/CN2012/070699
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English (en)
French (fr)
Inventor
何志峰
杨育程
Original Assignee
联正电子(深圳)有限公司
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Application filed by 联正电子(深圳)有限公司 filed Critical 联正电子(深圳)有限公司
Priority to US13/982,207 priority Critical patent/US9252677B2/en
Priority to EP12739642.2A priority patent/EP2670039B1/en
Publication of WO2012100740A1 publication Critical patent/WO2012100740A1/zh

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/337—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration
    • H02M3/3376—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/337—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration
    • H02M3/3372—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration of the parallel type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present invention relates to the field of power electronics, and more particularly to a quasi-resonant push-pull converter and a control method therefor.
  • filter inductors, capacitors, and transformers account for a large proportion of volume and weight. Taking effective measures to reduce the size and weight of these components is the main way to miniaturize and reduce weight. Increasing the switching frequency can increase the cutoff frequency of the filter accordingly, resulting in smaller inductors and capacitors, reducing the size and weight of the filter. Increasing the switching frequency also reduces the size and weight of the transformer.
  • Soft switching technology has emerged for these problems, which can solve the switching loss problem in the converter, and can also solve the electromagnetic interference EMI (Electromagnetic Interference) problem caused by the hard switch.
  • Soft switching technology usually refers to zero voltage switch ZVS (Zero Voltage Switch) and zero current switch ZCS (Zero Current Switch), or similar zero voltage switch and zero current switch.
  • Soft-switching converters for DC-DC converters include: Resonant Converters, Quasi Resonant Converters, Multiple Resonant Converters, Zero-Switching PWM Converters (Zero Switching PWM Converter) ) and a variety of soft switching technologies such as Zero Transition PWM Converter.
  • the resonant converter is actually a DC switching power supply load resonant converter, passed in the standard
  • the PWM converter is obtained by simply adding a resonant element to the structure.
  • the resonance mode of the resonant element it can be divided into two types: a series resonant converter and a parallel resonant converter; according to the connection relationship between the load and the resonant circuit, it can be divided into a series load resonant converter and a parallel load resonant converter.
  • the working principle is mainly to resonate the resonance element with the load, so that the current or voltage passing through the switching element is shaped into a sinusoidal waveform, and the switching element is turned on or off at the zero crossing of the current or voltage to realize the soft switching process.
  • the Quasi Resonant Converter is called quasi-resonant because its circuit operates in resonance for only a fraction of a switching cycle.
  • the quasi-resonant converter changes the current or voltage on the switching device by a sinusoidal law by resonance, thereby generating a zero current or zero voltage switching condition, which greatly reduces the switching loss and switching noise of the converter.
  • Multiple Resonant Converters are called multiple resonances because of the resonant topology and parameters in the circuit. It generally implements a zero voltage switch of the switch, but it can only use a frequency control method.
  • the zero-voltage multi-resonant converter is often used in practice, mainly because it absorbs the junction capacitance of the switching transistor and the rectifier diode, and at the same time realizes zero-voltage switching of the switching transistor and the rectifier diode.
  • the zero-switch PWM converter consists of a zero-voltage PWM converter and a zero-current PWM converter. Based on the quasi-resonant soft-switch, an auxiliary switching transistor is added to control the resonant process of the resonant component to achieve PWM control.
  • the commutation is realized only by the resonance, and the PWM operation mode is still adopted after the commutation, so that the defects of the hard switching PWM in the switching process can be overcome, and the low steady-state loss and the low steady-state stress of the hard-switching PWM converter can be retained. advantage.
  • the zero-conversion PWM converter includes a ZVS-PWM converter and a ZCS-PWM converter.
  • This type of converter combines a quasi-resonant converter with a conventional PWM converter to block the resonance process by an additional auxiliary active switch, allowing the circuit to operate in a ZCS or ZVS quasi-resonant mode for a period of time.
  • the other part of the time runs in PWM mode, which has the characteristics of soft switch and PWM constant frequency duty cycle adjustment.
  • the resonant inductor is connected in series in the main power loop, so there is always a large circulating current in the circuit, which inevitably increases the conduction loss of the circuit.
  • the energy storage of the inductor has a lot to do with the input voltage and output load.
  • the widespread use of soft switching technology has revolutionized the design of power electronic converters.
  • the application of soft switching technology enables the power electronic converter to have higher efficiency, greatly reducing its own loss, higher power density, its own volume, weight reduction, and higher reliability; and can effectively reduce power
  • the electromagnetic pollution and environmental pollution caused by the conversion device provide an effective way and method for vigorously developing green power electronic products. Summary of the invention
  • the present invention provides a control method of a quasi-resonant push-pull converter for use in a push-pull converter having a resonant capacitor, the push-pull converter comprising: a DC input power supply for providing a converter a DC input; a first and a second power input unit respectively connected to the DC input power source for respectively providing inputs to the converter in different periods, comprising first and second power switching tubes, a first primary winding And a second primary winding; a power output circuit for providing an output of the converter, comprising a secondary winding and a full bridge rectifier circuit; first and second output capacitors coupled to the power output circuit for storing power
  • the DC power output by the output circuit is characterized in that
  • a resonant element is disposed in the power output circuit, and a quasi-resonant switching circuit is implemented by voltage feedback;
  • a switching circuit is controlled by voltage feedback to control the turns ratio of the primary winding and the secondary winding of the push-pull converter.
  • the disposing the resonant element in the power output circuit is specifically: setting a transformer leakage inductance, a resonant capacitor, and first and second inductances in the power output circuit.
  • the method further comprises: setting a first full bridge rectifier circuit, a second full bridge rectifier circuit and a switching element in the power output circuit, controlling the working turns of the secondary winding by controlling the on or off of the switching element, and changing the primary level
  • the turns ratio of the winding and the secondary winding is such that the operation of the resonant circuit does not differ depending on the input voltage, so that the primary and second power switches of the primary side operate in the switching mode in the resonant mode.
  • the controlling a switching circuit by voltage feedback is specifically: setting a voltage value, When the DC input voltage is detected to be higher than the set value, the switch circuit is set to be turned off; when the DC input voltage is detected to be lower than the set value, the switch circuit is set to be turned on, thereby controlling the secondary The number of working turns of the winding.
  • the invention also provides a quasi-resonant push-pull converter comprising a DC input power source for providing a DC input to the converter; and first and second power input units respectively connected to the DC input power source for different periods Providing inputs to the converter, respectively, comprising first and second power switching tubes, a first primary winding and a second primary winding; a power output circuit for providing an output of the converter, including a secondary winding and a full a bridge rectifier circuit; the first and second output capacitors are coupled to the power output circuit for storing DC power output by the power output circuit, wherein
  • the power output circuit further includes a transformer leakage inductance, a resonance capacitance and an inductance, and a quasi-resonant switching circuit is realized by voltage feedback;
  • the power output circuit also controls a switching circuit by voltage feedback to control the turns ratio of the primary winding and the secondary winding of the push-pull converter.
  • the method further comprises: setting a first full bridge rectifier circuit, a second full bridge rectifier circuit and a switching element in the power output circuit, controlling the working turns of the secondary winding by controlling the on or off of the switching element, and changing the primary level
  • the turns ratio of the winding and the secondary winding is such that the operation of the resonant circuit does not differ depending on the input voltage, so that the primary and second power switches of the primary side operate in the switching mode in the resonant mode.
  • the switching element includes a first switch and a second switch, and the first switch and the second switch are respectively connected between the first full bridge rectifier circuit and the second full bridge rectifier circuit.
  • the resonant frequency is set to be twice the switching frequency.
  • the secondary winding includes a first secondary winding, a second secondary winding, a third secondary winding, and a fourth secondary winding, when the first switch and the second switch are both turned on, by the first time
  • the stage winding, the second secondary winding, the third secondary winding, and the fourth secondary winding provide an output; when both the first switch and the second switch are off, the output is provided by the second secondary winding and the third secondary winding.
  • the method further includes setting a voltage value, and when the DC input voltage is higher than a set value, setting the first switch and the second switch to be off; when the DC input voltage is lower than the set value, Then, both the first switch and the second switch are set to be turned on.
  • FIG. 1 is a structural view of a push-pull converter according to the prior art
  • FIG. 2 is a structural diagram of a quasi-resonant push-pull converter according to the prior art
  • Figure 3 illustrates the loss of Q1 and Q2 in hard-switch mode and resonant mode
  • Figure 4 illustrates a comparison of the power transient efficiencies of Figures 1 and 2 at different input voltages
  • Figure 5 is a block diagram of a quasi-resonant push-pull converter in accordance with an embodiment of the present invention
  • Figure 6 illustrates a comparison of the power transient efficiencies of Figures 1, 2, and 5 at different input voltages.
  • the terms “a”, “an”, “the” it should be understood that the terms “comprising”, “comprising”, “”,”” Or a plurality of features, entities, steps, operations, units, components, and/or groups having them. It will be understood that when a unit is referred to as “connected,” or “coupled” to another unit, it can be directly connected or coupled to another unit, and the intermediate unit can be present. Further, “connected” or “coupled” as used herein includes a wireless connection or coupling. The term “and / or” as used herein includes any and all combinations of one or more of the associated items listed.
  • the push-pull converter has a simple structure, and energy is transmitted through the alternate conduction of the two switching tubes.
  • the transformer in the push-pull converter is bidirectionally excited, and the push-pull converter can transmit a large amount of power with a high utilization rate.
  • only one of the input loops has a conduction voltage drop, which results in a relatively small conduction loss, making it particularly suitable for power supply systems with low input voltages.
  • the push-pull converter has a magnetic bias problem, and the circuit must have good symmetry, otherwise it is easy to cause DC bias to cause the core to saturate.
  • Switch management is subject to twice the input voltage, but due to the presence of leakage inductance, the voltage spike when the switch is turned off is greater than this value. Therefore, the transformer windings must be tightly coupled to reduce the leakage inductance. In addition, new requirements are imposed on the withstand voltage of the switching tubes.
  • the primary side of the push-pull converter includes: DC input supply, MOSFET switches Q1 and Q2, input capacitor Cin, and primary windings with turns N1 and N2.
  • the connection relationship is as follows: The positive pole of the DC input power is connected to the source of the MOSFET switch Q1 through the primary winding of the number N1, the drain of the MOSFET switch Q1 is grounded; the positive pole of the DC input power is connected through the primary winding of the number N2 The source of the MOSFET switch Q2, the drain of the MOSFET switch Q2 is grounded; and the input capacitor Cin is connected in parallel with the DC input power supply, one end is connected to the same name end of the primary winding of the number N1, and the other end is connected to the ground.
  • the secondary side of the push-pull converter includes: a secondary winding with a number of turns N4, a secondary winding with a number of turns N5, a full-bridge rectifier, inductors L2 and L3, and output capacitors Col and Co2.
  • the full bridge rectifier includes: diodes Dl, D2, D5 and D6.
  • connection relationship is: the same name of the secondary winding with the number of turns N4 and the other end of the secondary winding with the number of turns N5 are grounded, and the other end of the secondary winding with the number of turns N4 is connected to the anode of the diode D1 and the diode D5
  • the cathode of the secondary winding having the number of turns N5 is connected to the anode of the diode D2 and the cathode of the diode D6.
  • the cathode of diode D1 is connected to the cathode of diode D2 and one end of inductor L2, the other end of inductor L2 is connected to the anode of output capacitor Col, the cathode of output capacitor Col is grounded; the anode of diode D5 is connected to the anode of diode D6 and one end of inductor L3 The other end of the inductor L3 is connected to the cathode of the output capacitor Co2, and the anode of the output capacitor Co2 is grounded.
  • FIG. 1 The circuit action of Figure 1 is specifically as follows: During the first cycle, the MOSFET switch Q1 is turned on, The current flows along the primary winding with the number of turns N1 to the MOSFET switch Q1. Correspondingly, on the secondary side, the current flows from the other end of the secondary winding having the number of turns N4 to the same end through the output capacitor Co2, the inductor L3 and the diode D5; meanwhile, the current is from the secondary winding of the number N5 The other end flows to its end of the same name and flows to the output capacitor Col through diode D2 and inductor L2.
  • the MOSFET switch Q2 is turned on, and the current flows along the primary winding with the number of turns N2 to the MOSFET switch Q2. Accordingly, on the secondary side, the current flows from the same-name end of the secondary winding having the number of turns N5 to the other end through the output capacitor Co2, the inductor L3, and the diode D6; meanwhile, the current is from the secondary winding having the number of turns N4 The same name flows to the other end, and flows to the output capacitor Col through the diode D1 and the inductor L2.
  • Push-pull converters are easy to control and low cost. However, when the input power is high, the push-pull converter cannot provide high transmission efficiency. Especially at higher frequencies or high power densities, the switching losses of the power switches Q1 and Q2 become more severe.
  • the present invention proposes a quasi-resonant push-pull converter.
  • the primary side of the push-pull converter consists of: a DC input supply, MOSFET switches Q1 and Q2, an input capacitor Cin, and a primary winding of N1 and N2.
  • connection relationship is as follows: The positive pole of the DC input power is connected to the source of the MOSFET switch Q1 through the primary winding of the number N1, the drain of the MOSFET switch Q1 is grounded; the positive pole of the DC input power is connected through the primary winding of the number N2 The source of the MOSFET switch Q2, the drain of the MOSFET switch Q2 is grounded; and the input capacitor Cin is connected in parallel with the DC input power supply, one end is connected to the same name end of the primary winding of the number N1, and the other end is connected to the ground.
  • the secondary side of the push-pull converter includes: a secondary winding with a number of turns N4, a secondary winding with a number of turns N5, an equivalent leakage inductance of the transformer L1, a full-bridge rectifier, a resonant capacitor C3, inductors L2 and L3, and an output capacitor Col and Co2.
  • the full bridge rectifier includes: diodes D1, D2, D5 and D6.
  • the connection relationship is as follows: the same name of the secondary winding with the number of turns N4 and the other end of the secondary winding with the number of turns N5 are grounded, and the other end of the secondary winding with the number of turns N4 is connected to the equivalent leakage inductance L1 of the transformer.
  • the other end of the transformer leakage inductance L1 is connected to the anode of diode D1 and diode D5.
  • the cathode of the secondary winding having the number of turns N5 is connected to the anode of the diode D2 and the cathode of the diode D6.
  • the cathode of the diode D1 is connected to the cathode of the diode D2, one end of the resonant capacitor C3 and one end of the inductor L2, the other end of the inductor L2 is connected to the anode of the output capacitor Col, the cathode of the output capacitor Col is grounded, and the anode of the diode D5 is connected to the diode D6.
  • the anode, the other end of the resonant capacitor C3, and one end of the inductor L3, the other end of the inductor L3 is connected to the cathode of the output capacitor Co2, and the anode of the output capacitor Co2 is grounded.
  • Figure 3 illustrates the losses of Q1 and Q2 in hard-switch mode and resonant mode. It can be clearly seen that in the hard switching mode, the losses of the MOSFET switches Q1 and Q2 are significantly greater than the losses of Q1 and Q2 in the resonant mode.
  • the design of the resonant circuit is usually designed in accordance with the requirements of the low voltage input, so the resonant circuit cannot At the same time, it meets the requirements of high voltage input and discharge of low voltage after a period of time, and the optimal resonance effect cannot be maintained. Therefore, when the input voltage of the battery is high, the duty becomes small and the resonance effect is not good. When the Q1 and Q2 switches enter the hard switching mode, the switching loss is large and the conduction loss is also very large.
  • Figure 4 illustrates a comparison of the power transient efficiencies of Figures 1 and 2 at different input voltages.
  • the abscissa in Figure 4 represents the input battery voltage, increasing from left to right, with a minimum voltage of 60 volts and a maximum voltage of 84 volts, where each coordinate is incremented by 2 volts.
  • the ordinate represents the ratio of output power to input power, expressed as a percentage, with a minimum ratio of 79% and a highest ratio of 88%, with each coordinate increasing by 1%.
  • the rectangular points represent the power transient efficiency values of the circuit shown in Figure 2 for different input voltages.
  • the diamond points represent the power transient efficiency values of the circuit shown in Figure 1 for different input voltages.
  • the ratio of the output power to the input power of the circuit shown in Figure 1 and the circuit shown in Figure 2 decreases with the input battery voltage. And rise.
  • the ratio of the output power to the input power of the circuit shown in Fig. 1 decreases as the input battery voltage decreases.
  • the output power of the circuit shown in Figure 2 is the same as the input power, which is less than the output power and input of the input battery voltage of 70 volts.
  • the ratio of power is greater than the ratio of the output power to the input power of the input battery voltage of 60 volts.
  • the ratio of the output power of the two to the input power reaches their respective maximum values, and when the input battery voltage is 84 volts, the ratio of the output power of the two to the input power reaches their respective minimum values. .
  • the ratio of the output power to the input power of the circuit shown in Figure 1 is greater than the ratio of the output power to the input power of the circuit shown in Figure 2.
  • the ratio of the output power to the input power of the circuit shown in Figure 1 is less than the ratio of the output power to the input power of the circuit shown in Figure 2.
  • the efficiency of the quasi-resonant push-pull converter shown in FIG. 2 is higher than that of the push-pull converter shown in FIG. 1, but when the battery is input with high voltage, for example, 70 volts to 84 volts. In the volt interval, the efficiency of the quasi-resonant push-pull converter is still not high. Therefore, it is desirable to provide a converter that provides higher efficiency even when the input DC power supply has a wide voltage range.
  • FIG. 5 is a block diagram of a quasi-resonant push-pull converter in accordance with an embodiment of the present invention.
  • the primary side of the push-pull converter consists of: DC input supply, MOSFET switches Q1 and Q2, input capacitor Cin, and primary windings for turns N1 and N2.
  • connection relationship is as follows: The positive pole of the DC input power is connected to the source of the MOSFET switch Q1 through the primary winding of the number N1, the drain of the MOSFET switch Q1 is grounded; the anode of the DC input power is connected through the primary winding of the number N2 The source of the MOSFET switch Q2, the drain of the MOSFET switch Q2 is grounded; and the input capacitor Cin is connected in parallel with the DC input power supply, one end is connected to the same name end of the primary winding of the number N1, and the other end is connected to the ground.
  • the secondary side of the push-pull converter includes: a secondary winding having a number of turns N3, N4, N5 and N6, a first full bridge rectifier, a second full bridge rectifier, first and second switches, a resonant capacitor C3, an inductor L2 and L3, output capacitors Col and Co2.
  • the first full bridge rectifier comprises: diodes D1, D2, D5 and D6;
  • the second full bridge rectifier comprises: D3, D4, D7 and D8.
  • the first full-bridge rectifier connects the other end of the secondary winding with the number of turns and the same name of the secondary winding with the number of turns N6; the second full-bridge rectifier connects the same-numbered end and the number of turns of the secondary winding with the number N3 The other end of the secondary winding of N6.
  • the specific connection relationship is as follows: The other end of the secondary winding with the number of turns N3 is connected to the equivalent leakage inductance L1 of the transformer, one end of the transformer, the leakage inductance L1 of the transformer, and the other end of the diode is connected to the anode of the diode D1 and the anode of the D5.
  • the same end of the secondary winding with the number of turns N3 is connected to the end of the transformer equivalent leakage inductance L1
  • the other end of the leakage inductance L1 of the transformer is connected to the anode of the diode D3, the cathode of the diode D7 and the secondary winding of the number N4 another side.
  • the same name of the secondary winding with the number of turns N4 and the other end of the secondary winding with the number of turns N5 are grounded, the same name of the secondary winding of the number N5 is connected to the anode of the diode D4, and the cathode and the number of turns of the diode D8
  • the other end of the secondary winding of N6; the same name of the secondary winding of the number N6 is connected to the anode of the diode D2 and the cathode of the diode D6.
  • the cathode of diode D1 is connected to the cathode of diode D2, one end of switch K1.
  • the switch K1 can be replaced with another circuit capable of implementing the switching function.
  • the other end of the switch K1 is connected to the cathodes of the diodes D3 and D4, one end of the resonant capacitor C3, and one end of the inductor L2; the other end of the inductor L2 is connected to the anode of the output capacitor Col, and the negative pole of the output capacitor Col is grounded.
  • the anode of diode D5 is connected to the anode of diode D6, one end of switch K2.
  • the switch K2 can be replaced with another circuit capable of implementing the switching function.
  • the other end of the switch ⁇ 2 is connected to the anodes of the diodes D7 and D8, the other end of the resonant capacitor C3, and one end of the inductor L3; the other end of the inductor L3 is connected to the cathode of the output capacitor Co2, and the anode of the output capacitor Co2 is grounded.
  • the integrated resonant push-pull converter shown in Fig. 6 can maintain the resonance effect in a good state.
  • the secondary windings operating on the secondary side are N4 and N5.
  • a preferred embodiment of the present invention increases the number of turns N3 and N6 of the secondary side of a group of transformers, and designs the resonant circuit in a state of outputting a low voltage; when both switches K1 and K2 are turned on, the number of turns of the secondary side is increased to (N3+N4) and (N5+N6).
  • the first switch K1 and the second switch K2 are both turned off, and the secondary windings of the secondary side are N4 and N5, so that The duty of the Q1 and Q2 switches becomes larger, the resonant capacitor C3 and the leakage inductance L1 resonate, and the Q1 and Q2 switches enter the resonant mode; when the battery voltage drops to a set value due to discharge, for example, when the DC input voltage is lower than the setting
  • the first switch K1 and the second K2 are both turned on, and the number of turns of the secondary side is increased to (N3+N4) and (N5+N6), and the number of turns of the primary winding and the secondary winding are changed.
  • the resonant capacitor C3 and the leakage inductance L1 In order to maintain the value of the output voltage, the resonant capacitor C3 and the leakage inductance L1, the resonant, Q1 and Q2 switches operate in the resonant mode as well. Thus, in the process of discharging the entire battery, the effects of the Q1 and Q2 switching resonances are maintained in a good state, thereby improving the overall discharge efficiency.
  • Figure 6 illustrates a comparison of the power transient efficiencies of Figures 1, 2, and 5 at different input voltages.
  • the abscissa of Figure 6 represents the input battery voltage, increasing from left to right, with a minimum voltage of 60 volts and a maximum voltage of 84 volts, where each coordinate is incremented by 2 volts.
  • the ordinate represents the ratio of output power to input power, expressed as a percentage, with a minimum ratio of 79% and a highest ratio of 88%, with each coordinate increasing by 1%.
  • the triangle points represent the power transient efficiency values of the circuit shown in Figure 5 under different input voltages.
  • the rectangular points represent the power transient efficiency values of the circuit shown in Figure 2 under different input voltages.
  • the diamond points represent different input voltages.
  • Figure 1 The power transient efficiency value of the circuit shown.
  • the ratio of the output power to the input power of the circuit shown in Figure 1 and the circuit shown in Figure 2 decreases with the input battery voltage. And rise.
  • the ratio of the output power to the input power of the circuit shown in Fig. 1 decreases as the input battery voltage decreases.
  • the output power of the circuit shown in Figure 2 is the same as the input power, which is less than the ratio of the output power to the input power of the input battery voltage of 70 volts.
  • the ratio of the output power to the input power of the circuit shown in Figure 5 increases as the input battery voltage decreases, at the input battery voltage from 76 volts to 70 volts.
  • the ratio of the output power to the input power of the circuit shown in Figure 5 decreases as the input battery voltage decreases.
  • the output power of the circuit shown in Figure 5 is in the range of the input battery voltage from 70 volts to 60 volts.
  • the ratio of the input power to the input power is the same as the ratio of the output power to the input power of the circuit shown in Figure 2.
  • the ratio of the output power to the input power of the circuit shown in Figure 1 and the circuit shown in Figure 2 reaches their respective maximum values, and the input battery voltage is 76 volts, the circuit shown in Figure 5.
  • the ratio of the output power to the input power reaches a maximum.
  • the ratio of the output power of the three to the input power reaches their respective minimum values.
  • the ratio of the output power to the input power of the circuit shown in Figure 1 is greater than the ratio of the output power to the input power of the circuit shown in Figure 2.
  • the ratio of the output power to the input power of the circuit shown in Figure 1 is less than the ratio of the output power to the input power of the circuit shown in Figure 2.
  • the ratio of the output power to the input power of the circuit shown in Fig. 5 is the same as the ratio of the output power to the input power of the circuit shown in Fig. 2.
  • the ratio of the output power to the input power of the circuit shown in Figure 5 is greater than the ratio of the output power to the input power of the circuit shown in Figure 1 and the circuit shown in Figure 2. .
  • the efficiency of the quasi-resonant push-pull converter shown in FIG. 5 is higher than that of the push-pull converter shown in FIG. 1 and FIG. 2, and the integrated quasi-resonant push-pull converter Achieve higher efficiency.

Description

准谐振推挽变换器及其控制方法
技术领域
本发明涉及电力电子技术领域， 更具体地， 涉及准谐振推挽变换器及 其控制方法。
背景技术
目前， 电力电子技术的发展对电子产品提出了小型化、轻量化的要求， 同时也对效率和电磁兼容性提出了更高的要求。 在电力电子装置中， 滤波 电感、 电容和变压器在体积和重量上占很大比例， 采取有效措施减小这些 元器件的体积和重量是小型化、 轻量化的主要途径。 提高开关频率可以相 应的提高滤波器的截止频率， 从而选用较小的电感和电容， 降低滤波器的 体积和重量。 提高开关频率同样可以降低变压器的体积和重量。
但是， 提高开关频率的同时， 开关损耗增大， 感性关断、 容性开通、 二极管反向恢复等问题将加剧， 导致电路效率下降， 电磁干扰增大。 针对 这些问题出现了软开关技术， 它可以解决变换器中的开关损耗问题， 同时 也能解决由硬开关引起的电磁干扰 EMI ( Electro Magnetic Interference ) 问题。 软开关技术通常是指零电压开关 ZVS ( Zero Voltage Switch )和零 电流开关 ZCS ( Zero Current Switch ) ， 或者近似的零电压开关与零电流 开关。
DC-DC 变换器的软开关变换器包括： 谐振变换器（Resonant Converter) , 准谐振变换器 (Quasi Resonant Converter) , 多谐振变换器 (Multiple Resonant Converter) , 零开关 PWM 变换器 (Zero Switching PWM Converter)以及零转换 PWM 变换器（Zero Transition PWM Converter)等多种软开关技术。
谐振变换器实际上是直流开关电源负载谐振变换器， 通过在标准 PWM 变换器结构上简单地附加谐振元件的方法而得到。 按照谐振元件的 谐振方式， 可分为串联谐振变换器和并联谐振变换器两类； 按负载与谐振 电路的连接关系， 又可分为串联负载谐振变换器和并联负载谐振变换器。 其工作原理主要是通过谐振元件与负载的谐振， 使经过开关元件的电流或 电压被整形为正弦波形， 开关元件在电流或电压的过零处开通或关断， 实 现软开关过程。
准谐振变换器 (Quasi Resonant Converter)， 因其电路工作在谐振的时 间只占一个开关周期中的一部分， 故称为准谐振。 准谐振变换器通过谐振 使开关器件上的电流或电压按准正弦规律变化， 从而产生零电流或零电压 开关条件， 极大地减小了变换器的开关损耗和开关噪声。
多谐振变换器 (Multiple Resonant Converter), 由于电路中谐振拓朴和 参数不止一个， 故称为多谐振。 其一般能实现开关管的零电压开关， 但还 是只能采用频率控制方法。 实际常常用零电压多谐振变换器， 主要是因为 它吸收了开关管和整流二极管的结电容， 同时实现了开关管和整流二极管 零电压开关。
零开关 PWM变换器包括零电压 PWM变换器和零电流 PWM变换器， 它们是在准谐振软开关的基础上， 加入一个辅助开关管， 来控制谐振元件 的谐振过程， 实现 PWM控制。 只利用谐振实现换相， 换相完毕后仍采用 PWM工作方式， 从而既能克服硬开关 PWM在开关过程中的缺陷， 又能 保留硬开关 PWM变换器的低稳态损耗和低稳态应力的优点。
零转换 PWM变换器包括 ZVS-PWM变换器和 ZCS-PWM变换器。 这种类型的变换器， 将准谐振变换器和常规的 PWM变换器相结合， 通过 附加的辅助有源开关阻断谐振过程,使电路在一周期内，一部分时间按 ZCS 或 ZVS准谐振方式运行， 另一部分时间按 PWM方式运行， 既具有软开关 的特点， 又具有 PWM恒频占空比调节的特点。
在 ZVS-PWM变换器和 ZCS-PWM变换器中， 谐振电感串联在主功 率回路中， 因此电路中总是存在着很大的环流能量， 这不可避免地增加了 电路的导通损耗。 另外， 电感储能与输入电压和输出负载有很大关系， 这 对软开关技术的广泛应用， 使电力电子变换器的设计出现了革命性的 变化。 软开关技术的应用使电力电子变换器可以具有更高的效率一自身损 耗大大降低， 更高的功率密度一自身体积、 重量大大减小， 以及更高的可 靠性； 并可有效地减小电能变换装置引起的电磁污染和环境污染， 为大力 绿色电力电子产品提供了有效的方式和方法。 发明内容
为了解决上述问题， 本发明提供一种准谐振推挽变换器的控制方法， 应用于具有谐振电容的推挽变换器中， 所述推挽变换器包括： 直流输入电 源， 用于为变换器提供直流输入； 第一和第二功率输入单元， 分别连接所 述直流输入电源， 用于在不同的周期内分别为变换器提供输入， 其包括第 一和第二功率开关管， 第一原级绕组和第二原级绕组； 功率输出电路， 用 于提供变换器的输出， 其包括次级绕组和全桥整流电路； 第一和第二输出 电容， 连接于所述功率输出电路， 用于储存功率输出电路所输出的直流电 能， 其特征在于，
在功率输出电路中设置谐振元件， 通过电压反馈来实现准谐振开关电 路； 以及
通过电压反馈来控制一开关电路， 从而控制推挽变换器的原级绕组和 次级绕组的匝数比。
优选地， 所述在功率输出电路中设置谐振元件具体为： 在功率输出电 路中设置变压器漏感、 谐振电容以及第一和第二电感。
优选地， 还包括在功率输出电路中设置第一全桥整流电路、 第二全桥 整流电路和开关元件， 通过控制开关元件的导通或截止来控制次级绕组的 工作匝数， 改变原级绕组和次级绕组的匝数比， 以保持谐振电路的工作不 因输入电压的高低而不同， 使原边的第一和第二功率开关管均工作在谐振 模式下的开关模式。
优选地，所述通过电压反馈来控制一开关电路具体为：设定一电压值， 当侦测到直流输入电压高于该设定值时， 将开关电路设置为截止； 当侦测 到直流输入电压低于该设定值时， 将该开关电路设置为导通， 从而控制次 级绕组的工作匝数。
本发明还提供一种准谐振推挽变换器， 包括直流输入电源， 用于为变 换器提供直流输入； 第一和第二功率输入单元， 分别连接所述直流输入电 源， 用于在不同的周期内分别为变换器提供输入， 其包括第一和第二功率 开关管， 第一原级绕组和第二原级绕组； 功率输出电路， 用于提供变换器 的输出， 其包括次级绕组和全桥整流电路； 第一和第二输出电容， 连接于 所述功率输出电路， 用于储存功率输出电路所输出的直流电能， 其特征在 于，
功率输出电路还包括变压器漏感、 谐振电容和电感， 通过电压反馈来 实现准谐振开关电路； 以及
功率输出电路还通过电压反馈来控制一开关电路， 从而控制推挽变换 器的原级绕组和次级绕组的匝数比。
优选地， 还包括在功率输出电路中设置第一全桥整流电路、 第二全桥 整流电路和开关元件， 通过控制开关元件的导通或截止来控制次级绕组的 工作匝数， 改变原级绕组和次级绕组的匝数比， 以保持谐振电路的工作不 因输入电压的高低而不同， 使原边的第一和第二功率开关管均工作在谐振 模式下的开关模式。
优选地， 所述开关元件包括第一开关和第二开关， 所述第一开关和第 二开关分别连接于所述第一全桥整流电路和第二全桥整流电路之间。
优选地， 其中谐振频率设置为开关频率的 2倍。
优选地， 所述次级绕组包括第一次级绕组、 第二次级绕组、 第三次级 绕组和第四次级绕组， 当第一开关和第二开关均导通时， 由第一次级绕组、 第二次级绕组、 第三次级绕组和第四次级绕组提供输出； 当第一开关和第 二开关均截止时， 由第二次级绕组和第三次级绕组提供输出。
优选地， 还包括设定一电压值， 当直流输入电压高于一该设定值时， 则将第一开关和第二开关均设置为截止；当直流输入电压低于该设定值时， 则将第一开关和第二开关均设置为导通。 附图说明
图 1为根据现有技术的推挽式变换器的结构图；
图 2为根据现有技术的准谐振推挽式变换器的结构图；
图 3说明了在硬开关模式和谐振模式下 Q1和 Q2的损失；
图 4说明了在不同的输入电压下图 1和图 2的功率暂态效率的比较图； 图 5根据本发明实施方式的准谐振推挽式变换器的结构图；
图 6说明了在不同的输入电压下图 1、 图 2和图 5的功率暂态效率的 比较图。 具体实施方式
现在参考附图介绍本发明的示例性实施方式， 然而， 本发明可以用许 多不同的形式来实施， 并且不局限于此处描述的实施例， 提供这些实施例 是为了详尽地且完全地公开本发明， 并且向所属技术领域的技术人员充分 传达本发明的范围。 对于表示在附图中的示例性实施方式中的术语并不是 对本发明的限定。 在附图中， 相同的单元 /元件使用相同的附图标记。
除非另有说明， 此处使用的"一"、 "一个"、 "所述 "和"该"也包括复数 形式。 此外， 应当理解的是， 本说明书中使用的术语"包括"、 "包含 "和 / 或"含有"， 指定了一些特征、 实体、 步骤、 操作、 单元、 和 /或元件， 但并 不排除一个或多个特征、 实体、 步骤、 操作、 单元、 元件和 /或有它们组成 的组。 应当理解的是， 当单元被称为"连接，，或"耦合，，到另一个单元时， 它 可以是直接和另一单元连接或耦合， 也可以存在中间单元。 此外， 此处所 指的"连接"或"耦合"包括无线连接或耦合。此处使用的术语 "和 /或"包括一 个或以上所列相关项目的任意组合和全部组合。
除非另有说明， 此处使用的术语 (包括科技术语）对所属技术领域的 技术人员具有通常的理解含义。 另外， 可以理解的是， 以通常使用的词典 限定的术语， 应当被理解为与其相关领域的语境具有一致的含义， 而不应 该被理解为理想化的或过于正式的意义。
推挽变换器结构简单， 通过两个开关管的交替导通实现能量的传递。 推挽变换器中的变压器是双向励磁， 推挽变换器可以传输较大的功率， 利 用率高。 在工作过程中， 输入回路中只有一个开关管的导通压降， 产生的 导通损耗相对较小， 因此特别适用于输入电压较低的电源系统。 但是推挽 变换器存在磁偏的问题， 电路必须具有良好的对称性， 否则容易引起直流 偏磁导致磁芯饱和。 开关管理论上承受两倍的输入电压， 但由于漏感的存 在， 开关管关断时的电压尖峰大于该值。 所以要求变压器绕组必须紧密藕 合， 以减小漏感， 另外对开关管的耐压也提出了新的要求。
图 1为根据现有技术的推挽式变换器的结构图。 推挽变换器的原边侧 包括： 直流输入电源， MOSFET开关管 Q1和 Q2, 输入电容 Cin以及匝 数 N1和 N2的一次绕组。 其连接关系为： 直流输入电源的正极通过匝数 N1的一次绕组的连接 MOSFET开关管 Q1的源极， MOSFET开关管 Q1 的漏极接地； 直流输入电源的正极通过匝数 N2 的一次绕组的连接 MOSFET开关管 Q2的源极， MOSFET开关管 Q2的漏极接地； 以及输入 电容 Cin与直流输入电源并联，一端连接到匝数 N1的一次绕组的同名端， 另一端与地线连接。
推挽变换器的副边侧包括： 由匝数为 N4的二次绕组， 匝数为 N5的二 次绕组， 全桥整流器， 电感 L2和 L3， 输出电容 Col和 Co2。 其中， 全桥 整流器包括： 二极管 Dl， D2, D5和 D6。 其连接关系为： 匝数为 N4的二 次绕组的同名端与匝数为 N5的二次绕组的另一端均接地，匝数为 N4的二 次绕组的另一端连接二极管 D1的阳极和二极管 D5的阴极， 匝数为 N5的 二次绕组的同名端连接二极管 D2的阳极和二极管 D6的阴极。 二极管 D1 的阴极连接到二极管 D2的阴极和电感 L2的一端， 电感 L2的另一端连接 输出电容 Col的正极， 输出电容 Col的负极接地； 二极管 D5的阳极连接 到二极管 D6的阳极和电感 L3的一端，电感 L3的另一端连接输出电容 Co2 的负极， 输出电容 Co2的正极接地。
图 1的电路动作具体为： 在第一周期内， MOSFET开关管 Q1导通， 电流沿匝数为 Nl的一次绕组， 流向 MOSFET开关管 Ql。 相应地， 在副 边侧， 电流经输出电容 Co2、 电感 L3和二极管 D5， 从匝数为 N4的二次 绕组的另一端流向其同名端； 同时， 电流从匝数为 N5的二次绕组的另一 端流向其同名端， 并经过二极管 D2和电感 L2流向输出电容 Col。
在第二周期内， 在第一周期内， MOSFET开关管 Q2导通， 电流沿匝 数为 N2的一次绕组， 流向 MOSFET开关管 Q2。 相应地， 在副边侧， 电 流经输出电容 Co2、 电感 L3和二极管 D6, 从匝数为 N5的二次绕组的同 名端流向其另一端； 同时， 电流从匝数为 N4的二次绕组的同名端流向其 另一端， 并经过二极管 D1和电感 L2流向输出电容 Col。
推挽式变换器容易控制并且成本低。 但是， 当输入功率较高时， 推挽 式变换器无法提供较高的传输效率。尤其是在更高频率或者大功率密度时， 功率开关管 Q1和 Q2的开关损耗变得更为严重。为了解决上述问题， 降低 功率开关管的开关损耗， 提供大功率密度， 高效率以及高功率节省的推挽 式变换器， 本发明提出一种准谐振推挽式变换器。
图 2为根据现有技术的的准谐振推挽式变换器的结构图。 推挽变换器 的原边侧包括： 直流输入电源， MOSFET开关管 Q1和 Q2,输入电容 Cin 以及匝数 Nl和 N2的一次绕组。其连接关系为： 直流输入电源的正极通过 匝数 N1的一次绕组的连接 MOSFET开关管 Q1的源极， MOSFET开关 管 Q1的漏极接地； 直流输入电源的正极通过匝数 N2的一次绕组的连接 MOSFET开关管 Q2的源极， MOSFET开关管 Q2的漏极接地； 以及输入 电容 Cin与直流输入电源并联，一端连接到匝数 N1的一次绕组的同名端， 另一端与地线连接。
推挽变换器的副边侧包括： 匝数为 N4的二次绕组， 匝数为 N5的二次 绕组， 变压器等效漏感 Ll， 全桥整流器， 谐振电容 C3， 电感 L2和 L3， 输出电容 Col和 Co2。 其中， 全桥整流器包括： 二极管 Dl, D2, D5和 D6。 其连接关系为： 匝数为 N4的二次绕组的同名端与匝数为 N5的二次 绕组的另一端均接地， 匝数为 N4的二次绕组的另一端连接变压器等效漏 感 L1的一端，变压器漏感 L1的另一端连接二极管 D1的阳极和二极管 D5 的阴极， 匝数为 N5的二次绕组的同名端连接二极管 D2的阳极和二极管 D6的阴极。 二极管 D1的阴极连接到二极管 D2的阴极、 谐振电容 C3的 一端以及电感 L2的一端， 电感 L2的另一端连接输出电容 Col的正极，输 出电容 Col的负极接地； 二极管 D5的阳极连接到二极管 D6的阳极、 谐 振电容 C3的另一端以及电感 L3的一端， 电感 L3的另一端连接输出电容 Co2的负极， 输出电容 Co2的正极接地。
图 3说明了在硬开关模式和谐振模式下 Q1和 Q2的损耗。可以明显的 看出， 在硬开关的模式下， MOSFET开关管 Q1和 Q2的损耗明显要大于 谐振模式下 Q1和 Q2的损耗。
但是, 当输入直流电源为一电池时， 因为电池放电的特性会维持在低 压的时间较久， 为维持整体放电的效率， 谐振电路的设计通常配合低电压 输入的要求而设计， 因此谐振电路无法同时符合电池高电压输入和放电一 段时间后低电压输入的要求， 无法维持最佳的谐振效果。 因此， 使得当电 池输入电压较高时， duty变小， 谐振效果不佳， Q1和 Q2开关进入硬开关 模式时， 开关损耗大， 同时导通损耗也非常大。
图 4说明了在不同的输入电压下图 1和图 2的功率暂态效率的比较图。 图 4 中的横坐标表示输入电池电压， 从左向右依次递增， 最低电压为 60 伏， 最高电压为 84伏， 其中每个坐标递增电压 2伏。 纵坐标表示输出功率 与输入功率的比值， 以百分率表示， 最低比值为 79%， 最高比值为 88%， 其中每个坐标递增 1%。 矩形点表示不同的输入电压下图 2所示电路的功 率暂态效率值， 菱形点表示不同的输入电压下图 1所示电路的功率暂态效 率值。
如图 4所示， 在输入电池电压从 84伏到 70伏的区间内， 图 1所示的 电路和图 2所示的电路的输出功率与输入功率的比值， 均随着输入电池电 压的降低而升高。 在输入电池电压从 70伏到 60伏的区间内， 图 1所示的 电路的输出功率与输入功率的比值， 随着输入电池电压的降低而降低。 在 输入电池电压为 68伏， 66伏， 64伏和 62伏的时， 图 2所示电路的输出功 率与输入功率的比值相同，均小于输入电池电压为 70伏的输出功率与输入 功率的比值， 均大于输入电池电压为 60伏的输出功率与输入功率的比值。 输入电池电压为 70伏时，两者的输出功率与输入功率的比值分别达到 各自的最大值， 而输入电池电压为 84伏时， 两者的输出功率与输入功率的 比值分别达到各自的最小值。 在输入电池电压为 84伏和 82伏时， 图 1所 示的电路的输出功率与输入功率的比值要大于图 2所示的电路的输出功率 与输入功率的比值。 而在其余各个点上， 图 1所示的电路的输出功率与输 入功率的比值均小于图 2所示的电路的输出功率与输入功率的比值。
由此可知， 在大部分情况下， 图 2所示的准谐振推挽式变换器的效率 比图 1所示的推挽式变换器要高， 但是在电池输入高压时， 例如 70伏至 84伏区间， 准谐振推挽式变换器的效率仍然不高。 因此， 需要提供一种在 输入的直流电源的电压范围较宽时仍能提供较高效率的变换器。
图 5根据本发明实施方式的准谐振推挽式变换器的结构图。 推挽变换 器的原边侧包括： 直流输入电源， MOSFET开关管 Q1和 Q2， 输入电容 Cin以及匝数 N1和 N2的一次绕组。 其连接关系为： 直流输入电源的正极 通过匝数 N1的一次绕组的连接 MOSFET开关管 Q1的源极， MOSFET 开关管 Q1的漏极接地；直流输入电源的正极通过匝数 N2的一次绕组的连 接 MOSFET开关管 Q2的源极， MOSFET开关管 Q2的漏极接地； 以及 输入电容 Cin与直流输入电源并联， 一端连接到匝数 N1的一次绕组的同 名端， 另一端与地线连接。
推挽变换器的副边侧包括： 匝数分别为 N3， N4, N5和 N6的二次绕 组， 第一全桥整流器， 第二全桥整流器， 第一和第二开关， 谐振电容 C3 , 电感 L2和 L3， 输出电容 Col和 Co2。 其中， 第一全桥整流器包括： 二极 管 Dl， D2, D5和 D6; 第二全桥整流器包括： D3 , D4, D7和 D8。 第一 全桥整流器连接匝数为 的二次绕组的另一端和匝数为 N6的二次绕组的 同名端； 第二全桥整流器连接匝数为 N3的二次绕组的同名端和匝数为 N6 的二次绕组的另一端。
具体连接关系为： 匝数为 N3 的二次绕组的另一端连接变压器等效漏 感 L1，的一端， 变压器漏感 L1，的另一端连接二极管 D1的阳极和 D5的阴 极， 匝数为 N3的二次绕组的同名端连接变压器等效漏感 L1的一端， 变压 器漏感 L1的另一端连接二极管 D3的阳极，二极管 D7的阴极和匝数为 N4 的二次绕组的另一端。匝数为 N4的二次绕组的同名端与匝数为 N5的二次 绕组的另一端均接地，匝数为 N5的二次绕组的同名端连接二极管 D4的阳 极， 二极管 D8的阴极和匝数为 N6的二次绕组的另一端； 匝数为 N6的二 次绕组的同名端连接二极管 D2的阳极和二极管 D6的阴极。
二极管 D1的阴极连接到二极管 D2的阴极， 开关 K1的一端。 可替换 地， 使用其它能够实现开关功能的电路来替换开关 Kl。 开关 K1的另一端 连接二极管 D3和 D4的阴极， 谐振电容 C3的一端， 电感 L2的一端； 电 感 L2的另一端连接输出电容 Col的正极， 输出电容 Col的负极接地。
二极管 D5的阳极连接到二极管 D6的阳极， 开关 K2的一端。 可替换 地， 使用其它能够实现开关功能的电路来替换开关 K2。 开关 Κ2的另一端 连接二极管 D7和 D8的阳极， 谐振电容 C3的另一端， 电感 L3的一端； 电感 L3的另一端连接输出电容 Co2的负极， 输出电容 Co2的正极接地。
通过控制开关电路 K1和 K2的同时导通和截止，图 6所示的集成谐振 推挽变换器能够实现谐振的效果维持在良好的状态。当 K1和 K2均截止时， 副边工作的二次绕组为 N4和 N5。 本发明的优选实施方式增加一组变压器 的二次侧的圏数 N3及 N6， 并以输出低压的状态来设计谐振电路； 当开关 K1和 K2均导通时， 副边工作的匝数增加为 (N3+N4)以及 (N5+N6)。 由此 可知， 通过本发明的改进实施方式， 在谐振推挽变换器中能够控制输出电 压与输出电压的不同比值， 以满足不同的需要， 使得输入高压的谐振效果 像输入 时一样良好。
当电池为高电压输入时， 例如， 当直流输入电压高于一设定值 Θ时， 使第一开关 K1和第二开关 K2均截止， 副边工作的二次绕组为 N4和 N5， 如此让 Q1和 Q2开关的 duty变大， 谐振电容 C3和漏感 L1谐振， Q1和 Q2开关进入谐振模式； 当电池电压因放电而下降至一设定值时，例如， 当 直流输入电压低于该设定值 Θ时，将第一开关 K1和第二 K2均导通，副边 工作的匝数增加为 (N3+N4)以及 (N5+N6)，改变原边绕组和副边绕组的匝数 比， 以保持输出电压的值， 谐振电容 C3和漏感 L1，谐振， Q1和 Q2开关同 样操作在谐振模式。 如此以使整个电池放电的过程中， Q1和 Q2开关谐振 的效果维持在良好的状态， 从而提高整体的放电效率。
图 6说明了在不同的输入电压下图 1、 图 2和图 5的功率暂态效率的 比较图。 图 6的横坐标表示输入电池电压， 从左向右依次递增， 最低电压 为 60伏， 最高电压为 84伏， 其中每个坐标递增电压 2伏。 纵坐标表示输 出功率与输入功率的比值， 以百分率表示， 最低比值为 79%， 最高比值为 88%, 其中每个坐标递增 1%。 三角形点表示不同的输入电压下图 5所示 电路的功率暂态效率值， 矩形点表示不同的输入电压下图 2所示电路的功 率暂态效率值， 菱形点表示不同的输入电压下图 1所示电路的功率暂态效 率值。
如图 6所示， 在输入电池电压从 84伏到 70伏的区间内， 图 1所示的 电路和图 2所示的电路的输出功率与输入功率的比值， 均随着输入电池电 压的降低而升高。 在输入电池电压从 70伏到 60伏的区间内， 图 1所示的 电路的输出功率与输入功率的比值， 随着输入电池电压的降低而降低。 在 输入电池电压为 68伏， 66伏， 64伏和 62伏的时， 图 2所示电路的输出功 率与输入功率的比值相同，均小于输入电池电压为 70伏的输出功率与输入 功率的比值， 均大于输入电池电压为 60伏的输出功率与输入功率的比值。 在输入电池电压从 84伏到 76伏的区间内， 图 5所示电路的输出功率与输 入功率的比值， 随着输入电池电压的降低而升高， 在输入电池电压从 76 伏到 70伏的区间内， 图 5所示的电路的输出功率与输入功率的比值， 随着 输入电池电压的降低而降低， 在输入电池电压从 70伏到 60伏的区间内， 图 5所示电路的输出功率与输入功率的比值， 与图 2所示电路的输出功率 与输入功率的比值相同。
输入电池电压为 70伏时，图 1所示的电路和图 2所示的电路的输出功 率与输入功率的比值分别达到各自的最大值， 而输入电池电压为 76伏时， 图 5所示电路的输出功率与输入功率的比值达到最大值。 输入电池电压为 84伏时， 三者的输出功率与输入功率的比值分别达到各自的最小值。 在输 入电池电压为 84伏和 82伏时， 图 1所示的电路的输出功率与输入功率的 比值要大于图 2所示的电路的输出功率与输入功率的比值。 而在其余各个 点上， 图 1所示的电路的输出功率与输入功率的比值均小于图 2所示的电 路的输出功率与输入功率的比值。
在输入电池电压从 70伏到 60伏的区间内， 图 5所示电路的输出功率 与输入功率的比值， 与图 2所示电路的输出功率与输入功率的比值相同。 在输入电池电压从 84伏到 70伏的区间内， 图 5所示电路的输出功率与输 入功率的比值要大于图 1所示的电路和图 2所示的电路的输出功率与输入 功率的比值。
由此可知， 在大部分情况下， 图 5所示的准谐振推挽式变换器的效率 比图 1和图 2所示的推挽式变换器要高， 集成的准谐振推挽式变换器实现 了更高的效率。
从图 6可以看出， 整个 Push-Pull谐振电路的效率不因输入电池高低压的 原因而使效率变差， 效率都维持在相当的高点， 整个 Battery mode (电池 模式）的整体效率因此提高。

Claims

权 利 要 求
1. 一种准谐振推挽变换器的控制方法，应用于具有谐振电容的推挽变 换器中， 所述推挽变换器包括： 直流输入电源， 用于为变换器提供直流输 入； 第一和第二功率输入单元， 分别连接所述直流输入电源， 用于在不同 的周期内分别为变换器提供输入， 其包括第一和第二功率开关管， 第一原 级绕组和第二原级绕组； 功率输出电路， 用于提供变换器的输出， 其包括 次级绕组和全桥整流电路； 第一和第二输出电容， 连接于所述功率输出电 路， 用于储存功率输出电路所输出的直流电能， 其特征在于，
在功率输出电路中设置谐振元件， 通过电压反馈来实现准谐振开关电 路； 以及
通过电压反馈来控制一开关电路， 从而控制推挽变换器的原级绕组和 次级绕组的匝数比。
2.根据权利要求 1所述的方法， 其特征在于， 所述在功率输出电路中 设置谐振元件具体为： 在功率输出电路中设置变压器漏感、 谐振电容以及 第一和第二电感。
3.根据权利要求 2所述的方法， 其特征在于， 还包括在功率输出电路 中设置第一全桥整流电路、 第二全桥整流电路和开关元件， 通过控制开关 元件的导通或截止来控制次级绕组的工作匝数， 改变原级绕组和次级绕组 的匝数比， 以保持谐振电路的工作不因输入电压的高低而不同， 边的 第一和第二功率开关管均工作在谐振模式下的开关模式。
4.根据权利要求 2或 3所述的方法， 其特征在于， 所述通过电压反馈 来控制一开关电路具体为： 设定一电压值， 当侦测到直流输入电压高于该 设定值时， 将开关电路设置为截止； 当侦测到直流输入电压低于该设定值 时， 将该开关电路设置为导通， 从而控制次级绕组的工作匝数。
5. 一种准谐振推挽变换器， 其特征在于， 包括：
直流输入电源， 用于为变换器提供直流输入； 第一和第二功率输入单元， 分别连接所述直流输入电源， 用于在不同 的周期内分别为变换器提供输入， 其包括第一和第二功率开关管， 第一原 级绕组和第二原级绕组；
功率输出电路， 用于提供变换器的输出， 其包括次级绕组和全桥整流 电路；
第一和第二输出电容， 连接于所述功率输出电路， 用于储存功率输出 电路所输出的直流电能， 其特征在于，
功率输出电路还包括变压器漏感、 谐振电容和电感， 通过电压反馈来 实现准谐振开关电路； 以及
通过电压反馈来控制一开关电路， 从而控制推挽变换器的原级绕组和 次级绕组的匝数比。
6.根据权利要求 5所述的准谐振推挽变换器， 其特征在于， 还包括在 功率输出电路中设置第一全桥整流电路、 第二全桥整流电路和开关元件， 通过控制开关元件的导通或截止来控制次级绕组的工作匝数， 改变原级绕 组和次级绕组的匝数比， 以保持谐振电路的工作不因输入电压的高低而不 同， 使原边的第一和第二功率开关管均工作在谐振模式下的开关模式。
7.根据权利要求 6所述的准谐振推挽变换器， 其特征在于， 所述开关 元件包括第一开关和第二开关， 所述第一开关和第二开关分别连接于所述 第一全桥整流电路和第二全桥整流电 之间。
8.根据权利要求 5所述的准谐振推挽变换器， 其特征在于， 其中谐振 频率设置为开关频率的 2倍。
9.根据权利要求 7所述的准谐振推挽变换器， 其特征在于， 所述次级 绕组包括第一次级绕组、 第二次级绕组、 第三次级绕组和第四次级绕组， 当第一开关和第二开关均导通时， 由第一次级绕组串联第二次级绕组和第 三次级绕组串联第四次级绕组提供输出；当第一开关和第二开关均截止时， 由第二次级绕组和第三次级绕组提供输出。
10.根据权利要求 7所述的准谐振推挽变换器， 其特征在于， 还包括 设定一电压值， 当直流输入电压高于一该设定值时， 则将第一开关和第二 开关均设置为截止； 当直流输入电压低于该设定值时， 则将第一开关和第 二开关均设置为导通。
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