WO2015106701A1 - 一种交流-直流变换电路及其控制方法 - Google Patents

一种交流-直流变换电路及其控制方法 Download PDF

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Publication number
WO2015106701A1
WO2015106701A1 PCT/CN2015/070808 CN2015070808W WO2015106701A1 WO 2015106701 A1 WO2015106701 A1 WO 2015106701A1 CN 2015070808 W CN2015070808 W CN 2015070808W WO 2015106701 A1 WO2015106701 A1 WO 2015106701A1
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Prior art keywords
circuit
input
inverter
isolation transformer
secondary side
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PCT/CN2015/070808
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English (en)
French (fr)
Inventor
李伦全
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深圳市保益新能电气有限公司
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Publication of WO2015106701A1 publication Critical patent/WO2015106701A1/zh
Priority to US15/202,576 priority Critical patent/US9748854B2/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4216—Arrangements for improving power factor of AC input operating from a three-phase input voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4258—Arrangements for improving power factor of AC input using a single converter stage both for correction of AC input power factor and generation of a regulated and galvanically isolated DC output voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/01—Resonant DC/DC converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33571—Half-bridge at primary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02P—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN THE PRODUCTION OR PROCESSING OF GOODS
    • Y02P80/00—Climate change mitigation technologies for sector-wide applications
    • Y02P80/10—Efficient use of energy, e.g. using compressed air or pressurized fluid as energy carrier

Definitions

  • the invention relates to a switching power supply, in particular to an AC-DC conversion circuit and a control method thereof.
  • PF aligners Single-stage power factor (PF) aligners, active clamp flybacks, active clamp forwards, and improvements thereof, which are soft switching technologies, have been widely used in power supplies, and one disclosed in US Pat. No. 7,301,785 B2.
  • the switching power supply circuit the voltage stress of the switching tube on the primary side of the converter circuit varies with the load condition, and is highly offset at full load, which does not fundamentally overcome the shortcomings of the typical resonance technology.
  • the range of available power ranges, input and output voltage variations is limited.
  • Chinese patent document CN101692595B discloses an active clamp positive flyback circuit which is improved compared to other already disclosed circuits, but its primary side does not require a large capacity electrolytic capacitor because of the need for AC rectification filtering.
  • the actual use still requires additional high frequency filter capacitors, and the AC input side must also use full bridge rectification.
  • the secondary side rectification circuit is complicated, and whether it is an input rectification circuit or an output rectification circuit, a plurality of diodes have large series loss.
  • the output voltage and current ripple are large; in practical use, it is difficult to apply to a large-power AC-DC converter circuit, and it is generally suitable to apply to a DC-DC converter.
  • the clamp circuit may increase the loss due to the unsatisfactory resonance condition in the low-voltage input portion, and reduce the loss. The stability and reliability of the circuit.
  • the technical problem to be solved by the present invention is to make up for the deficiencies of the above prior art, and to propose an AC-DC conversion circuit and a control method thereof.
  • the present invention adopts the following technical solutions:
  • An AC-DC conversion circuit comprising an input rectification circuit, a primary input filter capacitor, a primary side inverter circuit, a drive circuit, an isolation transformer, a secondary side rectifier circuit, a third side to a fourth side capacitor, and a secondary side output filter capacitor And a control circuit connected between the secondary side output filter capacitor and the driving circuit, the driving circuit is connected to the primary side inverter circuit;
  • the input rectifying circuit includes first to second input rectifying diodes, and the first input is integrated
  • the anode of the flow diode and the cathode of the second input rectifier diode are connected to an AC input live line
  • the primary input filter capacitor comprises two input high frequency filter capacitors connected in series with the AC input neutral line as a midpoint, and two input high frequency
  • the opposite ends of the filter capacitors are respectively connected to the positive terminal and the negative terminal of the input rectifier circuit;
  • the primary side inverter circuit includes first to second inverter switch tubes, and the first inverter switch tube and the second inverter switch tube are respectively connected to a positive end and a negative end of the input rectifier circuit, One end of the primary winding of the isolation transformer is connected in series with the first and second inverter switching tubes, and the other end of the primary winding of the isolation transformer is connected to the input neutral line; the primary side inverter circuit and the primary input filter capacitor Forming a loop to form a clamp resonant circuit, the inverter switch tube operating in a zero voltage switching state;
  • the secondary side rectifier circuit includes third to fourth diodes, and a cathode of the third diode is connected to an anode of the fourth diode and one output end of the secondary winding of the isolation transformer, a cathode of the fourth diode is connected to an output positive end of the power supply and an end of the secondary output filter capacitor, an anode of the third diode and a negative output end of the power supply, and the secondary output filter capacitor Connected to the other end, the other output end of the secondary winding of the isolation transformer is connected to an intermediate point of the third to fourth capacitors of the secondary side, and the secondary capacitor of the secondary side is connected to the output positive end of the power supply, the secondary side
  • the four capacitors are connected to the output negative terminal of the power supply; the secondary side rectifier circuit has two working modes of forward and reverse.
  • the isolation transformer is an isolation transformer in which the magnetic core is an air gap or an isolation transformer in which a primary side has a resonant inductor or an isolation transformer in which a secondary side has a storage inductor.
  • the isolation transformer is a transformer of a single secondary winding multiplexed between a forward working coil and a flyback working coil.
  • the third and fourth capacitors of the secondary side are two series-connected infinite capacitors or polar capacitors, and in the case of a polar capacitor, the anode of the secondary side third capacitor is connected to the output positive terminal of the power source, The negative terminal of the fourth capacitor of the secondary side is connected to the output negative terminal of the power supply.
  • An AC-DC conversion circuit comprising two, three or more of the aforementioned AC-DC conversion circuits, the input of each AC-DC conversion circuit being connected to an AC having two, three or more phase inputs Different phase inputs of the source.
  • each AC source has an input zero line and a midpoint of the primary side input filter capacitor of each AC-DC conversion circuit is connected to the input zero line, or each AC source has no input zero line and each intersection The midpoint of the primary input filter capacitor of the current-to-dc converter circuit is connected to the same point.
  • a control method of the AC-DC conversion circuit the circuit has two working modes of forward and reverse excitation, and the first and second inverter switching tubes in the circuit are used as an inverter switch tube according to a working period Or a clamp switch tube is used.
  • the primary side input filter capacitor in the circuit Used as an input filter capacitor or a clamp capacitor according to the working period
  • the primary input filter capacitor resonates with the leakage inductance of the isolation transformer in the circuit when the first or second inverter switch is turned off, so that the clamp is clamped
  • the switch tube and the inverter switch tube obtain zero voltage switching, and the energy of the leakage inductance of the isolation transformer is transmitted to the secondary side of the isolation transformer by resonance.
  • the operating states of the first and second inverter switching tubes are controlled by a control circuit to make the input current coincide with the fundamental wave of the input voltage.
  • the clamp resonant circuit in the circuit adopts the following two control modes in the full cycle of the AC rectification: no driving signal is generated in a low voltage phase lower than the set voltage threshold, and a high voltage phase not lower than the set voltage threshold The drive signal is sent to perform the reverse resonance of the clamp current.
  • the invention provides a single-phase or multi-phase AC input single-stage high power factor wide range AC-DC conversion circuit.
  • the input filter capacitor in addition to being used as an input filtering function, the input filter capacitor also functions as a clamp function to assist in realizing the soft switching operation of the primary side inverter switch tube, and the inverter switch tube has a function other than the inverter switch function. Also acts as a clamp switch. That is, the circuit of the present invention can fully utilize the difference of the switching working state of the positive and negative half cycles of the AC input, and multiplex the negative (positive) terminal switching transistor and the negative (positive) terminal input filter capacitor to realize the clamp switch tube and the clamp capacitor.
  • the function similar to three-level rectification, reduces diode losses in the rectifier loop.
  • the primary input filter capacitor can resonate with the leakage inductance of the isolation transformer when the first or second inverter switch is turned off, so that the clamp switch tube and the inverter switch tube obtain zero voltage switching (ZVS), and the resonance will be high.
  • the energy of the leakage inductance of the frequency isolation transformer is transmitted to the secondary side, avoiding the energy loss of the leakage inductance and instantaneously causing the voltage spike of the inverter switch tube.
  • the invention can reduce the diode loss in the rectification loop, realize the soft switching of the positive (negative) end inverter switch tube, and reduce the voltage stress and the switching loss of the primary side switch tube and the secondary side rectifier diode.
  • the double-voltage rectification of the secondary side is used to reduce the voltage stress of the secondary side rectifier diode, and at the same time, the forward energy and the flyback energy form different loops, thereby subtly implementing a similar power factor correction similar energy transfer.
  • the output voltage is a proportional input voltage (excited energy) plus a proportional inductor stored in energy (flyback energy).
  • the clamp circuit is divided into the full cycle of AC rectification.
  • the two control modes that is, no driving signal is sent at the input low voltage part, and the driving signal is sent to the reverse resonance of the clamp current in the relatively high voltage part, effectively controlling the loss of the clamp circuit and improving the reliability and stability of the circuit. .
  • the invention is particularly suitable for applications where three-phase or multi-phase AC input is high and the output voltage is high, the semiconductor device cannot withstand high voltage, and where power factor, power density and volume constraints are strict.
  • Embodiment 1 is a circuit diagram of Embodiment 1 of an AC-DC conversion circuit of the present invention
  • FIG. 2 is a schematic diagram of a transformer equivalent structure of the circuit shown in FIG. 1;
  • Figure 3 is a timing diagram of driving of the circuit switch of Figure 1;
  • Embodiment 2 is a circuit diagram of Embodiment 2 of the AC-DC conversion circuit of the present invention.
  • Fig. 5 is a circuit diagram showing a third embodiment of the AC-DC conversion circuit of the present invention.
  • the AC-DC conversion circuit shown in FIG. 1 includes an input rectification circuit, a primary input filter capacitor C1, C2, a primary side inverter circuit, a drive circuit, a high frequency isolation transformer T1, a secondary side rectifier circuit, and a secondary side third. a fourth capacitor C3, C4, a secondary output filter capacitor C5, and a control circuit connected between the secondary output filter capacitor C5 and the drive circuit.
  • the input rectifying circuit comprises two input rectifying diodes D1 and D2.
  • the anode of the input rectifying diode D1 and the cathode of the input rectifying diode D2 are connected to the AC input live line L.
  • the primary input input filter capacitor includes a neutral input zero line N as a midpoint.
  • Two input high frequency filter capacitors C1 and C2 connected in series, and opposite ends of the two input high frequency filter capacitors C1 and C2 are respectively connected to the positive terminal and the negative terminal of the input rectifier circuit.
  • the primary side inverter circuit includes first and second inverter switching tubes Q1 and Q2, and the first and second inverter switching tubes Q1 and Q2 are respectively connected to a positive end and a negative end of the input rectifying circuit.
  • One end of the primary winding of the isolation transformer T1 is connected in series with the first and second inverter switching tubes Q1 and Q2, and the other end is connected to the AC input neutral line N.
  • the primary side inverter circuit and the primary input input filter capacitor C1 and C2 also form a loop to form a clamp resonant circuit, and the inverter switching transistors Q1 and Q2 operate in a ZVS state.
  • the DC voltage can be converted into a pulse voltage and applied to the primary winding of the isolation transformer.
  • the on/off of the resonant current reverse loop can be controlled by controlling the switching tubes of the inverter switching tubes Q1 and Q2 that function as clamps.
  • the secondary side rectifier circuit is a forward and flyback working rectifier circuit, including a third diode D3 and a fourth diode D4, a cathode of the third diode D3 and the fourth diode D4 An anode is connected to one of the output ends of the secondary winding of the isolation transformer, and a cathode of the fourth diode D4 is connected to an output positive terminal V+ of the power supply and an end of the secondary output filter capacitor C5.
  • the anode of the three diode D3 is connected to the output negative terminal V- of the power supply and the other end of the secondary output filter capacitor C5, and the other output terminal of the secondary winding of the isolation transformer and the third and fourth capacitors of the secondary side C3, C4 intermediate point connection, the positive side of the secondary side third capacitor C3 (when using a polar capacitor) is connected to the output positive terminal V+ of the power supply, and the secondary side of the secondary capacitor C4 is negative (when the pole is used When the capacitor is used, the output terminal of the power supply is connected.
  • the third and fourth capacitors C3 and C4 of the secondary side may also have a polar capacitor.
  • the capacitance of the forward working circuit and the voltage of the flyback working circuit have the same or complementary waveform as the input AC rectified waveform, and the voltage of the forward loop capacitor and the input voltage of the primary side The change has an approximately linear relationship.
  • the operating states of the first and second inverter switching tubes are controlled by the control circuit to make the input current coincide with the fundamental wave of the input voltage, thereby achieving high input power factor correction.
  • the input circuit rectifies the AC voltage, the input high-frequency filter capacitors C1 and C2 have a small capacity, and the parameters are mainly determined by the resonant frequency used for clamping. Therefore, this circuit ensures that the input current can conditionally follow the input voltage to ensure the input power factor and total harmonic content of the power supply (Total Harmonics Distortion, THD for short).
  • the inverter circuit is composed of an inverter switch tube Q1 (Q2) and an isolation transformer T1.
  • the primary side clamp resonant circuit is composed of clamp capacitor C2 (C1) and clamp switch tube Q2 (Q1).
  • a high-frequency PWM signal is applied to the inverter switching transistor Q1 through the driving circuit, and the driving voltage on the clamp switch Q2 is a PWM voltage which is approximately complementary to Q1. There is a certain dead zone delay relationship between the two PWM drive voltages.
  • the isolation transformer is an isolation transformer in which the magnetic core is an air gap or an isolation transformer in which a primary side has a resonant inductor or an isolation transformer in which a secondary side has a storage inductor.
  • the size of the air gap of the magnetic core is proportional to the forward and reverse excitation. Together with the system input and output parameters, the original and secondary side coupling coefficients do not need to be specially set.
  • the core of the isolation transformer T1 has an air gap and a certain leakage inductance, so that the isolation transformer T1 can work in both forward and flyback states.
  • the leakage inductance is obtained by a natural winding process, and at the same time, according to actual needs, a leakage feeling that can be large or small can be obtained by a change in the winding process.
  • an inductor can be applied to the secondary side.
  • the isolation transformer does not deliberately distinguish the end point connection points of the primary side and the secondary side, that is, the starting end of the isolation transformer is not intentionally considered.
  • the primary exciting inductance Lm and the leakage inductance Lr on the primary side are determined.
  • the leakage inductance of the primary side and the resonant frequency of the resonant capacitor C2 (C1) satisfy the following relationship with respect to the switching frequency:
  • the input is rectified by diode D1, and then capacitor C1 is subjected to high frequency filtering.
  • the control circuit calculates the result, and applies a high frequency PWM signal to the inverter switch Q1 through the drive circuit.
  • the inverter switch Q1 When the inverter switch Q1 is turned on, the transformer primary excitation inductance Lm and the resonance inductance Lr1 start linear charging.
  • the primary current is equal to the excitation current
  • the parasitic capacitance of the inverter switch Q1 When the inverter switch Q1 is turned off, the parasitic capacitance of the inverter switch Q1 is charged, and the charging process is also resonant, except that the parasitic capacitance is small and the charging time is short, which can be regarded as linear.
  • the secondary leakage inductance or the potential VLr2 of the external inductor Lr2 is deflected, trying to maintain the original current direction and size, but as time passes, the current through the inductor or the rectifier diode D4 necessarily begins to decrease.
  • clamp switch Q2 When the parasitic capacitor voltage of the inverter switch Q1 is charged high enough to be about the voltage VC2+Vin, the clamp switch Q2 is reversed and the diode is forward-biased. Clamp capacitor C2 clamps the voltage of resonant inductor Lr1 and magnetizing inductor Lm to voltage VC2, because clamp capacitor C2 is much larger than the parasitic capacitance of inverter switch Q1, most of the resonant current enters clamp capacitor C2, clamp The capacitor C2 and the resonant inductor Lr1 start to resonate; when the current of the primary side is equal to the exciting current, the output current of the secondary side of the transformer is equal to zero, and the potential of the secondary side coupling voltage V2 alternates.
  • the coupling voltage V2 corresponding to the secondary side is sufficient to cause the diode D3 to be forward biased.
  • the voltage reflected from the secondary side to the primary side is about n (Vo-Vc3-VLr2), which provides a condition for the clamp switch Q2 to obtain ZVS.
  • the driving voltage of the clamp switch Q2 will become a high level. And turned on.
  • the energy cup originally stored in the transformer air gap is released. This state is a typical flyback transformer operation. Since the voltage across capacitor C4 is linear with the input voltage, the voltage across capacitor C3 is combined with the voltage across capacitor C4 to synthesize the output voltage.
  • the driving voltage of the inverter switching transistor Q1 is turned to a high level and turned on; the inductance of the primary side is linearly charged, a new cycle is started, and the above state process is repeated.
  • FIG. 3 it is a circuit breaker driving timing diagram of the present invention.
  • this circuit when the input voltage is in the first and third regions, since the duty ratio of the inverter switch tube is large, the inverter switch tube is During the turn-off period, the time is short, and the aforementioned resonant current has not yet reached the reverse direction. At the same time, since the input voltage is low, the voltage stress on the semiconductor element in the circuit is small. In order to avoid the uncertainty of the resonance state and the unreliability of the circuit, it is preferable to explicitly pass the first setting.
  • the input voltage limit of the third region turns off the driving of the clamp tube for a long time, so that the clamp capacitance value acts as a common peak absorption function, and also reduces the driving loss; when the input voltage is in the second and fourth regions, the clamp
  • the bit switch tube Q2 (or Q1) works according to the above clamp method, effectively improving the stability and reliability of the circuit.
  • the isolation transformer is like a linear transformer, and the linear ratio of the input voltage is reduced to the capacitor C4, and the isolation transformer is better than the conventional one.
  • the PFC inductor in the power factor correction circuit releases the energy stored in the turn-on of the switch to the capacitor C3 when the switch is turned off. Therefore, the voltage on the capacitors C3 and C4 constitutes a linear ratio and a conventional non-isolated PFC voltage. . Therefore, the circuit well implements an isolated PFC that is not available in a single-stage isolated AC-DC converter circuit of known advantages.
  • the inverter circuit is composed of the inverter switch tube Q2 and the primary transformer of the isolation transformer T1.
  • the primary clamp clamp resonance circuit is composed of the clamp capacitor C1, the clamp switch tube Q1 and the transformer primary coil.
  • the voltage waveforms of the capacitors C3 and C4 are also symmetrical, the capacitor C3 acts as the output capacitor of the forward loop, and C4 acts as the output capacitor of the flyback loop.
  • Other working state principles and control methods are consistent with the aforementioned positive half wave.
  • the primary inverter switch tube and the high-frequency filter capacitor are subtly multiplexed into the clamp switch tube and the clamp capacitor when the input voltage is positive and negative for half a cycle, and the input rectifier circuit only A diode is required, and the output only needs to pass through a diode to form a rectification loop. Therefore, the circuit of the circuit is simple and efficient.
  • the invention also provides an AC-DC conversion circuit with two-phase, three-phase or more phase inputs.
  • a three-phase four-wire input AC-DC conversion circuit has the same basic circuit composition and beneficial effects as the first embodiment. The difference is that the input is three-phase.
  • the advantage is that theoretically, an output voltage better than that of the first embodiment can be obtained, the output voltage is smoother, and the ripple voltage is smaller.
  • D1a, D2a, D1b, D2b, D1c, and D2c represent primary side rectifier diodes
  • C1a, C2a, C1b, C2b, C1c, and C2c represent primary side filter capacitors
  • Q1a, Q2a, Q1b, Q2b, Q1c, and Q2c represent Primary inverter switch
  • T1a, T1b, T1c represent transformers
  • D3a, D4a, D3b, D4b, D3c, D4c represent secondary rectifier diodes, C3a, C4a, C5b, C3b, C4b, C5b, C3c, C4c, C5c Secondary filter capacitor.
  • the present invention also provides an AC-DC conversion circuit of another two-phase, three-phase or more phase input.
  • a three-phase three-wire input AC-DC conversion circuit has the same basic circuit composition and beneficial effects as the second embodiment. The difference is that the input has no zero line input.
  • the advantage is that the above performance can still be achieved in the actual three-phase three-wire zero-line input environment.
  • the circuit of the invention has two different working modes of forward and flyback, and can realize a large input and output voltage adjustment range, and the input current can also track the input voltage to achieve power factor correction.
  • the circuit due to the function of the active clamp circuit, the reverse recovery voltage spike and switching loss caused by the leakage inductance of the isolation transformer and the secondary side rectifier diode can be reduced, and the efficiency is improved; the circuit is more compact than the conventional circuit;
  • the multi-state clamp control mode method can effectively control the loss of the clamp circuit; the invention is particularly suitable for applications in three-phase (multi-phase) AC input and high output voltage.

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  • Power Engineering (AREA)
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Abstract

一种交流－直流变换电路及其控制方法。该交流一直流变换电路包括输入整流电路（D1，D2）、原边输入滤波电容（C1，C2）、原边逆变电路（Q1，Q2）、驱动电路、隔离变压器（T1）、副边整流电路（D3，D4）、副边电容（C3，C4）、副边输出滤波电容（C5）以及连接在副边输出滤波电容与驱动电路之间的控制电路。驱动电路连接原边逆变电路。原边逆变电路与原边输入滤波电容构成回路，形成箝位谐振电路，原边逆变电路的逆变开关管（Q1，Q2）工作在零电压切换状态。副边整流电路具有正激及反激两种工作模式，原边逆变电路中的逆变开关管根据工作时段作为逆变开关管或箝位开关管使用。交流－直流变换电路及其控制方法能减少整流回路中的二极管损耗，实现逆变开关管的软开关，降低原边开关管及副边整流二极管的电压应力和开关损耗，提高效率。

Description

一种交流-直流变换电路及其控制方法 技术领域
本发明涉及开关电源，特别是一种交流-直流变换电路及其控制方法。
背景技术
属于软开关技术的单级功率因数(Power Factor，简称PF)矫正器、有源箝位反激、有源箝位正激及其改进方案已经在电源中广泛运用，美国专利文献US7301785B2公开的一种开关电源电路，该变换器电路原边的开关管的电压应力会随着负载的状况而变化，且在满载时会偏移到很高，未从根本上克服典型谐振技术的缺点，因此，其可用的功率范围、输入及输出电压变化范围受到限制。中国专利文献CN101692595B公开了一种有源箝位正反激电路，该电路相比其他已经公开的电路有所改进，但其一次侧因为交流整流滤波的需要，虽然可以不需要大容量的电解电容，但实际使用还是需要额外添加的高频滤波电容，交流输入侧也必须运用全桥整流。二次侧整流电路复杂，无论是输入整流回路还是输出整流回路，多个二极管串联损耗较大。同时其在无大容量的输入电解电容时，输出电压电流纹波都很大；在实际使用中难以运用到较大功率的交流-直流变换电路场所，一般运用到直流-直流变换场所较为合适。此外，该变换器在输入是交流时，由于输入电压的周期性变化引起对应驱动占空比的变化，使得箝位电路在低压输入部分会因为谐振条件不满足而出现损耗加大的情况，降低了电路的稳定性、可靠性。
发明内容
本发明所要解决的技术问题是弥补上述现有技术的不足，提出一种交流-直流变换电路及其控制方法。
为实现上述目的，本发明采用以下技术方案：
一种交流-直流变换电路，包括输入整流电路、原边输入滤波电容、原边逆变电路、驱动电路、隔离变压器、副边整流电路、副边第三至第四电容、副边输出滤波电容以及连接在所述副边输出滤波电容与所述驱动电路之间的控制电路，所述驱动电路连接所述原边逆变电路；
所述输入整流电路包括第一至第二输入整流二极管，所述第一输入整 流二极管的阳极和所述第二输入整流二极管的阴极连接交流输入火线，所述原边输入滤波电容包括以交流输入零线为中点串联的两个输入高频滤波电容，两个输入高频滤波电容的相反端分别与所述输入整流电路的正端及负端连接；
所述原边逆变电路包含第一至第二逆变开关管，所述第一逆变开关管和所述第二逆变开关管分别与所述输入整流电路的正端和负端连接，所述隔离变压器原边绕组的一端与第一、第二逆变开关管串联，所述隔离变压器原边绕组的另一端与输入零线连接；所述原边逆变电路与原边输入滤波电容也构成回路，形成箝位谐振电路，所述逆变开关管工作在零电压切换状态；
所述副边整流电路包括第三至第四二极管，所述第三二极管的阴极与所述第四二极管的阳极及所述隔离变压器副边绕组的其中一输出端相连，所述第四二极管的阴极与电源的输出正端以及所述副边输出滤波电容的一端连接，所述第三二极管的阳极与电源的输出负端以及所述副边输出滤波电容的另一端相连，所述隔离变压器副边绕组的另一个输出端与副边第三至第四电容的中间点连接，所述副边第三电容连接电源的输出正端，所述副边第四电容连接电源的输出负端；所述副边整流电路具有正激及反激两种工作模式。
进一步地：
所述隔离变压器是磁芯是开有气隙的隔离变压器或原边串联有谐振电感的隔离变压器或副边串联有储能电感的隔离变压器。
所述隔离变压器是正激工作线圈与反激工作线圈复用的单个副边绕组的变压器。
所述副边第三、第四电容是两个串联的无极电容或者有极性电容，且在为有极性电容的情况下，所述副边第三电容的正极接电源的输出正端，所述副边第四电容的负极接电源的输出负端。
一种交流-直流变换电路，包括2个、3个或者更多个前述的交流-直流变换电路，各交流-直流变换电路的输入连接到具有2个、3个或者更多个相位输入的交流源的不同相位输入。
进一步地，各交流源有输入零线且各交流-直流变换电路的原边输入滤波电容的中点都连接至所述输入零线，或者，各交流源无输入零线且各交 流-直流变换电路的原边输入滤波电容的中点连接同一点。
一种所述的交流-直流变换电路的控制方法，所述电路具有正激及反激两种工作模式，所述电路中的第一、第二逆变开关管根据工作时段作为逆变开关管或箝位开关管使用，所述第一、第二逆变开关管中的一个开关管工作在逆变状态时另外一个开关管工作在箝位工作状态，所述电路中的原边输入滤波电容根据工作时段作为输入滤波电容或箝位电容使用，所述原边输入滤波电容在第一或第二逆变开关管关断时与所述电路中的隔离变压器的漏感产生谐振，使箝位开关管和逆变开关管获得零电压切换，通过谐振将所述隔离变压器漏感的能量传递到所述隔离变压器副边。
通过控制电路控制所述第一、第二逆变开关管的工作状态以使输入电流与输入电压的基波一致。
所述电路中的所述箝位谐振电路在交流整流的全周期采用以下两种控制方式：在低于设定电压阈值的低压阶段不发驱动信号，在不低于设定电压阈值的高压阶段才发驱动信号以进行箝位电流的反向谐振。
本发明的有益技术效果：
本发明提供一种单相或者多相交流输入单级高功率因数宽范围交流-直流变换电路。本发明交流-直流变换电路中，输入滤波电容除了用作输入滤波功能外，还充当箝位功能，辅助实现原边逆变器开关管的软开关工作，逆变开关管除了逆变开关功能外，还充当箝位开关。即，本发明的电路能够充分利用交流输入正负半周的开关工作状态差异，复用负(正)端逆变开关管及负(正)端输入滤波电容，实现箝位开关管以及箝位电容的功能，类似三电平整流技术，减少整流回路中的二极管损耗。原边输入滤波电容可以在第一或第二逆变开关管关断时与隔离变压器的漏感产生谐振，使箝位开关管和逆变开关管获得零电压切换(ZVS)，通过谐振将高频隔离变压器漏感的能量传递到副边，避免漏感的能量损耗及瞬间造成逆变开关管的电压尖峰。
本发明可以减少整流回路中的二极管损耗，实现正(负)端逆变开关管的软开关，降低原边开关管及副边整流二极管的电压应力和开关损耗。同时利用副边的倍压整流，既降低了副边整流二极管的电压应力，也同时将正激能量和反激能量形成不同的回路，从而巧妙的实现了类似常规的功率因素矫正类似的能量传送。即输出电压是成比例的输入电压(正激能量)加上成比例的电感储存能量(反激能量)的电压。
此外，通过多态的箝位控制模式，让箝位电路在交流整流的全周期分 为两种控制工作方式，即在输入低压部分不发驱动信号，在相对高压部分才发驱动信号进行箝位电流的反向谐振，有效地控制箝位电路的损耗并提高电路的可靠和稳定性。
本发明尤其适合在三相或多相交流输入且输出电压较高、半导体器件无法承受高压的场合，以及对功率因素，功率密度及体积限制较严格的场合。
附图说明
图1是本发明交流-直流变换电路实施例一的电路图；
图2是图1所示电路的变压器等效结构一示意图；
图3是图1所示电路开关管驱动时序图；
图4是本发明交流-直流变换电路实施例二的电路图；
图5是本发明交流-直流变换电路实施例三的电路图。
具体实施方式
以下结合附图对本发明的实施例作详细说明。应该强调的是，下述说明仅仅是示例性的，而不是为了限制本发明的范围及其应用。
实施例一
如图1所示的交流-直流变换电路，包括输入整流电路、原边输入滤波电容C1、C2、原边逆变电路、驱动电路、高频隔离变压器T1、副边整流电路、副边第三、第四电容C3、C4、副边输出滤波电容C5以及连接在所述副边输出滤波电容C5与所述驱动电路之间的控制电路。
输入整流电路包括两个输入整流二极管D1、D2，输入整流二极管D1的阳极和输入整流二极管D2的阴极连接交流输入火线L，所述原边输入滤波电容包括以交流输入零线N为中点的串联的两个输入高频滤波电容C1、C2，两个输入高频滤波电容C1、C2的相反端分别与所述输入整流电路的正端及负端连接。
所述原边逆变电路包括第一、第二逆变开关管Q1、Q2，所述第一、第二逆变开关管Q1、Q2分别与所述输入整流电路的正端和负端连接，所述隔离变压器T1原边绕组的一端与所述第一、第二逆变开关管Q1、Q2串联，另一端与交流输入零线N连接，所述原边逆变电路与原边输入滤波电容C1、C2也构成回路，形成箝位谐振电路，所述逆变开关管Q1、Q2工作在ZVS状态。通过控制逆变开关管Q1、Q2的门极电压，可将直流电压转换成脉冲电压加在隔离变压器的原边绕组。通过控制逆变开关管Q1、Q2中起箝位作用的开关管，可以控制谐振电流反向回路的通断。
所述副边整流电路为正激及反激工作整流回路，包括第三二极管D3和第四二极管D4，所述第三二极管D3的阴极与所述第四二极管D4的阳极及所述隔离变压器副边绕组的其中一输出端相连，所述第四二极管D4的阴极与电源的输出正端V+以及所述副边输出滤波电容C5的一端连接，所述第三二极管D3的阳极与电源的输出负端V-以及所述副边输出滤波电容C5的另一端相连，所述隔离变压器副边绕组的另一个输出端与副边第三、第四电容C3、C4的中间点连接，所述副边第三电容C3的正极(当采用有极性电容时)接电源的输出正端V+，所述副边第四电容C4的负极(当采用有极性电容时)接电源的输出负端。副边第三、第四电容C3、C4也可以采用有极性电容。
根据电路连接的原理，正激工作回路的电容与反激工作回路的电容上电压有与输入交流整流后的波形趋势相同或者互补的形态，且正激回路电容的电压及与初级侧的输入电压变化有近似线性变化关系。
通过所述控制电路控制所述第一、第二逆变开关管的工作状态，以使输入电流与输入电压的基波一致，从而实现高输入功率因数校正。
由于输入电路是对交流电压进行整流，故输入高频滤波电容C1、C2的容量不大，其参数主要是由用来箝位的谐振频率来决定。所以本电路保证了输入电流可以有条件跟随输入电压，以保证电源的输入功率因数和总谐波含量(Total Harmonics Distortion，简称THD)。
当交流输入时，正(负)半周(以下括号内容均对应交流负半周)将通过输入整流二极管D1(D2)进行整流，然后原边输入滤波电容C1(C2)会进行高频滤波。逆变线路则由逆变开关管Q1(Q2)和隔离变压器T1共同构成。此时原边箝位谐振电路由箝位电容C2(C1)、箝位开关管Q2(Q1)
及变压器原边线圈共同构成。在工作周期内，根据控制电路计算的结果，通过驱动电路，给逆变开关管Q1施加一个高频的PWM信号，同时给箝位开关管Q2上的驱动电压是一个与Q1近似互补的PWM电压，两个PWM驱动电压之间有一定的死区延迟关系。
所述隔离变压器是磁芯是开有气隙的隔离变压器或原边串联有谐振电感的隔离变压器或副边串联有储能电感的隔离变压器，磁芯气隙的大小由正、反激的比例和系统输入输出参数共同决定，原、副边耦合系数无需另外做特定的设置。
隔离变压器T1的磁芯开有气隙，有一定漏感，使隔离变压器T1工作能够在正激及反激两个状态。其漏感通过自然的绕制工艺得到，同时，根据实际的需要，可以通过绕制工艺的改变来获得可大可小的漏感。当然，如果自然绕制的漏感感量不足够，也可以在次级侧外加电感。
隔离变压器不用刻意区分原边及副边的端点连接点，即不用刻意考虑隔离变压器的起始端。
参阅图2，当隔离变压器T1绕制完成，其初级侧的主励磁电感Lm及漏感Lr确定。原边的漏感与谐振电容C2(C1)的谐振频率相对开关频率满足以下关系式：
Figure PCTCN2015070808-appb-000001
相关的工作回路及原理如下：
由于在交流输入时，正半周和负半周具有对称性，因此，以下以交流输入正半周为例。
状态一：
正半周将输入通过二极管D1进行整流，然后电容C1会进行高频滤波。在工作周期内，根据输入电压的反馈，控制电路计算出结果，通过驱动电路，给逆变开关管Q1施加一个高频的PWM信号。当逆变器开关管Q1开通的时候，变压器原边励磁电感Lm及谐振电感Lr1开始线性充电，当原边的电流等于励磁电流时，副边耦合的电压V2上升到VLr2+V C4时，二极管D4导通，即电压V2被箝位；副边电流为I2，原边的电流近似ILr＝ILm+I2/n。此状态对于输出整流就与正常的正激一样，同时因为输入整流电压是正弦型，输入高频滤波电容容值较小，所以电容C4上的电压波形也成为近似正弦型，与输入电压Vin有近似1/n的线性关系。
状态二：
当逆变器开关管Q1关断的时候，逆变器开关管Q1寄生电容被充电，其充电过程也是谐振，只是因为寄生电容较小，充电时间很短，可以视为线性的。同时次级漏感或者外接电感Lr2的电势VLr2发生偏转，试图维持原来的电流方向及大小不变，但随着时间的推移，其通过电感或者整流二极管D4的电流必然开始下降。
状态三：
当逆变器开关管Q1其寄生电容电压被充电至足够高，约为电压VC2+Vin，箝位开关管Q2反并二极管被正偏导通。箝位电容C2将谐振电感Lr1和励磁电感Lm的电压箝制在电压VC2，因为箝位电容C2比逆变开关管Q1的寄生电容大的多，绝大部分谐振电流进入箝位电容C2，箝位电容C2与谐振电感Lr1开始谐振；当原边的电流与励磁电流相等时，变压器副边的输出电流等于零，同时副边耦合电压V2电势发生交变。
状态四：
当原边电压下降到足够低，副边对应的耦合电压V2足够使二极管D3正偏导通。此时副边反射到原边的电压约为n(Vo-Vc3-VLr2)，为箝位开关管Q2能够获得ZVS提供了条件，此时箝位开关管Q2的驱动电压将变为高电平而导通。在这个工作状态模式下，原来存储在变压器气隙中的能量杯释放出来。该状态是一个典型的反激式变压器工作。由于电容C4上的电压与输入电压是一个线性关系，所以电容C3上的电压与电容C4上的电压互补地合成输出电压。
状态五：
当箝位开关管Q2关断时，迫使箝位电容C2脱离原谐振回路，同时谐振电感Lr1将与逆变开关管Q1的寄生电容形成新的谐振；以释放寄生电容的电荷，为逆变开关管Q1的ZVS做准备。
状态六：
当Q2驱动关断一定时间后，通过状态五中的谐振将逆变开关管Q1的寄生电容的电荷完全释放，同时通过逆变开关管Q1的反并二极管进行续流，此时逆变开关管Q1获得ZVS开通条件。
状态七
此时，将逆变开关管Q1的驱动电压变为高电平而导通；原边的电感将被线性充电，开始新的周期，重复以上的状态过程。
如图3所示，是本发明的电路开关管驱动时序图，在本电路中，当输入电压在第①、第③区域时，由于逆变开关管的占空比较大，在逆变开关管关断期间，时间较短，前述提到的谐振电流还没有来得及反向。同时由于输入电压较低，对于电路中的半导体元件电压应力较小，为了避免造成谐振状态的不确定性和电路的不可靠性，较佳地，明确通过在设定第①、 第③区域输入电压限值，长时间关掉箝位管的驱动，使箝位电容值充当普通的尖峰吸收功能，同时也减少了驱动损耗；当输入电压在第②、第④区域时，箝位开关管Q2(或者Q1)才按照上述的箝位方法工作，有效的提高电路的稳定可靠性。
由以上的工作模式中的状态一级状态四的分析可知，在交流-直流变换电路中，隔离变压器好比是线性的变压器，输入电压的线性比例的降低输入到电容C4，同时隔离变压器又好比常规功率因素矫正电路中PFC电感，将在开关管导通中存储的能量在开关管关断的时候释放至电容C3，因此电容C3、C4上的电压构成了一个线性比例与常规无隔离的PFC电压。所以，电路很好地实现了隔离式PFC，而这种优势已知的单级隔离式交流-直流变换电路所不具备的。
当输入电压为交流的负半周时，将通过二极管D2整流，然后电容C2会进行输入高频滤波。逆变线路则由逆变开关管Q2和隔离变压器T1原边线圈共同构成；此时原边箝位谐振电路由箝位电容C1、箝位开关管Q1及变压器原边线圈共同构成。同时电容C3、C4的电压波形也对称性调转，电容C3充当正激回路的输出电容，C4充当反激回路的输出电容。其他工作状态原理及控制方法与前述正半波的一致。
由以上分析可知，本电路中，原边的逆变开关管、高频滤波电容在输入电压正负半周的时候被巧妙地分时复用为箝位开关管及箝位电容，输入整流回路只需通过一个二极管即可，同时输出也只需要通过一个二极管就构成整流回路。因此，本电路的线路简洁，效率高。
实施例二
本发明还提供一种二相、三相或更多相输入的交流-直流变换电路。如图4所示为一种三相四线输入交流-直流变换电路，基本电路组成及有益效果与实施例一相同，区别是：输入是三相。其好处是理论上可以获得比实施例一输出电压特性更好的输出电压，输出电压更加平滑，纹波电压更小。图4中，D1a、D2a、D1b、D2b、D1c、D2c表示原边整流二极管，C1a、C2a、C1b、C2b、C1c、C2c表示原边滤波电容，Q1a、Q2a、Q1b、Q2b、Q1c、Q2c表示原边逆变开关管，T1a、T1b、T1c表示变压器，D3a、D4a、D3b、D4b、D3c、D4c表示副边整流二极管，C3a、C4a、C5b、C3b、C4b、C5b、C3c、C4c、C5c表示副边滤波电容。
实施例三
本发明还提供另一种二相、三相或更多相输入的交流-直流变换电路。如图5所示为一种三相三线输入交流-直流变换电路，基本电路组成及有益效果与实施例二相同，区别是：输入无零线输入。其好处是在实际三相三线无零线输入使用环境中，依然可以实现前述性能。
本发明的电路工作时有正激及反激两个不同的工作模式，可以实现较大的输入、输出电压的调节范围，同时输入电流还可以跟踪输入电压，实现功率因素的矫正。同时由于有源箝位电路功能，可以降低原边开关管及副边整流二极管因为隔离变压器漏感等因素引起的反向恢复电压尖峰及开关损耗，提高效率；相比传统的电路更加简洁；此外，可以通过多态的箝位控制模式方法，有效的控制箝位电路的损耗；本发明尤其适合广泛运用在三相(多相)交流输入，及输出电压较高的场合。
以上内容是结合具体的优选实施方式对本发明所作的进一步详细说明，不能认定本发明的具体实施只局限于这些说明。对于本发明所属技术领域的普通技术人员来说，在不脱离本发明构思的前提下，还可以做出若干简单推演或替换，都应当视为属于本发明的保护范围。

Claims (10)

  1. 一种交流-直流变换电路，其特征在于，包括输入整流电路、原边输入滤波电容、原边逆变电路、驱动电路、隔离变压器、副边整流电路、副边第三至第四电容、副边输出滤波电容以及连接在所述副边输出滤波电容与所述驱动电路之间的控制电路，所述驱动电路连接所述原边逆变电路；
    所述输入整流电路包括第一至第二输入整流二极管，所述第一输入整流二极管的阳极和所述第二输入整流二极管的阴极连接交流输入火线，所述原边输入滤波电容包括以交流输入零线为中点串联的两个输入高频滤波电容，两个输入高频滤波电容的相反端分别与所述输入整流电路的正端及负端连接；
    所述原边逆变电路包含第一至第二逆变开关管，所述第一逆变开关管和所述第二逆变开关管分别与所述输入整流电路的正端和负端连接，所述隔离变压器原边绕组的一端与第一、第二逆变开关管串联，所述隔离变压器原边绕组的另一端与输入零线连接；所述原边逆变电路与原边输入滤波电容也构成回路，形成筘位谐振电路，所述逆变开关管工作在零电压切换状态；
    所述副边整流电路包括第三至第四二极管，所述第三二极管的阴极与所述第四二极管的阳极及所述隔离变压器副边绕组的其中一输出端相连，所述第四二极管的阴极与电源的输出正端以及所述副边输出滤波电容的一端连接，所述第三二极管的阳极与电源的输出负端以及所述副边输出滤波电容的另一端相连，所述隔离变压器副边绕组的另一个输出端与副边第三至第四电容的中间点连接，所述副边第三电容连接电源的输出正端，所述副边第四电容连接电源的输出负端；所述副边整流电路具有正激及反激两种工作模式。
  2. 如权利要求1所述的交流-直流变换电路，其特征在于：所述隔离变压器是磁芯是开有气隙的隔离变压器或原边串联有谐振电感的隔离变压器或副边串联有储能电感的隔离变压器。
  3. 如权利要求1所述的交流-直流变换电路，其特征在于：所述隔离变压器是正激工作线圈与反激工作线圈复用的单个副边绕组的变压器。
  4. 如权利要求1所述的交流-直流变换电路，其特征在于：所述副边 第三、第四电容是两个串联的无极电容或者有极性电容，且在为有极性电容的情况下，所述副边第三电容的正极接电源的输出正端，所述副边第四电容的负极接电源的输出负端。
  5. 一种交流-直流变换电路，其特征在于：包括2个、3个或者更多个如权利要求1-4任一项所述的交流-直流变换电路，各交流-直流变换电路的输入连接到具有2个、3个或者更多个相位输入的交流源的不同相位输入。
  6. 如权利要求5所述的交流-直流变换电路，其特征在于：各交流源有输入零线且各交流-直流变换电路的原边输入滤波电容的中点都连接至所述输入零线，或者，各交流源无输入零线且各交流-直流变换电路的原边输入滤波电容的中点连接同一点。
  7. 一种如权利要求1至4任一项所述的交流-直流变换电路的控制方法，其特征在于：
    所述电路具有正激及反激两种工作模式，所述电路中的第一、第二逆变开关管根据工作时段作为逆变开关管或筘位开关管使用，所述第一、第二逆变开关管中的一个开关管工作在逆变状态时另外一个开关管工作在筘位工作状态，所述电路中的原边输入滤波电容根据工作时段作为输入滤波电容或筘位电容使用，所述原边输入滤波电容在第一或第二逆变开关管关断时与所述电路中的隔离变压器的漏感产生谐振，使筘位开关管和逆变开关管获得零电压切换，通过谐振将所述隔离变压器漏感的能量传递到所述隔离变压器副边。
  8. 如权利要求7所述的控制方法，其特征在于：通过控制电路控制所述第一、第二逆变开关管的工作状态以使输入电流与输入电压的基波一致。
  9. 如权利要求7或8所述的控制方法，其特征在于：所述电路中的所述筘位谐振电路在交流整流的全周期采用以下两种控制方式：在低于设定电压阈值的低压阶段不发驱动信号，在不低于设定电压阈值的高压阶段才发驱动信号以进行筘位电流的反向谐振。
  10. 一种如权利要求5至6任一项所述的交流-直流变换电路的控制方法，其特征在于：
    所述电路具有正激及反激两种工作模式，所述电路中的第一、第二逆变开关管根据工作时段作为逆变开关管或筘位开关管使用，所述第一、第二逆变开关管中的一个开关管工作在逆变状态时另外一个开关管工作在 筘位工作状态，所述电路中的原边输入滤波电容根据工作时段作为输入滤波电容或筘位电容使用，所述原边输入滤波电容在第一或第二逆变开关管关断时与所述电路中的隔离变压器的漏感产生谐振，使筘位开关管和逆变开关管获得零电压切换，通过谐振将所述隔离变压器漏感的能量传递到所述隔离变压器副边。
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CN109245568A (zh) * 2018-09-12 2019-01-18 杭州海兴电力科技股份有限公司 一种交流转直流隔离开关电源电路
CN109245568B (zh) * 2018-09-12 2024-05-24 杭州海兴电力科技股份有限公司 一种交流转直流隔离开关电源电路

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