Classifications

• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
  • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
  • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
  • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
  • H03B5/1228—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device the amplifier comprising one or more field effect transistors
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
  • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
  • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
  • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
  • H03B5/1206—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device using multiple transistors for amplification
  • H03B5/1212—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device using multiple transistors for amplification the amplifier comprising a pair of transistors, wherein an output terminal of each being connected to an input terminal of the other, e.g. a cross coupled pair
  • H03B5/1215—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device using multiple transistors for amplification the amplifier comprising a pair of transistors, wherein an output terminal of each being connected to an input terminal of the other, e.g. a cross coupled pair the current source or degeneration circuit being in common to both transistors of the pair, e.g. a cross-coupled long-tailed pair
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
  • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
  • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
  • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
  • H03B5/1237—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator
  • H03B5/124—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator the means comprising a voltage dependent capacitance
  • H03B5/1246—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator the means comprising a voltage dependent capacitance the means comprising transistors used to provide a variable capacitance
  • H03B5/1253—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator the means comprising a voltage dependent capacitance the means comprising transistors used to provide a variable capacitance the transistors being field-effect transistors
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
  • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
  • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
  • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
  • H03B5/1237—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator
  • H03B5/1293—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator having means for achieving a desired tuning characteristic, e.g. linearising the frequency characteristic across the tuning voltage range
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03J—TUNING RESONANT CIRCUITS; SELECTING RESONANT CIRCUITS
  • H03J2200/00—Indexing scheme relating to tuning resonant circuits and selecting resonant circuits
  • H03J2200/36—Circuit arrangements for, e.g. increasing the tuning range, linearizing the voltage-capacitance relationship, lowering noise, constant slope in different bands

Definitions

• the present invention relates to a tunable, capacitive component and an LC oscillator with the tunable, capacitive component.
• Tunable, capacitive components are usually implemented using varactor diodes.
• the junction capacitance depends on an applied control voltage.
• Varactor diodes or tunable, capacitive components are used, for example, on a large scale in voltage-controlled oscillators, so-called voltage controlled oscillators (VCO).
• VCO voltage controlled oscillators
• these usually comprise a tunable capacitance and a fixed value inductance and are therefore referred to as LC oscillators.
• the oscillation frequency of the oscillator is set by varying the capacitance value of the tunable capacitance.
• Such voltage-controlled oscillators are required, for example, in transmitters and receivers in mobile radio.
• FIG. 1 shows a basic circuit arrangement for a voltage-controlled LC oscillator with two inductors 1, two tunable capacitors 2 and two cross-coupled MOS transistors 3 according to the prior art with a symmetrical structure.
• the oscillation frequency F osz of the LC VCO according to FIG. 1 is determined to a good approximation by the resonance frequency of the LC
• the frequency V osz of the LC-VCO can therefore be controlled by the voltage-controlled capacitances 2 which can be set via the voltage Vtune.
• the LC oscillator comprising the actual LC resonant circuit 1, 2 and the damping amplifier 3, which is coupled to it, is supplied by a reference current source 4 fed.
• the controlling voltage Vfune is present at the controlled capacitors 2 at their connecting nodes, which form the circuit input 5.
• a pair of circuit nodes 6, 7 forms the output of the circuit, so that the control voltage Vtune applied from the outside against reference potential connection 8 does not directly impress the control voltage at the (capacitance) control inputs of the varactors 2.
• the voltages of the nodes 6, 7 oscillate 180 degrees out of phase with the frequency Fosz and the amplitude Uosz around a mean voltage value which, measured against ground, depends on the current Iref and the design of the transistors 3.
• the current Iref in the In practice, reference current source 4 shown in FIG. 1 is never completely independent of the supply voltage, so that disturbances in the supply voltage project into the reference current Iref. Furthermore, a reference current source itself is never completely free of noise and interference.
• Disturbances in the reference current Iref cause potential fluctuations in the same direction at the outputs A and B of the LC-VCO and thus project into the mean voltage value. Since the average capacitance value of the varactors 2 is determined directly by the tuning voltage that is present across the varactors 2, disturbances in the reference current Iref also simultaneously change the frequency F osz or the phase position of the LC-VCO. Random random disturbances in the frequency or phase of an oscillator oscillation are usually observed in the form of phase noise.
• the change in capacitance 2 with the controlling voltage is not unnecessarily high. It is therefore desirable if the voltage range over which the capacitance 2 can be controlled is as large as possible, and at the same time the voltage dependency of the capacitance 2 is linear over the entire control voltage range. So that amplitude noise does not transform into phase noise, it is further advantageous if the controlled capacitance value is not a function of the voltage amplitude present at capacitance 2.
• FIG. 2a shows a voltage-controlled capacitance 2 according to the prior art with two normally-off NMOS transistors 9, the four source / drain connections of which are connected to the tuning input 5.
• the output node pair 6, 7 is connected to a gate connection of the transistors 9.
• the circuit arrangement shown in FIG. 2a can be used unchanged for the LC-VCO in accordance with FIG.
• Potential fluctuations in the same direction at the connections 6, 7 with respect to the connection 5 each cause - within the control range of the capacitance between the gate and transistor channel - a change in the same direction of the capacitance of the NMOS transistors 9 between the gates and the interconnected drain and source connections.
• the change in the capacitance or the capacitance coverings with the control voltage which drops off effectively via the transistors 9 working as a varactor, is comparatively very large and only linear in a very small range, which is less than 50 mV. If the amplitude U osz at the outputs 6, 7 of the LC-VCO is very large, the situation improves somewhat.
• Figure 2b shows the electrical equivalent circuit of the tunable capacitance of Figure 2a.
• a controllable capacitance is specified in a VCO, which is constructed by means of MOS transistors.
• the known capacitive components which can be tuned and are suitable for use in VCO have the disadvantage in common that they do not have a large linear tuning range and / or have relatively high series resistances. High series resistances lead to poor quality, while an insufficient tuning range of a varactor or a tunable capacitance generally has the disadvantage of only a narrow frequency band in which the VCO can be tuned when used in a VCO. Finally, low linearity means that circuit properties are dependent on the current operating point of the capacitive component, which in turn has a disadvantageous effect on the implementation of control loops etc.
• the object of the present invention is therefore to provide a tunable, capacitive component which, with improved properties, is suitable for use in voltage-controlled oscillators. It is a further object of the invention to provide an oscillator with the improved component. According to the invention, the object relating to the tunable, capacitive component is achieved by a tunable, capacitive component, comprising:
• a tuning input for supplying a tuning voltage, a pair of circuit nodes between which a capacitance that can be tuned by means of the tuning voltage is provided,
• a reference signal input designed to supply a reference signal for setting the operating point of the MOS transistors, the reference signal input being connected to the resistors at its connecting node or to the four source / drain connections.
• the MOS transistors which are interconnected as a pair of transistors, form controllable capacitors.
• these controllable capacities are not directly connected to the pair of circuit nodes, but the dynamic connection to the circuit nodes is established by means of the two coupling capacitors.
• low-frequency and DC voltage fluctuations in the mean voltage value of the voltage between the switching nodes are no longer included in the tuning voltage.
• the coupling of the pair of MOS transistors to the tuning input can be carried out indirectly or directly.
• the two coupling capacitances preferably have capacitance values which are large compared to the maximum settable capacitance values of the MOS transistors.
• the coupling resistors preferably have resistance values which are large compared to the reciprocal of the product of 2 ⁇ , the oscillator frequency and the capacitance value of the MOS transistors.
• the present principle is based on the knowledge that by executing the capacitive component as a pair of MOS transistors in combination with the provision of a reference signal input which is designed to set the operating point of the MOS transistors by means of a reference signal, a tunable, capacitive component is realized , which has an adjustable tuning range.
• the tuning input is preferably connected to the source / drain connections of the transistor pair, alternatively to the gate connections of the transistors via coupling resistors.
• the reference signal input is coupled to the opposite input.
• MOS transistors have a bulk connection, this is preferably connected to a reference potential connection, that is to say to a ground connection.
• a further pair of MOS transistors or any number of further pairs of MOS transistors can be provided.
• the one or more pairs of MOS transistors are connected in parallel with the first-mentioned pair of MOS transistors in such a way that the gate connections are also connected to the pair of circuit nodes which form the output of the capacitive component via further pairs of coupling capacitances.
• Vote entrance and Reference signal inputs are designed in accordance with the circuitry in the first-mentioned pair of MOS transistors.
• the tuning input is connected to the four source / drain connections of the pair of MOS transistors, then all further pairs of MOS transistors with their four source / drain connections are also connected directly to one another and to the tuning input.
• the further reference signal inputs one of which is assigned to a pair of transistors, are connected in pairs to the gate connections of the pairs of transistors via further coupling resistors.
• the reference signal input and the further reference signal input (s) are connected to the four source / drain connections of the assigned transistor pair.
• the further MOS transistor pairs are each coupled in pairs to the common tuning input with a further pair of resistors.
• the tuning input can also be called a tuning input.
• the reference signal inputs can be coupled in and out with the transistor pairs for switching on and off respective reference signals.
• the reference signals that can be supplied to the individual transistor pairs and are assigned in each case can be different.
• the provision of the reference signals provided for setting the operating point can be made in a simple manner, for example with by means of a resistance chain, which is connected to a reference signal source and has tapping points for tapping respective reference signals, the tapping points being coupled directly or switchably to assigned reference signal inputs.
• Switches are preferably provided in each case, which couple the reference signal inputs to the assigned MOS transistor pairs for switching the respective reference signals on and off. This makes it possible to precharge stabilization capacitances connected to ground on the transistor side, for example, which in turn ensure that when switches are open, even in the event of faults, for example on the supply voltage, the linear relationship between the effective capacitance between the circuit nodes and the tuning voltage, that is to say with The present principle achieved high linearity of the tuning characteristic, is retained.
• the stabilizing capacities with their capacitance values are advantageously large compared to the transistor capacities.
• an LC oscillator with a tunable, capacitive component as described above comprising
• an attenuation amplifier which provides a negative impedance and is coupled to the resonator core, -
• the resonant circuit frequency can be detuned with the tuning voltage that can be supplied to the tuning input.
• the advantages of the tunable, capacitive component according to the invention namely a large variation ratio, highly linear and preferably temperature-stable tuning characteristic and low series resistance occur particularly advantageously when the component is used in an LC oscillator.
• a large variation ratio of the tunable capacitance ie a large quotient of the largest and smallest adjustable capacitance, enables the oscillator frequency of the LC oscillating circuit to be tuned over a wide frequency range. This feature is particularly advantageous when used in local oscillators of high-frequency transmitters and receivers, since a large number of widely spaced transmission channels can thus be addressed.
• the highly linear tuning characteristic of the tunable, capacitive component enables circuit properties to be dimensioned independently of the operating point, in particular when implementing control loops.
• the low series resistance of the capacitance leads to a high quality of the oscillator circuit.
• the LC oscillator with the tunable, capacitive component according to the present principle shows particularly low phase noise.
• FIG. 1 shows a basic circuit diagram of an LC-VCO with NMOS
• FIG. 2b shows the electrical equivalent circuit diagram of FIG. 2a
• FIG. 3a shows a first exemplary embodiment of a controllable capacitance with NMOS transistors according to the present principle
• FIG. 3b shows the electrical equivalent circuit of FIG. 3a
• Figure 4a shows a second embodiment of a controllable capacitance with NMOS transistors according to the present principle
• FIG. 4b shows the electrical equivalent circuit diagram of FIG. 4a
• FIG. 5 shows an LC-VCO with a tunable capacity according to FIG. 3a.
• FIG. 6 shows the tuning line of a controllable capacitance with NMOS transistors
• FIG. 8 shows the tuning characteristic of the controllable capacity from FIG. 7,
• FIG. 7 with pre-charging that can be switched off
• FIG. 10a shows an exemplary embodiment of a switch from FIG. 9,
• FIG. 10b the equivalent circuit of the switch from FIG. 10a
• FIG. 11 shows a further development of the tunable capacitance from FIG. 9 with additional temperature compensation of the working points
• FIG. 12 shows the subject of Figure 11, but with the
• FIG. 13 shows the tuning characteristic of the controllable capacity from FIG. 11,
• FIG. 14 is a diagram to illustrate the superposition of the tuning characteristic curves when several capacitors are connected in parallel with different operating points according to the present principle
• FIG. 15 shows the control characteristic of a VCO according to FIG. 5
• FIG. 3a shows an exemplary embodiment for a voltage-controlled capacitance in accordance with the present principle in a further development of the subject of FIG. 2a with two normally-off NMOS transistors 9 which, with their gate connections, are coupled dynamically to the nodes via coupling capacitances 10
• the four load connections of the transistors 9 are connected to one another and to the tuning input 5 with their source / drain connections. Furthermore, a reference signal input 11 is provided for setting the operating point of the transistors 9, which is connected via a coupling resistor 12 to a respective gate connection thereof.
• An advantage of the coupling capacitances is that DC and low-frequency voltage fluctuations in the mean voltage value no longer enter into the control voltage of the transistors 9.
• the proposed possibility for setting the operating point of the transistors 9 advantageously enables any good one Linearization of the tuning characteristic, as explained in more detail later.
• Ct represents the controllable capacitance value of the transistors 9 operating as a varactor.
• Figure 3b shows the electrical equivalent circuit of the object of Figure 3a.
• FIG. 4a shows an alternative embodiment of the subject of Figure 3a.
• this largely corresponds to that of FIG. 3a, but the connections for tuning voltage and reference signal 5, 11 are interchanged. Accordingly, the tuning input 5 is connected here via resistors 12 to the gates of the transistors 9, while the reference signal input 11 is connected to their source / drain connections.
• Figure 4b shows the electrical equivalent circuit of the object of Figure 4a.
• FIG. 5 shows a circuit arrangement for a voltage-controlled LC oscillator which is improved compared to FIG.
• NMOS transistors 9 according to the present principle, as shown in ren 3a and 4a shown.
• the controllable capacitances 2 are no longer directly connected to the nodes 6, 7, but via coupling capacitances 10, with the advantages and dimensioning rules already explained.
• FIG. 6 shows an example of the simulated small signal relationship between the control voltage V ⁇ - U ne unc ⁇ the effective capacitance C between the nodes 6, 7 for a voltage-controlled capacitance 2 constructed according to FIG. 3 a with self-blocking NMOS transistors 9, which are connected via the Coupling capacitances 10 are dynamically coupled to the nodes 6, 7.
• V re f 0 volts
• the control voltage range from V ⁇ ne shifts completely into the negative range.
• the value for V re f for the circuit arrangement according to FIG. 3a must generally always be chosen to be greater than or equal to V ⁇ O.
• the GND potential would be an ideal, interference-free reference potential for Vref.
• connections 5, 11 for V tune and V re f can also be used as a differential voltage control input for the between the nodes 6, 7 effective capacity can be understood, so that the objects according to FIG. 3a and FIG. 4a are actually identical from a technical point of view.
• FIG. 7 shows a circuit arrangement which, with regard to the connections, tuning input 5 and output pair node pair 6, 7, consists of N circuit arrangements according to FIG. However, the reference signal connections for supplying the reference signals V re f ] _ to V re f are not connected to one another.
• the reference signal connections 11, 13, 14 are at potentials which systematically increase or decrease systematically compared to the GND potential, technically advantageously in the same order of magnitude.
• the circuit arrangement of FIG. 7 approximately corresponds to the positioning range of a circuit arrangement with regard to the adjustment range of the capacitance between the nodes 6, 7 3a corresponds, the following conversion condition can be used, for example:
• the coupling capacitances are reduced by a factor of N compared to the object in FIG. 3a.
• the coupling resistances are made larger by a factor of N.
• Channel lengths of the transistors 9 are retained.
• Channel widths of the transistors 9 are reduced by a factor of N.
• circuit arrangement according to FIG. 7 in the manner described can also alternatively be formed with technically advantageously similar circuit arrangements according to FIG. 4a.
• Figure 8 shows an example of the simulated relationship between the control voltage Vt une and the effective capacitance between the nodes 6, 7 in accordance with a voltage controlled capacitance 7 is self-locking with NMOS transistors.
• the division factor N is 10.
• the simulation result shown in FIG. 8 shows that there is now an approximately linear relationship between the effective capacitance and the control voltage Vt une over a comparatively large range of approx.
• the comparison with the simulation result shown in FIG. 6 also shows that the influence of the temperature on the voltage dependence of the capacitance has decreased significantly over the entire control voltage range.
• the reference potentials Vrefl to VrefN can be derived from the supply voltage + V Q Q in such a way that any existing ones Do not project disturbances of the supply voltage + Vcc into the reference potentials V re f] _ to V re fu.
• FIG. 9 shows a development of the circuit arrangement according to FIG. 8.
• the potentials V re f_ to V re fj are generated by current supply with the aid of a resistor chain.
• a series connection of resistors 15 is provided with tapping points between the resistors 15, each of which is coupled to an assigned reference signal input of a tunable partial capacitance according to FIG. 3a.
• a switch 16 is provided for this coupling.
• each switch with a capacitance 17 is connected to the reference potential.
• the resistance chain which forms a voltage divider, is connected to a reference signal source 18 connected thereto, which is designed as a current source.
• the resistance chain is also connected to reference potential terminal 8.
• Both the gates of the paired NMOS transistors 9 and the storage capacitors 17 are charged to the respective potentials generated in this way via the electronic switches 16. After the end of the charging phase, the switches 16 are opened.
• the charges applied to the gates of the NMOS transistors 9 and to the storage capacitors 17 now ensure that the linear relationship between the effective capacitance between the nodes 6, 7 and the control voltage Vt une remains.
• the storage capacitors 17 each have a multiple (> 10) of the maximum as the capacitance value Have MOS capacitance of the NMOS transistors 9, as seen from the control input 5, the storage capacitors 17 each form a capacitive voltage divider with the NMOS transistors 9. After the switches 16 are opened, the amount of charge applied to the gates of the NMOS transistors 9 and the storage capacitors 17 can practically no longer change, for example due to disturbances at + Vcc.
• the above statement applies in particular exactly when the switches 16 are ideal switches.
• the electronic switches used in the monolithic integration preferably consist of real transistors with finite properties.
• Electronic MOS switches e.g. Leakage currents, the 'off' resistance is not infinitely large and the 'on' resistance is not zero.
• circuit arrangement according to FIG. 9 in the manner described is alternatively also conceivable with technically advantageously similar circuit arrangements according to FIG. 4a.
• the coupling capacitors 17 also act as storage capacitors.
• FIG. 10a shows an electronic switch 16, as can be used for example in the subject of FIG. 9.
• the switch 16 comprises the complementary switching transistors 19, 20, the controlled paths of which are connected in parallel and form the load path of the switch 16.
• an inverter 21 has its output connected to this control input.
• a further complementary pair of transistors 22, 23 with its controlled paths is also connected in parallel with the controlled paths of the transistors 19, 20. These transistors 22, 23 are connected as diodes.
• From the state level at the control input 24 of the inverter 21 and thus the switch 16, an inverted control signal for the transistor 19 is generated with the aid of the inverter 21.
• the control input 24 has an H level, both the transistor 19 and the transistor 20 are conductive.
• connection 25 forms the input of the switch and the connection 26 forms the output.
• the storage capacity 17, which is connected to the output 26, forms a time constant with the forward resistance between the nodes 25, 26.
• FIG. 10a for an electronic switch 16 is expanded by the NMOS transistors 22, 23, which bring about a so-called non-linear potential connection.
• the switch connections 25, 26 have the same potential, the transistors 22, 23 are blocked, i.e. maximum high impedance. If the switch connections 25, 26 have different potential, depending on the sign of the potential difference, either diode 22 or diode 23 becomes slightly conductive according to the magnitude of the potential difference.
• the diodes 22, 23 prevent the charges on the gates of the NMOS transistors 9 and the capacitances 17 in FIG. 9 from changing too greatly, owing to leakage currents.
• the leakage current losses are dependent on the leakage current
• FIG. 10b shows the electrical equivalent circuit diagram of the switch 16 according to FIG. 10a.
• FIG. 11 shows an expansion of the circuit arrangement according to FIG. 9. In structure and advantageous mode of operation, this largely corresponds to the subject of FIG. 9. In contrast, however, auxiliary transistors 27 with their controlled paths are connected between the tapping points of the resistor chain 15 and the switches 16.
• the source terminals of the transistors 27 are respectively connected to the tapping points, the re reference signals V re f] _ to provide V f.
• the drain connections are connected to the inputs 25 of the respective electronic switches 16.
• a reference current Ic which is fed in at the drain connections of the transistors 27 is in each case discharged again with the aid of current mirror arrangements 28 which are connected to the source connections of the transistors 27 in such a way that reference voltage tapping points V re fi to V re f (N-1) no current is fed in each case.
• the reference voltages V re fi 'to V re are applied to the inputs 25 of the respective electronic switches 16 shown in FIG f j ', which are each shifted in the manner described below with respect to the voltages V re f ] _ to V re f ⁇ by the voltage components Vtl, ... VtN.
• V ref N ' V re f N + VtN
• the voltage components Vtl,... VtN are generated with the aid of the auxiliary transistors 27, each of which a reference current Ic flows through.
• the magnitude of the reference current Ic is selected so that the voltage drop from drain to source via the auxiliary transistors 27 corresponds in good approximation to the threshold voltages Vtl, ... VtN.
• the generated voltage components Vtl, ... VtN ideally correspond to the respective operating voltages of the divided MOS capacities 9. If the voltage components Vtl, ... VtN are successfully generated and superimposed on V re f ] _ bis exactly, then the influence of the threshold voltages of the divided MOS capacitances 9 on the characteristic behavior between the effective capacitance between the nodes 6, 7 and the control voltage V £ une is completely eliminated. The reason for this is that the control area of a MOS capacitance is always centered around the MOS insertion voltage Vt.
• FIG. 12 shows a variant of the circuit arrangement according to FIG. 11 using the circuit arrangement analogously according to FIG. 4a instead of those according to FIG. 3a.
• Vrefi 1 Vrefi - Vti
• control relationship between the control voltage V and the effective capacitance between the nodes 6, 7 is accordingly inverted with respect to the relationship shown in FIG. 13 below.
• FIG. 13 shows an example of the simulated relationship between the control voltage Vt une and the effective capacitance between the nodes 6, 7 for a voltage-controlled capacitance according to FIG. 11 with normally-off NMOS transistors.
• the division factor N is 10.
• FIG. 15 shows a control characteristic curve measured at 27 degrees of an LC VCO according to FIG. 5, which has a controllable capacity according to the arrangement shown in FIG.
• the division factor N is 10.
• VCO control characteristic is approximately 1.5 volts according to the linear range of controllable capacity.
• circuit diagrams described above can also be implemented in complementary circuit technology.
• MOS capacitances can be implemented with both N-channel MOS transistors and P-channel MOS transistors.
• the level of the zero field threshold voltage V t ⁇ 0 of the MOS transistors 9 also plays no fundamental role.
• the function of a current mirror circuit can be carried out using simple or complex circuit technology, for example with cascodes.
• Alternative embodiments of the circuit structures specified here are therefore to be regarded as equivalent means for achieving the principle presented, which are within the scope of the invention.

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  • 2003-02-11 EP EP03743285A patent/EP1481470A1/de not_active Withdrawn
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