WO2023226723A1 - 双有源桥型微逆变器的电路建模方法及输出电流控制方法 - Google Patents

双有源桥型微逆变器的电路建模方法及输出电流控制方法 Download PDF

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WO2023226723A1
WO2023226723A1 PCT/CN2023/092409 CN2023092409W WO2023226723A1 WO 2023226723 A1 WO2023226723 A1 WO 2023226723A1 CN 2023092409 W CN2023092409 W CN 2023092409W WO 2023226723 A1 WO2023226723 A1 WO 2023226723A1
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microinverter
current
voltage
grid
signal model
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PCT/CN2023/092409
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English (en)
French (fr)
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李睿
杨佳涛
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上海交通大学
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33592—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H02J3/381—Dispersed generators
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/02—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc
    • H02M5/04—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters
    • H02M5/10—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using transformers
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J2203/00—Indexing scheme relating to details of circuit arrangements for AC mains or AC distribution networks
    • H02J2203/20—Simulating, e g planning, reliability check, modelling or computer assisted design [CAD]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J2300/00—Systems for supplying or distributing electric power characterised by decentralized, dispersed, or local generation
    • H02J2300/20—The dispersed energy generation being of renewable origin
    • H02J2300/22—The renewable source being solar energy
    • H02J2300/24—The renewable source being solar energy of photovoltaic origin
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E10/00—Energy generation through renewable energy sources
    • Y02E10/50—Photovoltaic [PV] energy
    • Y02E10/56—Power conversion systems, e.g. maximum power point trackers

Definitions

  • the present invention relates to the technical field of photovoltaic microinverters, and specifically to a circuit modeling method and an output current control method suitable for dual active bridge microinverters.
  • Microinverters generally refer to inverters in photovoltaic power generation systems with a power of less than or equal to 1000W and with component-level maximum power point tracking capabilities. Unlike centralized and string photovoltaic inverter systems, microinverters are directly connected to individual photovoltaic modules. The advantage is that each component can be independently controlled by MPPT, which greatly improves the overall efficiency while also avoiding problems such as DC high voltage, poor low-light effect, and barrel effect that exist in centralized inverters.
  • microinverters can be divided into three categories: DC bus structure, pseudo DC bus structure and no DC bus structure.
  • the microinverter without DC bus is a single-stage circuit and adopts matrix control. It uses a small number of switching devices and has high conversion efficiency, so it has more advantages.
  • the dual active bridge (DAB) type microinverter uses the smallest number of switching devices. On the basis of the wide-range soft switching characteristics of the DAB circuit, it also Improved the problem of low efficiency of DAB circuit under light load.
  • Small signal model derive the disturbance differential equations of grid current i g , grid current reference signal i g,ref , filter capacitor current reference signal and modulation signal u m in the control system; calculate grid-connected asymmetric cascade H-bridge multi-level Converter output voltage vi , derived without phase-locked loop
  • the system small signal model is derived; the small signal model of the phase-locked loop is derived, and the small signal model of the entire grid-connected asymmetric cascaded H-bridge multi-level converter system is established.
  • This method is also suitable for off-grid asymmetric cascaded H-bridge multi-level converter systems.
  • this method is applied to the circuit modeling of DAB microinverter, there are still the following problems:
  • the DAB microinverter belongs to a high-frequency chain AC-DC conversion system. Its working principle is essentially different from that of the cascaded H-bridge multi-level converter.
  • the primary and secondary voltages of the high-frequency transformer and the transformer current of the DAB microinverter are all AC components, so circuit modeling cannot be done directly through the modeling method in the above patent. Instead, these AC electrical quantities need to be equivalent;
  • the DAB type micro The transfer function gain of the inverter changes with changes in DC side voltage, AC side instantaneous voltage and instantaneous transmitted power. Therefore, this modeling method cannot accurately model DAB microinverters.
  • the Chinese invention patent "Negative Sequence Voltage Compensation Dual-loop Control Method and System Based on Virtual Synchronous Generator” with the authorization announcement number CN109193707B, in which the voltage loop uses a quasi-proportional resonant regulator to suppress the harmonic components of the output voltage, and the current loop uses proportional control to Speed up system response.
  • this method is applied to the output current closed-loop control of DAB microinverter, there are still the following problems:
  • the control loop of the DAB type microinverter only requires a current loop, and in order to achieve accurate control of the grid-connected current and suppression of the grid-connected current harmonics, a quasi-proportional resonant regulator needs to be used, using the ratio in the above patent.
  • the control cannot eliminate the control error well;
  • the current error generation link of the DAB micro-inverter is closely related to the positive and negative grid-side voltage, and the current error generation logic needs to be changed according to the sampling grid-side voltage, otherwise The control system will be unstable. Therefore, this method cannot effectively control the current of DAB microinverter.
  • the present invention provides a circuit modeling method and an output current control method of a dual active bridge microinverter.
  • a circuit modeling method of a dual-active bridge microinverter is provided.
  • the microinverter is equivalent to It is a standard dual active bridge (DAB) circuit, thereby further establishing the large signal model and small signal model of the microinverter, and completing the circuit modeling of the dual active bridge microinverter; among which:
  • DAB dual active bridge
  • the establishment of the equivalent circuit model of the dual active bridge microinverter includes:
  • the small signal model for establishing the microinverter is: third-order model Among them, x is the state variable of the small signal model; is the derivative of the state variable of the small signal model; u is the input variable of the small signal model; A 2 is the state variable coefficient matrix of the small signal model, with an order of 3 ⁇ 3; B 1 is the input variable coefficient matrix of the small signal model, with an order of The number is 3 ⁇ 4; the state variable x of the small signal model is:
  • the input variable u of the small signal model is: in, is the voltage disturbance value of the DC side bus capacitance Cbus of the equivalent circuit, is the disturbance value of the voltage of capacitor C g in the equivalent circuit, is the disturbance value of the output current of the equivalent circuit, is the disturbance value of the DC side input voltage of the equivalent circuit, is the disturbance value of the output voltage of the equivalent circuit, is the perturbation value of the internal phase shift angle, is the disturbance value of the external phase shift angle.
  • the voltage of the DC power supply V oc is the photovoltaic panel open circuit voltage; the DC source series resistance R pv is equal to the photovoltaic power at the maximum power point
  • the output voltage of the board is divided by the output current; the output currents of the controlled current source i ac,in and the controlled current source i ac,out are different according to the modulation mode of the dual active bridge microinverter. and change occurs;
  • v g is the output voltage of the equivalent circuit model, and the output current io of the equivalent circuit model is equal to ig ⁇ sgn(v g ), where ig is the sampled network
  • the side current, sgn(v g ) is the sign function of the grid side voltage.
  • the large-signal model of the microinverter is used to solve the changing relationship between the state variable X and the input variable U, and the obtained changing relationship is used to analyze the steady-state operating point of the circuit, The small signal model is solved and analyzed at the obtained steady-state operating point.
  • the small signal model of the microinverter is used to solve the transfer function from four input variables to three state variables, and is further used to analyze the parallelism of the microinverter under different circuit parameters and working conditions.
  • Network stability and dynamic performance among which:
  • I is a diagonal matrix with a diagonal element of 1 and an order of 3 ⁇ 3;
  • s is the Lagrange field symbol;
  • the expressions of matrices A 2 and B 2 are:
  • V bus is the DC bus capacitor voltage
  • R pv is the output voltage of the photovoltaic panel at the maximum power point divided by the output current
  • L g is the filter inductance on the AC grid side
  • the symbol variables ⁇ , ⁇ , and ⁇ are the same as the dual active Modulation mode related to bridge microinverter.
  • the transfer function from the input variable to the state variable is obtained.
  • the transfer function It can be used to describe the quantitative change relationship between the state variable Network stability is analyzed.
  • the modulation mode of the dual active bridge microinverter includes:
  • the angle between the negative rising edge of the square wave voltage on the primary side of the transformer and the positive rising edge of the square wave voltage on the primary side of the transformer is positioned as the internal phase shift angle D 1 , and the fundamental wave of the square wave voltage on the primary side of the transformer and the square wave voltage on the secondary side of the transformer are staggered.
  • the angle at which the fundamental wave is shifted is defined as the external phase shift angle D 2 , and the internal phase shift angle D 1 and the external phase shift angle D 2 are regarded as the two control degrees of freedom of the dual active bridge microinverter. , where the value range of D 1 is 0 ⁇ D 1 ⁇ 0.5, and the value range of D 2 is -0.5 ⁇ D 2 ⁇ 0.5;
  • the modulation mode of the transmission power is divided into mode one, mode two and mode three, where: when the external phase shift angle D 2 satisfies (1-D 1 )/2 ⁇ D 2 ⁇ 0.5 Or when -0.5 ⁇ D 2 ⁇ -(1-D 1 )/2, the positive level part of the primary side square wave voltage and the negative level part of the secondary side square wave voltage completely overlap, and the corresponding modulation mode is mode one.
  • the output current of the controlled current source i ac,in and the controlled current source i ac,out is close to a sine wave, and the output current has a maximum effective value; when the external phase shift angle D 2 satisfies 0 ⁇ D 2 ⁇ D 1 /2 or -D 1 /2 ⁇ D 2 ⁇ 0, the positive level part of the primary side square wave voltage and the positive level part of the secondary side square wave voltage completely overlap, and the corresponding modulation mode is mode three.
  • the output current of the controlled current source i ac,in and the controlled current source i ac,out is close to a triangular wave, and the effective value of the output current is the smallest; when the external phase shift angle D 2 satisfies D 1 /2 ⁇ D 2 When ⁇ (1-D 1 )/2 or -(1-D 1 )/2 ⁇ D 2 ⁇ -D 1 /2, part of the positive level of the primary side square wave voltage and the positive level of the secondary side square wave voltage The other part of the positive level of the primary side square wave voltage coincides with the negative level of the secondary side square wave voltage.
  • the corresponding modulation mode is mode two.
  • the controlled current source i ac,in and the receiver The output current of the controlled current source i ac,out is close to a trapezoidal wave, and the effective value of the output current is smaller than the effective value in mode one and greater than the effective value in mode three.
  • the symbolic variables ⁇ , ⁇ , and ⁇ are respectively:
  • f s is the switching frequency of the switching tube
  • L k is the value of the transformer leakage inductance converted to the secondary side
  • n is the turns ratio of the secondary primary side of the transformer
  • D 1 and D 2 are the internal and external phase shift angles respectively.
  • an output current control method of a dual active bridge microinverter is provided, based on the small signal model of the microinverter established by any of the circuit modeling methods described above,
  • the control logic of the current loop is determined according to the positive and negative voltage of the grid side.
  • the grid side current error is obtained through the determined control logic and input to the current closed-loop compensation controller.
  • the output of the current closed-loop compensation controller is the external phase shift of the microinverter.
  • Angle D 2 phase shift control is performed through the external phase shift angle D 2 to complete the closed-loop control of the output current of the dual active bridge microinverter;
  • the control logic of the current loop includes:
  • the grid-side current error is (i g -i g,ref ), where i g is the sampled grid-side current, and i g,ref is the grid-side current given value. ;
  • the grid-side current error is ( ig, ref -ig ).
  • the current closed-loop compensation controller adopts a quasi-proportional resonance controller.
  • the quasi-proportional resonance controller has three control parameters, namely: bandwidth adjustment parameter ⁇ c , proportion coefficient K p and resonance coefficient K r ; where :
  • the bandwidth adjustment parameter ⁇ c the resonance bandwidth used to change the frequency band characteristics of the current closed-loop compensation controller. The smaller the ⁇ c , the better the frequency selection characteristics, but the worse the ability to resist grid frequency disturbance;
  • the proportional coefficient K p is used to improve the dynamic characteristics of the dual active bridge microinverter control system.
  • the resonance coefficient K r is used to adjust the gain of the current closed-loop compensation controller.
  • the design method of the control parameters of the quasi-proportional resonant controller includes:
  • the current loop open-loop transfer function G io, d2 (s) from the external phase shift angle D 2 to the output current at the set time in the power frequency cycle is obtained; select The open-loop gain of the current loop at the power frequency frequency frequency ⁇ ac is greater than or equal to M; the crossover frequency of the open-loop transfer function of the current loop is selected to be ⁇ cr ; the phase margin of the open-loop transfer function of the current loop is selected to be greater than or equal to ⁇ margin ;
  • the current loop open-loop transfer function T c (j ⁇ ac ) G at the power frequency ⁇ ac is obtained io, d2 (j ⁇ ac ) ⁇ G QPR (j ⁇ ac ), where j is the imaginary unit.
  • the range of the proportional coefficient K p and the resonance coefficient K r can be solved, and the control parameters can be obtained accordingly.
  • the grid-side current error is (i g -i g, ref );
  • the grid-side current error is (ig , ref -ig ).
  • the present invention has at least one of the following beneficial effects:
  • the circuit modeling method of the dual active bridge microinverter provided by the present invention establishes a complete third-order microinverter large signal and small signal model, which on the one hand provides convenience for solving the steady-state operating point, and on the other hand
  • the small signal model can be used to further study the grid-connected stability and dynamic performance of the microinverter, and provide guidance for the design of control parameters.
  • the output current control method of the dual active bridge microinverter provided by the present invention provides correct current loop control logic and detailed current loop parameter design method, which is beneficial to improving the grid-connected stability and stability of the microinverter. Output current accuracy.
  • the circuit modeling method of the dual active bridge micro-inverter provided by the present invention can achieve accurate modeling of the DAB-type micro-inverter.
  • the output current control method of the dual active bridge type microinverter provided by the present invention can better control the DAB type microinverter. Perform current control.
  • Figure 1 is a schematic circuit diagram of a single-stage half-bridge DAB microinverter in a preferred embodiment of the present invention
  • Figure 2 shows a preferred embodiment of the present invention, when the transmission power direction is from the DC side to the AC side and the grid side voltage is positive, the driving signals of the switch tubes S1 to S8 in the three modulation modes, as well as the transformer primary side voltage, transformer Waveform diagram of secondary voltage and transformer secondary current;
  • Figure 3 is a schematic diagram of the internal phase shift angle and external phase shift angle ranges corresponding to three modulation modes in a preferred embodiment of the present invention
  • Figure 4 is a schematic equivalent circuit diagram of a dual active bridge microinverter in a preferred embodiment of the present invention.
  • Figure 5 is an overall control block diagram of a dual active bridge microinverter in a preferred embodiment of the present invention.
  • One embodiment of the present invention provides a circuit modeling method for a dual-active bridge microinverter.
  • the method establishes an equivalent circuit model of the dual-active bridge microinverter and converts the microinverter into an equivalent circuit. It is a standard dual active bridge circuit (DAB circuit), thereby further establishing the third-order large signal model and small signal model of the microinverter, and completing the circuit modeling of the dual active bridge microinverter, so as to It is used to solve the circuit state variables and then analyze the grid-connected stability and dynamic performance of the microinverter.
  • DAB circuit standard dual active bridge circuit
  • the dual active bridge microinverter includes: photovoltaic panels, DC side bus capacitors, primary side square wave generating circuit, high frequency transformer, secondary side square wave generating circuit, filter capacitor C g , AC
  • the positive and negative poles of the ports are connected.
  • the positive and negative poles of the AC side ports of the primary square wave generating circuit are connected to the positive and negative poles of the input ports of the high-frequency transformer.
  • the AC side ports of the secondary square wave generating circuit are respectively connected to the output ports of the high-frequency transformer.
  • the positive and negative poles are connected, and the positive pole of the DC side port of the secondary square wave generating circuit is in phase with the filter inductor L g on the AC grid side.
  • the filter inductor Lg on the AC grid side is connected in series with the series damping resistor R L and is connected to the positive pole of the AC grid; the negative pole of the DC side port of the secondary square wave generating circuit is connected to the negative pole of the AC grid; among them, the primary and secondary sides of the high-frequency transformer
  • the turns ratio is 1:n, which translates into the leakage inductance of the secondary side of the double active bridge microinverter transformer being L k .
  • the primary side of the dual active bridge microinverter transformer can output three levels ⁇ v pv ,0,-v pv ⁇ , and the secondary side can output two levels ⁇
  • v pv is the photovoltaic panel terminal voltage
  • is the absolute value of the grid side voltage.
  • the dual active bridge microinverter adjusts the output power through phase shift modulation.
  • the microinverter has two degrees of control freedom, namely the internal phase shift angle D 1 of the primary circuit and the external phase shift angle of the primary secondary circuit.
  • Phase shift angle D 2 where the internal phase shift angle is defined as the angle between the negative rising edge of the primary side square wave voltage and the positive rising edge of the primary side square wave voltage; the external phase shift angle is defined as the fundamental sum of the primary side square wave voltage of the transformer The angle at which the fundamental wave of the square wave voltage on the secondary side of the transformer is shifted.
  • the value range of D 1 is 0 ⁇ D 1 ⁇ 0.5
  • the value range of D 2 is -0.5 ⁇ D 2 ⁇ 0.5;
  • the dual-active bridge microinverter realizes the phase-shifting modulation mode of adjusting the output power.
  • D 1 and the external phase-shifting angle D 1 The value range of the phase shift angle D 2 is divided.
  • D 2 satisfies (1-D 1 )/2 ⁇ D 2 ⁇ 0.5 or -0.5 ⁇ D 2 ⁇ -(1-D 1 )/2, the positive level part of the primary side voltage and the negative level of the secondary side voltage Parts completely overlap, and the corresponding modulation mode is mode 1.
  • the transformer current is close to a sine wave, and the effective value of the transformer current is the largest, but the zero-voltage soft switching of the original and secondary switch tubes is easiest to implement; when D 2 satisfies D 1 /2 ⁇ D 2 ⁇ (1-D 1 )/2 or -(1-D 1 )/2 ⁇ D 2 ⁇ -D 1 /2, part of the positive level of the primary voltage coincides with the positive level of the secondary voltage, and the other part Part of it coincides with the negative level of the secondary voltage, and the corresponding modulation mode is mode 2.
  • the transformer current is close to a trapezoidal wave, and the effective value of the transformer current is moderate, that is, it is smaller than the effective value of mode 1 and greater than the effective value of mode 3.
  • the original and secondary The zero-voltage soft switching of the edge switch tube is easier to realize, that is, the ease is less than the ease of mode one and greater than the ease of mode three; when the external phase shift angle D 2 satisfies 0 ⁇ D 2 ⁇ D 1 /2 or -D When 1 /2 ⁇ D 2 ⁇ 0, the positive level part of the primary voltage and the positive level part of the secondary voltage completely overlap, and the corresponding modulation mode is mode three. At this time, the transformer current is close to a triangle wave, and the effective value of the transformer current is minimum. , but the zero-voltage soft switching of the original secondary switch tube is the most difficult to achieve.
  • establishing the equivalent circuit model of the dual active bridge microinverter includes: constructing the DC power supply V oc , the DC source series resistance R pv , the DC side bus capacitance C bus , and the controlled current source i ac,in , an ideal transformer with a primary and secondary turns ratio of 1:n, a controlled current source i ac,out , a capacitor C g , a filter inductor L g on the AC grid side, an inductor winding resistance R L and a DC pulsating power supply
  • the positive pole of the DC power supply V oc is connected to one end of the DC source series resistor R pv .
  • the other end of the DC source series resistor R pv is connected to the positive pole of the DC side bus capacitor C bus .
  • the positive pole of the DC side bus capacitor C bus is also connected. It is connected to the positive pole of the primary side of the ideal transformer, the negative pole of the primary side of the ideal transformer is connected to the negative pole of the DC power supply V oc , and the positive pole of the secondary side of the ideal transformer is connected to the controlled The negative pole of the current source i ac,in is connected.
  • the negative pole of the secondary side of the ideal transformer is connected to the positive pole of the controlled current source i ac,in . It is also connected to the negative pole of the controlled current source i ac,out .
  • the controlled current source i ac the positive electrode of out is connected to the positive electrode of capacitor C g , and at the same time connected to one end of the filter inductor L g on the AC grid side.
  • the other end of the inductor L g on the AC grid side is connected to one end of the inductor winding resistor R L.
  • the inductor winding The other end of the line resistor RL is connected to the positive pole of the DC pulsating power supply
  • the voltage of the DC power supply V oc is the open circuit voltage of the photovoltaic panel; the DC source series resistance R pv is equal to the output voltage of the photovoltaic panel at the maximum power point divided by the output current; the controlled current source i ac,in and the controlled current source i ac,out
  • the output current i.e., the transformer current
  • the output current changes according to the modulation mode of the dual active bridge microinverter.
  • the output voltage of the equivalent circuit constructed is
  • the large signal model of the microinverter is a third-order average model.
  • X is the state variable of the large signal model; is the derivative of the state variable of the large signal model with respect to time; U is the input variable of the large signal model; A 1 is the state variable coefficient matrix of the large signal model, with an order of 3 ⁇ 3; B 1 is the input variable coefficient matrix of the large signal model , the order is 3 ⁇ 2.
  • ⁇ v bus > is the average value of the voltage of the DC side bus capacitance C bus in the equivalent circuit during the switching cycle
  • ⁇ v cg > is the average value of the voltage of the capacitor C eq in the equivalent circuit during the switching cycle
  • ⁇ i o > is the average value of the output current of the equivalent circuit during the switching period
  • ⁇ v dc > is the average value of the DC side input voltage of the equivalent circuit during the switching period
  • ⁇ v o> is the output voltage of the equivalent circuit during average value over the switching cycle.
  • the large-signal model of the microinverter is used to solve the changing relationship between the state variable X and the input variable U.
  • the obtained changing relationship is used to analyze the steady-state operating point of the circuit. Solve and analyze the small signal model at the dynamic operating point.
  • the state variable coefficient matrix of the model has an order of 3 ⁇ 3; B 1 is the input variable coefficient matrix of the small signal model, and the order is 3 ⁇ 4.
  • the state variables of the small signal model are The input variables of the small signal model are in is the voltage disturbance value of the DC side bus capacitance Cbus of the equivalent circuit, is the disturbance value of the voltage of capacitor C g in the equivalent circuit, is the disturbance value of the output current of the equivalent circuit, is the disturbance value of the DC side input voltage of the equivalent circuit, is the disturbance value of the output voltage of the equivalent circuit, is the perturbation value of the internal phase shift angle, is the disturbance value of the external phase shift angle.
  • the small-signal model of the dual-active bridge microinverter can be used to solve the transfer function from four input variables to three state variables, and further used to analyze the grid-connected stability of the microinverter under different circuit parameters and working conditions. and dynamic performance.
  • I is a diagonal matrix with a diagonal element of 1 and an order of 3 ⁇ 3;
  • s is the Lagrange field symbol;
  • the expressions of matrices A 2 and B 2 are:
  • V bus is the DC bus capacitor voltage
  • R pv is the output voltage of the photovoltaic panel at the maximum power point divided by the output current
  • L g is the filter inductance on the AC grid side
  • the symbol variables ⁇ , ⁇ , ⁇ are related to the dual active bridge type Related to the modulation mode of the microinverter.
  • each symbolic variable is:
  • f s is the switching frequency of the switching tube
  • L k is the value of the transformer leakage inductance converted to the secondary side
  • n is the turns ratio of the secondary primary side of the transformer
  • D 1 and D 2 are the internal and external phase shift angles respectively.
  • the transfer function can be used to describe the quantitative change relationship between the state variable and the input variable at different frequencies.
  • the response of the state variable can be studied for various forms of input variables, so as to analyze the micro-inverter.
  • the grid connection stability is analyzed.
  • the method to further analyze the dynamic performance of the microinverter under different circuit parameters and working conditions is to change the three parameters of photovoltaic terminal voltage vpv , grid voltage amplitude Vm and grid-connected power Po , in each group Under the parameters, the transfer function from input variables to state variables is solved based on the small signal model of the microinverter, and the dynamic performance of the microinverter is analyzed based on the gain, crossover frequency, amplitude margin and phase margin of the transfer function.
  • the circuit modeling method provided by the above embodiments of the present invention equates the dual active bridge microinverter to a standard dual active bridge circuit by establishing an equivalent circuit model, thereby establishing a third-order microinverter.
  • the large-signal model and the small-signal model are used to solve the circuit state variables, and then analyze the grid-connected stability and dynamic performance of the microinverter.
  • This circuit modeling method establishes a complete large-signal and small-signal model of the microinverter. On the one hand, it provides convenience for solving the steady-state operating point. On the other hand, the transfer function obtained by using the small-signal model improves the dynamic performance of the microinverter. research is facilitated.
  • An embodiment of the present invention also provides an output current control method of a dual active bridge microinverter. Based on the small signal model of the microinverter established by the circuit modeling method in any of the above embodiments, according to The positive and negative of the grid-side voltage determines the control logic of the current loop. The grid-side current error obtained through the correct control logic is input to the current closed-loop compensation controller. Its output is the external phase shift angle D 2 of the microinverter, which can be used for shifting. Phase control to complete the closed-loop control of the output current of the dual active bridge microinverter. in:
  • control logic of the current loop is as follows:
  • the grid-side current error is (i g -i g, ref ), where i g is the sampled grid-side current, and i g, ref is the grid-side current given value;
  • the grid-side current error is (i g, ref -i g ).
  • the current closed-loop compensation controller is a quasi-proportional resonance (Q-PR) controller, which has three designable control parameters, namely 1 bandwidth adjustment parameter ⁇ c , 2 proportional coefficient K p , and 3 resonance coefficient K r .
  • Q-PR quasi-proportional resonance
  • 1Bandwidth adjustment parameter ⁇ c The resonance bandwidth used to change the frequency band characteristics of the controller. The smaller the ⁇ c , the better the frequency selection characteristics, but the worse the ability to resist grid frequency disturbance;
  • K p Proportional coefficient K p : used to improve the dynamic characteristics of the dual active bridge microinverter control system. The larger the K p , the shorter the reaction time, but the larger the overshoot;
  • K r used to adjust the gain of the controller.
  • K r the greater the gain of the controller is.
  • Step 1 According to the small signal model of the dual active bridge microinverter, obtain the transfer function G io, d2 (s) from the external phase shift angle to the output current at 3ms in the power frequency cycle; select the power frequency frequency ⁇ ac
  • the open-loop gain of the current loop is greater than or equal to M
  • the crossover frequency of the open-loop transfer function of the current loop is selected to be ⁇ cr
  • the phase margin of the open-loop transfer function of the current loop is selected to be greater than or equal to ⁇ margin ;
  • Step 4 According to the property that the gain of the current loop is 1 at the crossover frequency, an equality constraint relationship including K p and K r is obtained
  • 1;
  • Step 5 Based on the condition that the phase margin of the current loop at the crossover frequency is greater than or equal to ⁇ margin , an inequality constraint relationship ⁇ +angle [G io , d2 (j ⁇ cr ) ⁇ G QPR (j ⁇ cr )] ⁇ margin .
  • the current error of the input current closed-loop compensation controller is set to 0, and the calculation of the current error in the remaining time periods is performed according to normal principles. Specifically, When the grid voltage v g is less than -10V, the grid-side current error is (i g -i g, ref ); when the grid voltage v g is greater than 10V, the grid-side current error is (i g, ref - i g ).
  • the output current control method provided by the above embodiments of the present invention is a closed-loop control method of output current. Based on the small signal model of the dual active bridge microinverter, the correct current loop control logic is established and the current Ring parameter design method. This output current control method is conducive to improving the grid-connected stability and output current accuracy of the microinverter by providing correct current loop control logic and detailed current loop parameter design methods.
  • Figure 1 is a schematic circuit diagram of a single-stage half-bridge DAB microinverter.
  • the single-stage half-bridge DAB microinverter circuit is composed of photovoltaic panel components, DC bus capacitors, primary-side full-bridge circuit, high-frequency transformer, secondary-side half-bridge circuit, and grid-side low-pass filter.
  • the primary side full-bridge circuit includes switching tubes S1 ⁇ S4, and the secondary side half-bridge circuit includes switching tubes S5 ⁇ S8 and film capacitors C1/C2; the primary and secondary turns ratio of the high-frequency transformer is 1:n, which is converted to the excitation of the primary side.
  • the inductance is L m
  • the leakage inductance of the transformer converted to the secondary side is L k .
  • FIG 2 shows the driving waveforms of the switching transistors S1 to S8 in the three modulation modes of the single-stage half-bridge DAB microinverter in Embodiment 1 of the present invention and the voltage and current waveforms of the primary and secondary sides of the transformer.
  • the basic working mode of the single-stage half-bridge DAB microinverter is: the switching tubes S1 and S2 are complementary to each other at high frequency, and the switching tubes S3 and S4 are complementary to each other at high frequency.
  • the primary side of the single-stage half-bridge DAB microinverter transformer can output three levels ⁇ v pv ,0,-v pv ⁇ , and the secondary side can output two levels ⁇
  • v pv is the photovoltaic panel terminal voltage
  • is the absolute value of the grid side voltage.
  • the single-stage half-bridge DAB microinverter adjusts the output power through phase-shift modulation.
  • the microinverter has two degrees of control freedom, which are the internal phase-shift angle D 1 of the primary circuit and the original The external phase shift angle D 2 of the secondary circuit, where the internal phase shift angle is defined as the angle between the driving pulse of switch S4 and the driving pulse of switch S1; the external phase shift angle is defined as the fundamental wave of the square wave voltage of the primary side of the transformer. The angle offset from the fundamental wave of the square wave voltage on the secondary side of the transformer.
  • the value range of D 1 is 0 ⁇ D 1 ⁇ 0.5
  • the value range of D 2 is -0.5 ⁇ D 2 ⁇ 0.5.
  • Figure 3 is a schematic diagram of the internal phase shift angle and external phase shift angle ranges corresponding to the three modulation modes in Embodiment 1 of the present invention.
  • the corresponding modulation mode is mode one; when D 2 satisfies D 1 /2 ⁇ D 2 ⁇ (1-D 1 )/2 or -(1- When D 1 )/2 ⁇ D 2 ⁇ -D1/2, the corresponding modulation mode is mode 2; when the external phase shift angle D 2 satisfies 0 ⁇ D 2 ⁇ D 1 /2 or -D 1 /2 ⁇ D 2 ⁇ When 0, the corresponding modulation mode is mode three.
  • Figure 4 is the equivalent circuit of the single-stage half-bridge DAB micro-inverter in Embodiment 1 of the present invention.
  • the equivalent circuit of the micro-inverter consists of a DC power supply V oc and a DC source series resistor R pv , DC side bus capacitance C bus , controlled current source i ac,in , ideal transformer with primary and secondary turns ratio of 1:n, controlled current source i ac,out , capacitor C eq , filter inductor on the AC grid side It is composed of L g and inductor winding resistance RL .
  • the voltage of the DC power supply V oc is the open circuit voltage of the photovoltaic panel; the DC source series resistance R pv is equal to the output voltage of the photovoltaic panel at the maximum power point divided by the output current; the controlled current source i ac,in and the controlled current source i ac, The output current of out changes according to the different modulation modes; the capacitor C eq is the series connection of C1 and C2; the output voltage of the equivalent circuit is
  • the large-signal model of the circuit is the third-order average model where X is the state variable of the large signal model; is the derivative of the state variable of the large signal model with respect to time; U is the input variable of the large signal model; A 1 is the state variable coefficient matrix of the large signal model, with an order of 3 ⁇ 3; B 1 is the input variable coefficient matrix of the large signal model , the order is 3 ⁇ 2.
  • ⁇ v bus > is the average value of the DC bus capacitor voltage during the switching cycle
  • ⁇ v ceq > is the average value of the voltage of the capacitor C eq in the equivalent circuit during the switching cycle
  • ⁇ i o > is the output of the equivalent circuit
  • ⁇ v dc > is the average value of the DC side input voltage of the equivalent circuit during the switching period
  • ⁇ v o > is the average value of the output voltage of the equivalent circuit during the switching period.
  • ⁇ is related to the modulation mode of the microinverter.
  • the state variables of the small signal model are The input variables of the small signal model are in is the disturbance value of the DC bus capacitor voltage, is the disturbance value of the voltage of capacitor C eq in the equivalent circuit, is the disturbance value of the output current of the equivalent circuit, is the disturbance value of the DC side input voltage of the equivalent circuit, is the disturbance value of the output voltage of the equivalent circuit, is the perturbation value of the internal phase shift angle, is the disturbance value of the external phase shift angle.
  • each symbol variable is:
  • I is a diagonal matrix with a diagonal element of 1, and the order is 3 ⁇ 3; s is the Lagrange field symbol.
  • FIG. 5 shows the overall control block diagram of the single-stage half-bridge DAB microinverter in the second embodiment of the present invention.
  • the control block diagram is divided into six parts: maximum power point tracking (MPPT), voltage loop, phase locked loop, current loop, feedforward control and phase shift control.
  • MPPT maximum power point tracking
  • the controller After the controller samples the terminal voltage v pv and the output current i pv of the photovoltaic panel, it generates the photovoltaic panel terminal voltage given value V pv, ref ; V pv, ref and the actual sampling voltage v pv after going through the maximum power point tracking (MPPT) link.
  • MPPT maximum power point tracking
  • the controller After making the difference, the amplitude of the grid-side current given value I m is generated through the voltage loop; the controller samples the grid-side voltage v g and generates the grid-side voltage phase ⁇ through the phase-locked loop; ⁇ is multiplied by I m after sinusoidal transformation to obtain The grid-side current given value i g,ref is different from the actual grid-side current sampling value i g .
  • the external phase shift angle D 2 is obtained through the current closed-loop compensation controller; the internal phase shift angle D 1
  • the drive signals S1 ⁇ S4 are generated through the phase shift controller, and the external phase shift angle D2 is used to generate drive signals S5 ⁇ S8 through the phase shift controller, so that the single-stage half-bridge DAB microinverter outputs a given power.
  • control logic of the current loop is as follows:
  • the grid-side current error is (i g -i g, ref ), where i g is the sampled grid-side current, and i g, ref is the grid-side current given value;
  • the grid-side current error is (i g, ref -i g ).
  • the current closed-loop compensation controller is a quasi-proportional resonance (Q-PR) controller, which has three designable control parameters, namely 1 bandwidth adjustment parameter ⁇ c , 2 proportional coefficient K p , and 3 resonance coefficient K r .
  • Q-PR quasi-proportional resonance
  • 1Bandwidth adjustment parameter ⁇ c The resonance bandwidth used to change the frequency band characteristics of the controller. The smaller the ⁇ c , the better the frequency selection characteristics, but the worse the ability to resist grid frequency disturbance;
  • K p Proportional coefficient K p : used to improve the dynamic characteristics of the system. The larger K p, the shorter the reaction time, but the larger the overshoot;
  • K r used to adjust the gain of the controller.
  • K r the greater the gain of the controller is.
  • Step 1 Based on the small signal model of the single-stage half-bridge DAB microinverter, obtain the transfer function G io, d2 (s) from the external phase shift angle to the output current at 3ms in the power frequency cycle; select ⁇ ac The open-loop gain of the current loop is greater than or equal to M; select the current The crossover frequency of the open-loop transfer function is ⁇ cr ; select the phase margin of the open-loop transfer function of the current loop to be greater than or equal to ⁇ margin ;
  • Step 4 According to the property that the gain of the current loop is 1 at the crossover frequency, an equality constraint relationship including K p and K r is obtained
  • 1;
  • Step 5 According to the condition that the phase margin of the current loop at the crossover frequency is greater than or equal to ⁇ margin, an inequality constraint relationship ⁇ +angle [G io , d2 (j ⁇ cr ) ⁇ G QPR (j ⁇ cr )] ⁇ margin .
  • the ranges of the proportional coefficient K p and the resonance coefficient K r can be solved, and appropriate control parameters can be selected accordingly.
  • the circuit modeling method of the dual active bridge microinverter in the above embodiment of the present invention establishes a complete third-order microinverter large signal and small signal model. On the one hand, it provides convenience for solving the steady-state operating point. On the other hand, the small signal model can further study the dynamic performance of the microinverter and provide guidance for the design of control parameters.
  • the output current control method of the dual active bridge microinverter in the above embodiment of the present invention provides correct current loop control logic and detailed current loop parameter design method, which is beneficial to improving the grid connection stability of the microinverter. performance and output current accuracy.

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Abstract

本发明提供了一种双有源桥型微逆变器的电路建模方法，该方法通过建立等效电路模型，将双有源桥型微逆变器等效为标准的双有源桥电路，从而建立起微逆变器的三阶大信号模型和小信号模型，以便于进行电路状态变量的求解和动态性能的分析，该方法一方面为求解稳态工作点提供了便利，另一方面利用小信号模型得到的传递函数为微逆变器动态性能的研究提供了便利。本发明还提供了一种双有源桥型微逆变器的输出电流控制方法，该方法基于双有源桥型微逆变器的小信号模型，建立了正确的电流环控制逻辑，并给出了电流环参数的设计方法。该方法通过给出正确的电流环控制逻辑和详尽的电流环参数设计方法，有利于改善微逆变器的并网稳定性和输出电流精度。

Description

双有源桥型微逆变器的电路建模方法及输出电流控制方法 技术领域
本发明涉及光伏微逆变器技术领域，具体地，涉及一种适用于双有源桥型微逆变器的电路建模方法以及输出电流控制方法。
背景技术
微逆变器一般指是光伏发电系统中的功率小于等于1000W，且具备组件级最大功率点追踪能力的逆变器。与集中式和组串式光伏逆变系统不同，微逆变器直接与单个光伏组件进行连接。其优点是可以对每块组件进行独立的MPPT控制，在大幅提高整体效率的同时，也可以避免集中式逆变器存在的直流高压、弱光效应差、木桶效应等问题。
根据直流母线的位置和结构特点，可以将微逆变器分为三大类：直流母线结构，伪直流母线结构和无直流母线结构。其中无直流母线的微逆变器为单级式电路，采用矩阵式控制，其所用开关器件数量少，转换效率高，因此更具有优势。而在无直流母线结构的微逆变器中，双有源桥(Dual active bridge,DAB)型微逆变器使用的开关器件数量最少，在具备DAB电路宽范围软开关的特性基础上，还改善了DAB电路轻载时效率较低的问题。
目前已有文献对标准的DAB电路进行建模，但是该电路模型仅适用于DC-DC型DAB变换器，无法适用于DAB型微逆变器。此外，目前文献中DAB型微逆变器的电流环闭环控制补偿器为比例控制器或比例-积分控制器，且电流环控制逻辑没有跟随网侧电压变化而发生变化。因此有必要提供一种适用于DAB型微逆变器的电路建模方法，并基于此建模方法提供合理的电流环控制逻辑，以实现对微逆变器电路的闭环控制，改善输出电流精度和并网稳定性。
经过检索发现：
公开号为CN109408904A的中国发明专利申请《一种并网非对称级联H桥变换器系统小信号建模方法》，依据并网非对称级联H桥多电平变换器简化模型，推导功率电路小信号模型；推导控制系统中电网电流ig、电网电流参考信号ig,ref、滤波电容电流参考信号及调制信号um等的扰动微分方程；计算并网非对称级联H桥多电平变换器输出电压vi，推导不含锁相环 的系统小信号模型；推导锁相环的小信号模型，建立整个并网非对称级联H桥多电平变换器系统小信号模型。该方法同样适用于离网型非对称级联H桥多电平变换器系统。但是该方法如果应用于DAB型微逆变器的电路建模中，仍然存在如下问题：
首先DAB型微逆变器的属于高频链式交直流变换系统，其工作原理和级联H桥多电平变换器有本质不同，DAB型微逆变器的高频变压器原副边电压和变压器电流均为交流成分，因此无法直接通过上述专利中的建模方法进行电路建模，而是需要对这些交流电气量进行等效；其次，与级联H桥多电平变换器不同，DAB型微逆变器的传递函数增益会随直流侧电压、交流侧瞬时电压和瞬时传输功率的变化而改变。因此，该建模方法无法对DAB型微逆变器进行准确建模。
授权公告号为CN109193707B的中国发明专利《基于虚拟同步发电机的负序电压补偿双环控制方法及系统》，其中电压环采用准比例谐振调节器抑制输出电压的谐波分量，电流环采用比例控制以加快系统响应速度。但是该方法如果应用于DAB型微逆变器的输出电流闭环控制中，仍然存在如下问题：
首先DAB型微逆变器的控制回路中仅需要电流环，且为了实现对并网电流的准确控制以及对并网电流谐波的抑制，需要使用准比例谐振调节器，采用上述专利中的比例控制无法很好地消除控制误差；其次，与上述专利不同，DAB型微逆变器的电流误差生成环节与网侧电压正负密切相关，需要根据采样网侧电压改变电流误差的产生逻辑，否则控制系统会不稳定。因此，该方法不能对DAB型微逆变器进行有效的电流控制。
发明内容
本发明针对现有技术中存在的上述不足，提供了一种双有源桥型微逆变器的电路建模方法及输出电流控制方法。
根据本发明的一个方面，提供了一种双有源桥型微逆变器的电路建模方法，通过建立双有源桥型微逆变器的等效电路模型，将微逆变器等效为标准的双有源桥(DAB)型电路，从而进一步建立微逆变器的大信号模型和小信号模型，完成对双有源桥型微逆变器的电路建模；其中：
所述建立双有源桥型微逆变器的等效电路模型，包括：
构建直流电源Voc、直流源串联电阻Rpv、直流侧母线电容Cbus、受控电流源iac,in、原副边匝比为1:n的理想变压器、受控电流源iac,out、电容Cg、交流电网侧的滤波电感Lg、电感绕 线电阻RL以及直流脉动电源|vg|；所述直流电源Voc的正极与所述直流源串联电阻Rpv的一端相连，所述直流源串联电阻Rpv的另一端与所述直流侧母线电容Cbus的正极相连，所述直流侧母线电容Cbus的正极同时与所述理想变压器原边的正极相连，所述理想变压器原边的负极与所述直流电源Voc的负极相连，所述理想变压器副边的正极与所述受控电流源iac,in的负极相连，所述理想变压器副边的负极与所述受控电流源iac,in的正极相连，并同时与所述受控电流源iac,out的负极相连，所述受控电流源iac,out的正极与所述电容Cg的正极相连，并同时与所述交流电网侧的滤波电感Lg的一端相连，所述交流电网侧的电感Lg的另一端与所述电感绕线电阻RL的一端相连，所述电感绕线电阻RL的另一端与所述直流脉动电源|vg|的正极相连，所述直流脉动电源|vg|的负极与所述电容Cg的负极相连；
所述建立微逆变器的大信号模型为：三阶平均值模型其中，X为大信号模型的状态变量；为大信号模型状态变量对时间的导数；U为大信号模型的输入变量；A1为大信号模型的状态变量系数矩阵，阶数为3×3；B1为大信号模型的输入变量系数矩阵，阶数为3×2；所述大信号模型的状态变量X为：X＝[<vbus>，<vcg>，<io>]T；所述大信号模型的输入变量U为：U＝[<vdc>，<vo>]T；其中，<vbus>为等效电路的直流侧母线电容Cbus的电压在开关周期内的平均值，<vcg>为等效电路中电容Cg的电压在开关周期内的平均值，<io>为等效电路的输出电流在开关周期内的平均值，<vdc>为等效电路的直流侧输入电压在开关周期内的平均值，<vo>为等效电路的输出电压在开关周期内的平均值；
所述建立微逆变器的小信号模型为：三阶模型其中，x为小信号模型的状态变量；为小信号模型状态变量的导数；u为小信号模型的输入变量；A2为小信号模型的状态变量系数矩阵，阶数为3×3；B1为小信号模型的输入变量系数矩阵，阶数为3×4；所述小信号模型的状态变量x为：所述小信号模型的输入变量u为：其中，为等效电路的直流侧母线电容Cbus的电压的扰动值，为等效电路中电容Cg的电压的扰动值，为等效电路的输出电流的扰动值，为等效电路的直流侧输入电压的扰动值，为等效电路的输出电压的扰动值，为内移相角的扰动值，为外移相角的扰动值。
可选地，所述双有源桥型微逆变器的等效电路模型中，所述直流电源Voc的电压为光伏板开路电压；所述直流源串联电阻Rpv等于最大功率点处光伏板的输出电压除以输出电流；所述受控电流源iac,in和所述受控电流源iac,out的输出电流根据所述双有源桥型微逆变器的调制模式的不同而发生变化；
所述直流脉动电源|vg|为所述等效电路模型的输出电压，所述等效电路模型的输出电流io等于ig·sgn(vg)，其中，ig为采样得到的网侧电流，sgn(vg)为网侧电压的符号函数。
可选地，所述微逆变器的大信号模型，用于求解所述状态变量X随所述输入变量U的变化关系，求解得到的所述变化关系用于分析电路的稳态工作点，在得到的所述稳态工作点处对所述小信号模型进行求解和分析。
可选地，所述微逆变器的小信号模型，用于求解从四个输入变量到三个状态变量的传递函数，进一步用于分析微逆变器在不同电路参数和工况下的并网稳定性和动态性能；其中：
所述四个输入变量为所述三个状态变量为所述传递函数的表达式为：
x＝(sI-A2)-1B2u
其中I为对角线元素为1的对角矩阵，阶数为3×3；s为拉氏域符号；矩阵A2和B2的表达式为：

其中，Vbus为直流母线电容电压，Rpv为最大功率点处光伏板的输出电压除以输出电流，Lg为交流电网侧的滤波电感，符号变量α、β、γ与所述双有源桥型微逆变器的调制模式相关。
对于分析微逆变器在不同电路参数和工况下的并网稳定性和动态性能，包括：
根据微逆变器的小信号模型求解得到从输入变量到状态变量的传递函数，所述传递函数 可以用于描述不同频率下状态变量x随输入变量u的定量变化关系，当传递函数已知时，则可以针对各种不同形式的输入变量研究状态变量的响应，以便对微逆变器的并网稳定性进行分析。
改变光伏端电压vpv、电网电压幅值Vm和并网功率Po这三个参数，在每一组参数(vpv，Vm，Po)下，根据微逆变器的小信号模型求解得到从输入变量到状态变量的传递函数，并根据所述传递函数的增益、穿越频率、幅值裕度和相位裕度分析微逆变器的动态性能。
可选地，所述双有源桥型微逆变器的调制模式，包括：
将变压器原边方波电压负上升沿和变压器原边方波电压正上升沿错开的角度定位为内移相角D1，将变压器原边方波电压的基波和变压器副边方波电压的基波错开的角度定义为外移相角D2，将所述内移相角D1和所述外移相角D2作为所述双有源桥型微逆变器的两个控制自由度，其中，D1的取值范围为0≤D1≤0.5，D2的取值范围为-0.5≤D2≤0.5；
根据根据两个控制自由度，将传输功率的调制模式划分为模式一、模式二和模式三，其中：当所述外移相角D2满足(1-D1)/2＜D2≤0.5或-0.5＜D2≤-(1-D1)/2时，原边方波电压的正电平部分和副边方波电压的负电平部分完全重合，对应调制模式为模式一，此时所述受控电流源iac,in和所述受控电流源iac,out的输出电流接近正弦波，输出电流有效值最大；当所述外移相角D2满足0≤D2≤D1/2或-D1/2≤D2≤0时，原边方波电压的正电平部分和副边方波电压的正电平部分完全重合，对应的调制模式为模式三，此时所述受控电流源iac,in和所述受控电流源iac,out的输出电流接近三角波，输出电流有效值最小；当所述外移相角D2满足D1/2＜D2≤(1-D1)/2或-(1-D1)/2＜D2≤-D1/2时，原边方波电压的正电平的一部分和副边方波电压的正电平重合，原边方波电压的正电平的另一部分和副边方波电压的负电平重合，对应的调制模式为模式二，此时所述受控电流源iac,in和所述受控电流源iac,out的输出电流接近梯形波，输出电流有效值小于模式一时的有效值且大于模式三时的有效值。
可选地，对于所述模式二，所述符号变量α、β、γ分别为：
对于所述模式三，所述符号变量α、β、γ分别为：
其中，fs为开关管开关频率，Lk为变压器漏感折算到副边的值，n为变压器副原边匝比，D1和D2分别为内外移相角。
根据本发明的另一个方面，提供了一种双有源桥型微逆变器的输出电流控制方法，基于上述任一项所述的电路建模方法建立的微逆变器的小信号模型，根据网侧电压正负确定电流环的控制逻辑，通过确定的控制逻辑得到网侧电流误差并输入到电流闭环补偿控制器，所述电流闭环补偿控制器的输出为微逆变器的外移相角D2，通过所述外移相角D2进行移相控制，完成对双有源桥型微逆变器的输出电流闭环控制；其中：
所述电流环的控制逻辑包括：
当电网电压vg小于0时，所述网侧电流误差为(ig-ig，ref)，其中，ig为采样的到的网侧电流，ig,ref为网侧电流给定值；
当电网电压vg大于0时，所述网侧电流误差为(ig，ref-ig)。
可选地，所述电流闭环补偿控制器采用准比例谐振控制器，所述准比例谐振控制器具有三个控制参数，分别为：带宽调节参数ωc、比例系数Kp和谐振系数Kr；其中：
所述带宽调节参数ωc：用于改变所述电流闭环补偿控制器频带特性的谐振带宽，ωc越小则选频特性越好，但是抗电网频率扰动的能力越差；
所述比例系数Kp：用于改善双有源桥型微逆变器控制系统的动态特性，Kp越大反应时间越短，但是超调量越大；
所述谐振系数Kr：用于调节所述电流闭环补偿控制器的增益，Kr越大则控制器增益越大。
可选地，所述准比例谐振控制器的控制参数的设计方法，包括：
根据双有源桥型微逆变器的小信号模型，得到工频周期内设定时间处从外移相角D2到输出电流的电流环开环传递函数Gio，d2(s)；选择工频频率ωac处电流环的开环增益大于等于M；选择电流环开环传递函数的穿越频率为ωcr；选择电流环开环传递函数的相位裕度大于等于φmargin；
根据电网角频率最大允许变化范围[-Δωmax，Δωmax]，取ωc＝Δωmax，其中Δωmax为电网角频率的最大允许变化量；
根据电流环开环传递函数Gio，d2(s)和准比例谐振控制器的传递函数GQPR(s)得到工频频率ωac处的电流环开环传递函数Tc(jωac)＝Gio，d2(jωac)·GQPR(jωac)，其中j为虚数单位，令工频频率ωac处的电流环开环传递函数Tc(jωac)的增益大于等于M，得到一个包含比例系数Kp和谐振系数Kr的不等式约束关系|Gio，d2(jωac)·GQPR(jωac)|≥M；
根据电流环在穿越频率ωcr处增益为1的性质，得到一个包含比例系数Kp和谐振系数Kr的等式约束关系|Gio，d2(jωcr)·GQPR(jωcr)|＝1；
根据电流环在穿越频率ωcr处相位裕度需要大于等于φmargin这一条件，得到一个包含Kp和Kr的不等式约束关系π+angle[Gio，d2(jωcr)·GQPR(jωcr)]≥φmargin；
根据上述得到的两个不等式约束关系和一个等式约束关系，即能够求解出比例系数Kp和谐振系数Kr的范围，并据此得到控制参数。
可选地，还包括：
在网侧电压过零点附近满足|vg|≤10V时，输入电流闭环补偿控制器的电流误差置0；
当电网电压vg小于-10V时，所述网侧电流误差为(ig-ig，ref)；
当电网电压vg大于10V时，所述网侧电流误差为(ig，ref-ig)。
由于采用了上述技术方案，本发明与现有技术相比，具有如下以下至少一项的有益效果：
本发明提供的双有源桥型微逆变器的电路建模方法，建立了完整的三阶微逆变器大信号和小信号模型，一方面为求解稳态工作点提供了便利，另一方面通过小信号模型可以进一步研究微逆变器的并网稳定性和动态性能，并为控制参数的设计提供指导。
本发明提供的双有源桥型微逆变器的输出电流控制方法，给出了正确的电流环控制逻辑和详尽的电流环参数设计方法，有利于改善微逆变器的并网稳定性和输出电流精度。
本发明提供的双有源桥型微逆变器的电路建模方法，可以实现对DAB型微逆变器的准确建模。
本发明提供的双有源桥型微逆变器的输出电流控制方法，可以较好地对DAB型微逆变器 进行电流控制。
附图说明
通过阅读参照以下附图对非限制性实施例所作的详细描述，本发明的其它特征、目的和优点将会变得更明显：
图1为本发明一优选实施例中单级式半桥DAB型微逆变器的电路示意图；
图2为本发明一优选实施例中当传输功率方向为从直流侧到交流侧且网侧电压为正时，三种调制模式下开关管S1～S8的驱动信号，以及变压器原边电压、变压器副边电压和变压器副边电流的波形示意图；
图3为本发明一优选实施例中三种调制模式对应的内移相角和外移相角范围示意图；
图4为本发明一优选实施例中双有源桥型微逆变器的等效电路示意图；
图5为本发明一优选实施例中双有源桥型微逆变器的整体控制框图。
具体实施方式
下面结合具体的实施例对本发明进行详细的说明。以下实施例将有助于本领域的技术人员进一步理解本发明，但不以任何形式限制本发明。应当指出的是，对本领域的普通技术人员来说，在不脱离本发明构思的前提下，还可以做出若干变形和改进，这都属于本发明的保护范围。以下没有说明的部分，可以参照发明内容中记载或现有技术。
本发明一实施例提供了一种双有源桥型微逆变器的电路建模方法，该方法通过建立双有源桥型微逆变器的等效电路模型，将微逆变器等效为标准的双有源桥型电路(DAB型电路)，从而进一步建立微逆变器的三阶大信号模型和小信号模型，完成对双有源桥型微逆变器的电路建模，以便于进行电路状态变量的求解，进而实现对微逆变器的并网稳定性和动态性能的分析。
在该实施例中，双有源桥型微逆变器，包括：光伏板、直流侧母线电容、原边方波发生电路、高频变压器、副边方波发生电路、滤波电容Cg、交流电网侧的滤波电感Lg以及串联阻尼电阻RL；光伏板的正负极分别与直流侧母线电容正负极相连，直流侧母线电容的正负极分别与原边方波发生电路的直流侧端口正负极相连，原边方波发生电路的交流侧端口正负极分别与高频变压器的输入端口正负极相连，副边方波发生电路的交流侧端口分别与高频变压器的输出端口正负极相连，副边方波发生电路的直流侧端口正极与交流电网侧的滤波电感Lg相 连，交流电网侧的滤波电感Lg与串联阻尼电阻RL串联，并与交流电网正极相连；副边方波发生电路的直流侧端口负极与交流电网负极相连；其中，高频变压器原副边匝比为1:n，折算到双有源桥型微逆变器变压器副边的漏感为Lk。
双有源桥型微逆变器变压器原边可以输出三个电平{vpv,0,-vpv}，副边可以输出两个电平{|vg|/2,-|vg|/2}。其中vpv为光伏板端电压，|vg|为网侧电压的绝对值。
双有源桥型微逆变器通过移相调制实现对输出功率的调节，微逆变器具有两个控制自由度，分别为原边电路的内移相角D1和原副边电路的外移相角D2，其中内移相角定义为原边方波电压负上升沿和原边方波电压正上升沿错开的角度；外移相角定义为变压器原边方波电压的基波和变压器副边方波电压的基波错开的角度。D1的取值范围是0≤D1≤0.5，D2的取值范围是-0.5≤D2≤0.5；
进一步的，根据微逆变器的两个控制自由度，双有源桥型微逆变器实现对输出功率的调节的移相调制模式共有三种调制模式，根据内移相角D1和外移相角D2的取值范围划分。当D2满足(1-D1)/2＜D2≤0.5或-0.5＜D2≤-(1-D1)/2时，原边电压的正电平部分和副边电压的负电平部分完全重合，对应调制模式为模式一，此时变压器电流接近正弦波，变压器电流有效值最大，但原副边开关管的零电压软开关最容易实现；当D2满足D1/2＜D2≤(1-D1)/2或-(1-D1)/2＜D2≤-D1/2时，原边电压的正电平一部分和副边电压的正电平重合，另一部分和副边电压的负电平重合，对应的调制模式为模式二，此时变压器电流接近梯形波，变压器电流有效值适中，即小于模式一时的有效值且大于模式三时的有效值，原副边开关管的零电压软开关较易实现，即容易程度小于模式一时的容易程度且大于模式三时的容易程度；当外移相角D2满足0≤D2≤D1/2或-D1/2≤D2≤0时，原边电压的正电平部分和副边电压的正电平部分完全重合，对应的调制模式为模式三，此时变压器电流接近三角波，变压器电流有效值最小，但原副边开关管的零电压软开关最难实现。
在一优选实施例中，建立双有源桥型微逆变器的等效电路模型，包括：构建直流电源Voc、直流源串联电阻Rpv、直流侧母线电容Cbus、受控电流源iac,in、原副边匝比为1:n的理想变压器、受控电流源iac,out、电容Cg、交流电网侧的滤波电感Lg、电感绕线电阻RL以及直流脉动电源|vg|；直流电源Voc的正极与直流源串联电阻Rpv的一端相连，直流源串联电阻Rpv的另一端与直流侧母线电容Cbus的正极相连，直流侧母线电容Cbus的正极同时与理想变压器原边的正极相连，理想变压器原边的负极与直流电源Voc的负极相连，理想变压器副边的正极与受控 电流源iac,in的负极相连，理想变压器副边的负极与受控电流源iac,in的正极相连，并同时与受控电流源iac,out的负极相连，受控电流源iac,out的正极与电容Cg的正极相连，并同时与交流电网侧的滤波电感Lg的一端相连，交流电网侧的电感Lg的另一端与电感绕线电阻RL的一端相连，电感绕线电阻RL的另一端与直流脉动电源|vg|的正极相连，直流脉动电源|vg|的负极与电容Cg的负极相连。
其中：
直流电源Voc的电压为光伏板开路电压；直流源串联电阻Rpv等于最大功率点处光伏板的输出电压除以输出电流；受控电流源iac,in和受控电流源iac,out的输出电流(即变压器电流)根据双有源桥型微逆变器的调制模式的不同而发生变化。
构建得到的等效电路的输出电压为|vg|，输出电流io等于ig·sgn(vg)，其中，ig为采样得到的网侧电流，sgn(vg)为网侧电压的符号函数。
在一优选实施例中，微逆变器的大信号模型为三阶平均值模型其中X为大信号模型的状态变量；为大信号模型状态变量对时间的导数；U为大信号模型的输入变量；A1为大信号模型的状态变量系数矩阵，阶数为3×3；B1为大信号模型的输入变量系数矩阵，阶数为3×2。
其中：
大信号模型的状态变量为X＝[<vbus>，<vcg>，<io>]T；大信号模型的输入变量为U＝[<vdc>，<vo>]T。其中<vbus>为等效电路的直流侧母线电容Cbus的电压在开关周期内的平均值，<vcg>为等效电路中电容Ceq的电压在开关周期内的平均值，<io>为等效电路的输出电流在开关周期内的平均值，<vdc>为等效电路的直流侧输入电压在开关周期内的平均值，<vo>为等效电路的输出电压在开关周期内的平均值。
微逆变器的大信号模型，用于求解所述状态变量X随所述输入变量U的变化关系，求解得到的所述变化关系用于分析电路的稳态工作点，在得到的所述稳态工作点处对所述小信号模型进行求解和分析。
在一优选实施例中，微逆变器的小信号模型为三阶模型x＝A2x+B2u，其中x为小信号模型的状态变量；为小信号模型状态变量的导数；u为小信号模型的输入变量；A2为小信号 模型的状态变量系数矩阵，阶数为3×3；B1为小信号模型的输入变量系数矩阵，阶数为3×4。
其中：
小信号模型的状态变量为小信号模型的输入变量为其中为等效电路的直流侧母线电容Cbus的电压的扰动值，为等效电路中电容Cg的电压的扰动值，为等效电路的输出电流的扰动值，为等效电路的直流侧输入电压的扰动值，为等效电路的输出电压的扰动值，为内移相角的扰动值，为外移相角的扰动值。
双有源桥型微逆变器的小信号模型可用于求解从四个输入变量到三个状态变量的传递函数，进一步用于分析微逆变器在不同电路参数和工况下的并网稳定性和动态性能。
所述四个输入变量为所述三个状态变量为所述传递函数的表达式为：
x＝(sI-A2)-1B2u
其中，I为对角线元素为1的对角矩阵，阶数为3×3；s为拉氏域符号；矩阵A2和B2的表达式为：

其中，Vbus为直流母线电容电压，Rpv为最大功率点处光伏板的输出电压除以输出电流，Lg为交流电网侧的滤波电感，符号变量α、β、γ与双有源桥型微逆变器的调制模式相关。
进一步地，对于模式二，各符号变量为：
进一步地，对于模式三，各符号变量为：
其中，fs为开关管开关频率，Lk为变压器漏感折算到副边的值，n为变压器副原边匝比，D1和D2分别为内外移相角。
所述传递函数可以用于描述不同频率下状态变量随输入变量的定量变化关系，当传递函数已知时，则可以针对各种不同形式的输入变量研究状态变量的响应，以便对微逆变器的并网稳定性进行分析。
所述进一步分析微逆变器在不同电路参数和工况下动态性能的方法是：改变光伏端电压vpv、电网电压幅值Vm和并网功率Po这三个参数，在每一组参数下根据微逆变器的小信号模型求解得到从输入变量到状态变量的传递函数，并根据传递函数的增益、穿越频率、幅值裕度和相位裕度分析微逆变器的动态性能。
本发明上述实施例提供的电路建模方法，通过建立等效电路模型，将双有源桥型微逆变器等效为标准的双有源桥电路，从而建立起微逆变器的三阶大信号模型和小信号模型，以便于进行电路状态变量的求解，进而实现对微逆变器的并网稳定性和动态性能的分析。该电路建模方法建立了完整的微逆变器大信号和小信号模型，一方面为求解稳态工作点提供了便利，另一方面利用小信号模型得到的传递函数为微逆变器动态性能的研究提供了便利。
本发明一实施例还提供了一种双有源桥型微逆变器的输出电流控制方法，基于上述实施例中任一项的电路建模方法建立的微逆变器的小信号模型，根据网侧电压正负确定电流环的控制逻辑，通过正确的控制逻辑得到的网侧电流误差输入到电流闭环补偿控制器，其输出为微逆变器的外移相角D2，可用于进行移相控制，完成对双有源桥型微逆变器的输出电流闭环控制。其中：
电流环的控制逻辑具体如下：
当电网电压vg小于0时，网侧电流误差为(ig-ig，ref)，其中ig为采样得到的网侧电流，ig,ref为网侧电流给定值；
当电网电压vg大于0时，网侧电流误差为(ig，ref-ig)。
进一步的，电流闭环补偿控制器为准比例谐振(Q-PR)控制器，具有三个可设计控制参数，分别为①带宽调节参数ωc、②比例系数Kp、③谐振系数Kr。
三个参数的作用如下：
①带宽调节参数ωc：用于改变控制器频带特性的谐振带宽，ωc越小则选频特性越好，但是抗电网频率扰动的能力越差；
②比例系数Kp：用于改善双有源桥型微逆变器控制系统的动态特性，Kp越大反应时间越短，但是超调量越大；
③谐振系数Kr：用于调节控制器的增益，Kr越大则控制器增益越大。
Q-PR控制器的控制参数具体设计原则如下：
步骤一：根据双有源桥型微逆变器的小信号模型，得到工频周期内3ms处从外移相角到输出电流的传递函数Gio，d2(s)；选择工频频率ωac处电流环的开环增益大于等于M；选择电流环开环传递函数的穿越频率为ωcr；选择电流环开环传递函数的相位裕度大于等于φmargin；
步骤二：根据电网角频率最大允许变化范围[-Δωmax，Δωmax]，取ωc＝Δωmax，其中Δωmax为电网角频率的最大允许变化量；
步骤三：根据Gio，d2(s)和Q-PR控制器的传递函数GQPR(s)得到工频频率ωac处的电流环开环传递函数Tc(jωac)＝Gio，d2(jωac)·GQPR(jωac)，令其增益大于等于M，得到一个包含Kp和Kr的不等式约束关系|Gio，d2(jωac)·GQPR(jωac)|≥M；
步骤四：根据电流环在穿越频率处增益为1的性质，得到一个包含Kp和Kr的等式约束关系|Gio，d2(jωcr)·GQPR(jωcr)|＝1；
步骤五：根据电流环在穿越频率处相位裕度大于等于φmargin这一条件，得到一个包含Kp和Kr的不等式约束关系π+angle[Gio，d2(jωcr)·GQPR(jωcr)]≥φmargin。
根据步骤三至步骤五得到的两个不等式约束和一个等式约束即可求解出比例系数Kp和 谐振系数Kr的范围，并据此选择合适的控制参数。
在一具体应用实例中，在网侧电压过零点附近满足|vg|≤10V时，输入电流闭环补偿控制器的电流误差置0，其余时间段电流误差的计算按照正常原则进行，具体地，当电网电压vg小于-10V时，网侧电流误差为(ig-ig，ref)；当电网电压vg大于10V时，网侧电流误差为(ig，ref-ig)。
本发明上述实施例提供的输出电流控制方法，是一种输出电流闭环控制方法，基于双有源桥型微逆变器的小信号模型，建立了正确的电流环控制逻辑，并给出了电流环参数的设计方法。该输出电流控制方法，通过给出正确的电流环控制逻辑和详尽的电流环参数设计方法，有利于改善微逆变器的并网稳定性和输出电流精度。
下面结合附图，对本发明上述实施例提供的电路建模方法及输出电流控制方法进一步说明。
图1为单级式半桥DAB型微逆变器电路示意图。参照图1所示，单级式半桥DAB型微逆变器电路由光伏板组件、直流母线电容、原边全桥电路、高频变压器、副边半桥电路、网侧低通滤波器组成。其中原边全桥电路包含开关管S1～S4，副边半桥电路包含开关管S5～S8和薄膜电容C1/C2；高频变压器原副边匝比为1:n，折算到原边的励磁电感为Lm，折算到副边的变压器漏感为Lk。
图2为本发明实施例一中单级式半桥DAB型微逆变器的三种调制模式的开关管S1～S8的驱动波形和变压器原副边电压电流波形。参照图2所示，单级式半桥DAB型微逆变器的基本工作方式为：开关管S1和S2高频互补导通，开关管S3和S4高频互补导通。当网侧电压为正时，开关管S6和S8常通，开关管S5和S7高频互补导通；当网侧电压为负时，开关管S5和S7常通，开关管S6和S8高频互补导通。
参照图2所示，单级式半桥DAB型微逆变器变压器原边可以输出三个电平{vpv,0,-vpv}，副边可以输出两个电平{|vg|/2,-|vg|/2}。其中vpv为光伏板端电压，|vg|为网侧电压的绝对值。
进一步的，单级式半桥DAB型微逆变器通过移相调制实现对输出功率的调节，微逆变器具有两个控制自由度，分别为原边电路的内移相角D1和原副边电路的外移相角D2，其中内移相角定义为开关管S4的驱动脉冲和开关管S1的驱动脉冲错开的角度；外移相角定义为变压器原边方波电压的基波和变压器副边方波电压的基波错开的角度。D1的取值范围是0≤D1≤0.5，D2的取值范围是-0.5≤D2≤0.5。
图3为本发明实施例一中三种调制模式对应的内移相角和外移相角范围示意图，参照图3所示，当D2满足(1-D1)/2＜D2≤0.5或-0.5＜D2≤-(1-D1)/2时，对应调制模式为模式一；当D2满足D1/2＜D2≤(1-D1)/2或-(1-D1)/2＜D2≤-D1/2时，对应的调制模式为模式二；当外移相角D2满足0≤D2≤D1/2或-D1/2≤D2≤0时，对应的调制模式为模式三。
图4为本发明实施例一中单级式半桥DAB型微逆变器的等效电路，参照图4所示，微逆变器的等效电路由直流电源Voc、直流源串联电阻Rpv、直流侧母线电容Cbus、受控电流源iac,in、原副边匝比为1:n的理想变压器、受控电流源iac,out、电容Ceq、交流电网侧的滤波电感Lg以及电感绕线电阻RL组成。
其中直流电源Voc的电压为光伏板开路电压；直流源串联电阻Rpv等于最大功率点处光伏板的输出电压除以输出电流；受控电流源iac,in和受控电流源iac,out的输出电流根据调制模式的不同而发生变化；电容Ceq为C1和C2的串联；等效电路的输出电压为|vg|，输出电流io等于ig·sgn(vg)。
根据图4所示的单级式半桥DAB型微逆变器的等效电路，可以给出电路的大信号模型。电路的大信号模型为三阶平均值模型其中X为大信号模型的状态变量；为大信号模型状态变量对时间的导数；U为大信号模型的输入变量；A1为大信号模型的状态变量系数矩阵，阶数为3×3；B1为大信号模型的输入变量系数矩阵，阶数为3×2。
大信号模型的状态变量为X＝[<vbus>，<vceq>，<io>]T；大信号模型的输入变量为U＝[<vdc>，<vo>]T。其中<vbus>为直流母线电容电压在开关周期内的平均值，<vceq>为等效电路中电容Ceq的电压在开关周期内的平均值，<io>为等效电路的输出电流在开关周期内的平均值，<vdc>为等效电路的直流侧输入电压在开关周期内的平均值，<vo>为等效电路的输出电压在开关周期内的平均值。
大信号模型的通式为：
其中α与微逆变器的调制模式相关，对于模式二来说：对于模式三而言：
根据图4所示的单级式半桥DAB型微逆变器的等效电路，可以给出电路的小信号模型。电路的小信号模型为三阶模型x＝A2x+B2u，其中x为小信号模型的状态变量；为小信号模型状态变量的导数；u为小信号模型的输入变量；A2为小信号模型的状态变量系数矩阵，阶数为3×3；B1为小信号模型的输入变量系数矩阵，阶数为3×4。
小信号模型的状态变量为小信号模型的输入变量为其中为直流母线电容电压的扰动值，为等效电路中电容Ceq的电压的扰动值，为等效电路的输出电流的扰动值，为等效电路的直流侧输入电压的扰动值，为等效电路的输出电压的扰动值，为内移相角的扰动值，为外移相角的扰动值。
小信号模型的通式为：
其中符号变量α、β、γ与调制模式相关，对于模式二，各符号变量为：
对于模式三，各符号变量为：
根据上述单级式半桥DAB型微逆变器的小信号模型可用于求解从四个输入变量到三个 状态变量的传递函数，具体求解方式为：
x＝(sI-A2)-1B2u
其中I为对角线元素为1的对角矩阵，阶数为3×3；s为拉氏域符号。
如图5所示为本发明实施例二中单级式半桥DAB型微逆变器的整体控制框图。参照图5所示，控制框图分为最大功率点追踪(MPPT)、电压环、锁相环、电流环、前馈控制和移相控制共六个部分。控制器采样光伏板的端电压vpv和输出电流ipv后，经过最大功率点追踪(MPPT)环节后生成光伏板端电压给定值Vpv，ref；Vpv，ref和实际采样电压vpv做差后经过电压环生成网侧电流给定值的幅值Im；控制器采样网侧电压vg并经过锁相环生成网侧电压相位θ；θ经过正弦变换后与Im相乘得到网侧电流给定值ig,ref，并与实际网侧电流采样值ig做差，经过一定的控制逻辑后通过电流闭环补偿控制器得到外移相角D2；内移相角D1经过移相控制器生成驱动信号S1～S4，外移相角D2经过移相控制器生成驱动信号S5～S8，使得单级式半桥DAB型微逆变器输出给定功率。
进一步的，电流环的控制逻辑具体如下：
当电网电压vg小于0时，网侧电流误差为(ig-ig，ref)，其中ig为采样的到的网侧电流，ig,ref为网侧电流给定值；
当电网电压vg大于0时，网侧电流误差为(ig，ref-ig)。
在网侧电压过零点附近，即满足|vg|≤10V时，输入电流闭环补偿控制器的电流误差为0。
进一步的，电流闭环补偿控制器为准比例谐振(Q-PR)控制器，具有三个可设计控制参数，分别为①带宽调节参数ωc、②比例系数Kp、③谐振系数Kr。
具体地，三个参数的作用如下：
①带宽调节参数ωc：用于改变控制器频带特性的谐振带宽，ωc越小则选频特性越好，但是抗电网频率扰动的能力越差；
②比例系数Kp：用于改善系统的动态特性，Kp越大反应时间越短，但是超调量越大；
③谐振系数Kr：用于调节控制器的增益，Kr越大则控制器增益越大。
具体地，Q-PR控制器的具体设计原则如下：
步骤一：根据单级式半桥DAB型微逆变器的小信号模型，得到工频周期内3ms处从外移相角到输出电流的传递函数Gio，d2(s)；选择ωac处电流环的开环增益大于等于M；选择电流 环开环传递函数的穿越频率为ωcr；选择电流环开环传递函数的相位裕度大于等于φmargin；
步骤二：根据电网角频率最大允许变化范围[-Δωmax，Δωmax]，取ωc＝Δωmax，其中Δωmax为电网角频率的最大允许变化量；
步骤三：根据Gio，d2(s)和Q-PR控制器的传递函数GQPR(s)得到ωac处的电流环开环传递函数Tc(jωac)＝Gio，d2(jωac)·GQPR(jωac)，令其增益大于等于M，得到一个包含Kp和Kr的不等式约束关系|Gio，d2(jωac)·GQPR(jωac)|≥M；
步骤四：根据电流环在穿越频率处增益为1的性质，得到一个包含Kp和Kr的等式约束关系|Gio，d2(jωcr)·GQPR(jωcr)|＝1；
步骤五：根据电流环在穿越频率处相位裕度大于等于φmargin的条件，得到一个包含Kp和Kr的不等式约束关系π+angle[Gio，d2(jωcr)·GQPR(jωcr)]≥φmargin。
根据步骤三至步骤五得到的两个不等式约束和一个等式约束即可求解出比例系数Kp和谐振系数Kr的范围，并据此选择合适的控制参数。
当然，以上实施例的具体电路仅仅是本发明一种实现的优选实施例，并不用于限定本发明，在其他实施例中，也可以是实现相同功能的其他电路形式。
本发明上述实施例中双有源桥型微逆变器的电路建模方法，建立了完整的三阶微逆变器大信号和小信号模型，一方面为求解稳态工作点提供了便利，另一方面通过小信号模型可以进一步研究微逆变器的动态性能，并为控制参数的设计提供指导。
本发明上述实施例中双有源桥型微逆变器的输出电流控制方法，给出了正确的电流环控制逻辑和详尽的电流环参数设计方法，有利于改善微逆变器的并网稳定性和输出电流精度。
本发明上述实施例中未尽事宜均为本领域公知技术。
以上对本发明的具体实施例进行了描述。需要理解的是，本发明并不局限于上述特定实施方式，本领域技术人员可以在权利要求的范围内做出各种变形或修改，这并不影响本发明的实质内容。

Claims (10)

  1. 一种双有源桥型微逆变器的电路建模方法，其特征在于，通过建立双有源桥型微逆变器的等效电路模型，将微逆变器等效为标准的双有源桥型电路，从而进一步建立微逆变器的大信号模型和小信号模型，完成对双有源桥型微逆变器的电路建模；其中：
    所述建立双有源桥型微逆变器的等效电路模型，包括：
    构建直流电源Voc、直流源串联电阻Rpv、直流侧母线电容Cbus、受控电流源iac,in、原副边匝比为1:n的理想变压器、受控电流源iac,out、电容Cg、交流电网侧的滤波电感Lg、电感绕线电阻RL以及直流脉动电源|vg|；所述直流电源Voc的正极与所述直流源串联电阻Rpv的一端相连，所述直流源串联电阻Rpv的另一端与所述直流侧母线电容Cbus的正极相连，所述直流侧母线电容Cbus的正极同时与所述理想变压器原边的正极相连，所述理想变压器原边的负极与所述直流电源Voc的负极相连，所述理想变压器副边的正极与所述受控电流源iac,in的负极相连，所述理想变压器副边的负极与所述受控电流源iac,in的正极相连，并同时与所述受控电流源iac,out的负极相连，所述受控电流源iac,out的正极与所述电容Cg的正极相连，并同时与所述交流电网侧的滤波电感Lg的一端相连，所述交流电网侧的电感Lg的另一端与所述电感绕线电阻RL的一端相连，所述电感绕线电阻RL的另一端与所述直流脉动电源|vg|的正极相连，所述直流脉动电源|vg|的负极与所述电容Cg的负极相连；
    所述建立微逆变器的大信号模型为：三阶平均值模型其中，X为大信号模型的状态变量；为大信号模型状态变量对时间的导数；U为大信号模型的输入变量；A1为大信号模型的状态变量系数矩阵；B1为大信号模型的输入变量系数矩阵；所述大信号模型的状态变量X为：X＝[<vbus>，<vcg>，<io>]T；所述大信号模型的输入变量U为：U＝[<vdc>，<vo>]T；其中，<vbus>为等效电路的直流侧母线电容Cbus的电压在开关周期内的平均值，<vcg>为等效电路中电容Cg的电压在开关周期内的平均值，<io>为等效电路的输出电流在开关周期内的平均值，<vdc>为等效电路的直流侧输入电压在开关周期内的平均值，<vo>为等效电路的输出电压在开关周期内的平均值；
    所述建立微逆变器的小信号模型为：三阶模型其中，x为小信号模型的 状态变量；为小信号模型状态变量的导数；u为小信号模型的输入变量；A2为小信号模型的状态变量系数矩阵；B1为小信号模型的输入变量系数矩阵；所述小信号模型的状态变量x为：所述小信号模型的输入变量u为：其中，为等效电路的直流侧母线电容Cbus的电压的扰动值，为等效电路中电容Cg的电压的扰动值，为等效电路的输出电流的扰动值，为等效电路的直流侧输入电压的扰动值，为等效电路的输出电压的扰动值，为内移相角的扰动值，为外移相角的扰动值。
  2. 根据权利要求1所述的双有源桥型微逆变器的电路建模方法，其特征在于，所述双有源桥型微逆变器的等效电路模型中，所述直流电源Voc的电压为光伏板开路电压；所述直流源串联电阻Rpv等于最大功率点处光伏板的输出电压除以输出电流；所述受控电流源iac,in和所述受控电流源iac,out的输出电流根据所述双有源桥型微逆变器的调制模式的不同而发生变化；
    所述直流脉动电源|vg|为所述等效电路模型的输出电压，所述等效电路模型的输出电流io等于ig·sgn(vg)，其中，ig为采样得到的网侧电流，sgn(vg)为网侧电压的符号函数。
  3. 根据权利要求1所述的双有源桥型微逆变器的电路建模方法，其特征在于，所述微逆变器的大信号模型，用于求解所述状态变量X随所述输入变量U的变化关系，求解得到的所述变化关系用于分析电路的稳态工作点，在得到的所述稳态工作点处对所述小信号模型进行求解和分析。
  4. 根据权利要求1所述的双有源桥型微逆变器的电路建模方法，其特征在于，所述微逆变器的小信号模型，用于求解从四个输入变量到三个状态变量的传递函数，进一步用于分析微逆变器在不同电路参数和工况下的并网稳定性和动态性能；其中：
    所述四个输入变量为所述三个状态变量为所述传递函数的表达式为：
    x＝(xI-A2)-1B2u
    其中，I为对角线元素为1的对角矩阵；s为拉氏域符号；矩阵A2和B2的表达式为：

    其中，Vbus为直流母线电容电压，Rpv为最大功率点处光伏板的输出电压除以输出电流，Lg为交流电网侧的滤波电感，符号变量α、β、γ与所述双有源桥型微逆变器的调制模式相关。
  5. 根据权利要求2或4所述的双有源桥型微逆变器的电路建模方法，其特征在于，所述双有源桥型微逆变器的调制模式，包括：
    将变压器原边方波电压负上升沿和变压器原边方波电压正上升沿错开的角度定位为内移相角D1，将变压器原边方波电压的基波和变压器副边方波电压的基波错开的角度定义为外移相角D2，将所述内移相角D1和所述外移相角D2作为所述双有源桥型微逆变器的两个控制自由度，其中，D1的取值范围为0≤D1≤0.5，D2的取值范围为-0.5≤D2≤0.5；
    根据两个控制自由度，将传输功率的调制模式划分为模式一、模式二和模式三，其中：当所述外移相角D2满足(1-D1)/2＜D2≤0.5或-0.5＜D2≤-(1-D1)/2时，原边方波电压的正电平部分和副边方波电压的负电平部分完全重合，对应调制模式为模式一，此时所述受控电流源iac,in和所述受控电流源iac,out的输出电流接近正弦波，输出电流有效值最大；当所述外移相角D2满足0≤D2≤D1/2或-D1/2≤D2≤0时，原边方波电压的正电平部分和副边方波电压的正电平部分完全重合，对应的调制模式为模式三，此时所述受控电流源iac,in和所述受控电流源iac,out的输出电流接近三角波，输出电流有效值最小；当所述外移相角D2满足D1/2＜D2≤(1-D1)/2或-(1-D1)/2＜D2≤-D1/2时，原边方波电压的正电平的一部分和副边方波电压的正电平重合，原边方波电压的正电平的另一部分和副边方波电压的负电平重合，对应的调制模式为模式二，此时所述受控电流源iac,in和所述受控电流源iac,out的输出电流接近梯形波，输出电流有效值小于模式一时的有效值且大于模式三时的有效值。
  6. 根据权利要求5所述的双有源桥型微逆变器的电路建模方法，其特征在于，对于所述模式二，符号变量α、β、γ分别为：
    对于所述模式三，所述符号变量α、β、γ分别为：
    其中，fs为开关管开关频率，Lk为变压器漏感折算到副边的值，n为变压器副原边匝比，D1和D2分别为内外移相角；符号变量α、β、γ与所述双有源桥型微逆变器的调制模式相关。
  7. 一种双有源桥型微逆变器的输出电流控制方法，其特征在于，基于权利要求1-6中任一项所述的电路建模方法建立的微逆变器的小信号模型，根据网侧电压正负确定电流环的控制逻辑，通过确定的控制逻辑得到网侧电流误差并输入到电流闭环补偿控制器，所述电流闭环补偿控制器的输出为微逆变器的外移相角D2，通过所述外移相角D2进行移相控制，完成对双有源桥型微逆变器的输出电流闭环控制；其中：
    所述电流环的控制逻辑包括：
    当电网电压vg小于0时，所述网侧电流误差为(ig-ig，ref)，其中，ig为采样的到的网侧电流，ig,ref为网侧电流给定值；
    当电网电压vg大于0时，所述网侧电流误差为(ig，ref-ig)。
  8. 根据权利要求7所述的双有源桥型微逆变器的输出电流控制方法，其特征在于，所述电流闭环补偿控制器采用准比例谐振控制器，所述准比例谐振控制器具有三个控制参数，分别为：带宽调节参数ωc、比例系数Kp和谐振系数Kr；其中：
    所述带宽调节参数ωc：用于改变所述电流闭环补偿控制器频带特性的谐振带宽，ωc越小则选频特性越好，但是抗电网频率扰动的能力越差；
    所述比例系数Kp：用于改善双有源桥型微逆变器控制系统的动态特性，Kp越大反应时间越短，但是超调量越大；
    所述谐振系数Kr：用于调节所述电流闭环补偿控制器的增益，Kr越大则控制器增益越大。
  9. 根据权利要求8所述的双有源桥型微逆变器的输出电流控制方法，其特征在于，所述准比例谐振控制器的控制参数的设计方法，包括：
    根据双有源桥型微逆变器的小信号模型，得到工频周期内设定时间处从外移相角D2到输出电流的电流环开环传递函数Gio，d2(s)；选择工频频率ωac处电流环的开环增益大于等于M；选择电流环开环传递函数的穿越频率为ωcr；选择电流环开环传递函数的相位裕度大于等于φmargin；
    根据电网角频率最大允许变化范围[-Δωmax，Δωmax]，取ωc＝Δωmax，其中Δωmax为电网角频率的最大允许变化量；
    根据电流环开环传递函数Gio，d2(s)和准比例谐振控制器的传递函数GQPR(s)得到工频频率ωac处的电流环开环传递函数Tc(jωac)＝Gio，d2(jωac)·GQPR(jωac)，其中j为虚数单位，令工频频率ωac处的电流环开环传递函数Tc(jωac)的增益大于等于M，得到一个包含比例系数Kp和谐振系数Kr的不等式约束关系|Gio，d2(jωac)·GQPR(jωac)|≥M；
    根据电流环在穿越频率ωcr处增益为1的性质，得到一个包含比例系数Kp和谐振系数Kr的等式约束关系|Gio，d2(jωcr)·GQPR(jωcr)|＝1；
    根据电流环在穿越频率ωcr处相位裕度需要大于等于φmargin这一条件，得到一个包含Kp和Kr的不等式约束关系π+angle[Gio，d2(jωcr)·GQPR(jωcr)]≥φmargin；
    根据上述得到的两个不等式约束关系和一个等式约束关系，即能够求解出比例系数Kp和谐振系数Kr的范围，并据此得到控制参数。
  10. 根据权利要求7所述的双有源桥型微逆变器的输出电流控制方法，其特征在于，还包括：
    在网侧电压过零点附近满足|vg|≤10V时，输入电流闭环补偿控制器的电流误差置0；
    当电网电压vg小于-10V时，所述网侧电流误差为(ig-ig，ref)；
    当电网电压vg大于10V时，所述网侧电流误差为(ig，ref-ig)。
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