WO2013010385A1 - 一种自激推挽式变换器 - Google Patents

一种自激推挽式变换器 Download PDF

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Publication number
WO2013010385A1
WO2013010385A1 PCT/CN2012/070254 CN2012070254W WO2013010385A1 WO 2013010385 A1 WO2013010385 A1 WO 2013010385A1 CN 2012070254 W CN2012070254 W CN 2012070254W WO 2013010385 A1 WO2013010385 A1 WO 2013010385A1
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Prior art keywords
push
transistor
constant current
pull
circuit
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PCT/CN2012/070254
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English (en)
French (fr)
Inventor
王保均
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广州金升阳科技有限公司
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Priority to KR1020147000904A priority Critical patent/KR20140027463A/ko
Priority to JP2014520498A priority patent/JP2014521302A/ja
Priority to US14/128,628 priority patent/US9705421B2/en
Publication of WO2013010385A1 publication Critical patent/WO2013010385A1/zh

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/538—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a push-pull configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5383—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a self-oscillating arrangement
    • H02M7/53832—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a self-oscillating arrangement in a push-pull arrangement
    • H02M7/53835—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a self-oscillating arrangement in a push-pull arrangement of the parallel type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/337—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration
    • H02M3/3372—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration of the parallel type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/338—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement
    • H02M3/3382—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement in a push-pull circuit arrangement
    • H02M3/3384—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement in a push-pull circuit arrangement of the parallel type
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
    • H05B45/30—Driver circuits
    • H05B45/37—Converter circuits
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B20/00—Energy efficient lighting technologies, e.g. halogen lamps or gas discharge lamps
    • Y02B20/30—Semiconductor lamps, e.g. solid state lamps [SSL] light emitting diodes [LED] or organic LED [OLED]
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • This invention relates to DC-DC or DC-AC converters, and more particularly to the industrial control and lighting industry. Background technique
  • the existing self-excited push-pull converter the circuit structure is derived from the self-excited oscillation push-pull transistor single-transform DC converter invented by GH Royer in 1955, which is also the beginning of realizing the high-frequency switching control circuit;
  • the self-excited push-pull dual transformer circuit invented by Jen Sen (somewhere translated as "Jingsen") in 1957 is called the self-oscillating Jensen circuit; these two circuits are collectively called self-excited pushes. Pull converter.
  • Self-excited push-pull converters are described in Electronic Engineering Press, Principles and Designs of Switching Power Supplies, pages 67 to 70, ISBN ISBN 7-121-00211-6.
  • the main form of the circuit is the well-known Royer circuit and the self-oscillating Jensen circuit.
  • Figure 1-1 shows the common application of the self-excitation push-pull converter.
  • the circuit structure is a Royer circuit.
  • the capacitor C1 in parallel with the bias resistor R1 can be omitted in many cases.
  • the ZL patent number is 03273278. 3, date of publication: On August 25, 2004, the name is "Self-excited push-pull converter".
  • a Royer circuit with soft-start function is provided. See Figure 2, which solves the problem that the capacitor C1 in Figure 1_1 is used for push-pull at startup. The impact of the switching transistor.
  • Figure 1-2 is also an application.
  • the circuit structure is still a Royer circuit.
  • the original bias resistor R1 is split into two Rlu and Rid series, which are used for higher operating voltage input.
  • the capacitor C1 in parallel with the bias resistor Rlu can be omitted in many cases, so the capacitor C1 in Figure 1-2 is drawn in dashed lines.
  • Figure 3 is also a common Royer circuit, which simplifies the winding of the feedback winding.
  • the DC signal loop, the operating points of the transistors TR1 and TR2 are the same, but when the circuit is in the self-oscillating state, the transistors TR1 and TR2 work. difference.
  • the resistors Rla and Rlb are a cost-saving solution based on Figure 4.
  • the publication date is (2006-11-09) ((Switching power supply apparatus)), a resistance bias method similar to that of Fig. 4 is used as the unit circuit.
  • Figure 5 is a common Royer circuit. Since the inductor L1 is connected in series in the power supply circuit, and a capacitor CL is connected in parallel between the collector of the push-pull switch transistor, the circuit output is close to a sine wave, which is common in circuits such as energy-saving lamp electronic rectifiers, and the feedback winding can also be simplified.
  • the winding method is similar to that of Figures 3 and 4.
  • f is the oscillation frequency
  • BW is the working magnetic induction ( ⁇ )
  • ⁇ is the number of turns of the coil
  • S is the effective cross-sectional area of the core.
  • the circuit structure of Figure 1-1 is:
  • the input filter capacitor c is connected between the voltage input terminal and the ground to filter the input voltage;
  • the filtered input voltage is connected to the startup circuit, and the startup circuit is biased by the resistor R1 and the electric Capacitor CI is composed in parallel.
  • C1 can be omitted when the higher supply voltage is input.
  • the two ends of the bias resistor R1 are respectively coupled with the voltage input terminal and the coupling transformer B which provides positive feedback for the bases of the two push-pull transistors TR1 TR2.
  • the center taps of the side coils NB1 and NB2 are connected; the emitters of the two push-pull transistors TR1 TR2 are common, and the two collectors are respectively connected to the two ends of the primary windings NP1 and NP2 of the coupling transformer, and the base is connected to the primary side of the coupling transformer
  • the two ends of the coils NB1 and NB2, the center taps of the primary windings NP1 and NP2 are connected to the voltage input terminal; the secondary side coil NS of the coupling transformer B is connected to the output filter circuit to the voltage output
  • the Royer circuit uses the core saturation characteristic to perform push-pull oscillation.
  • the bias resistor R1 and the capacitor C1 are connected in parallel through the windings NB1 and NB2 to the transistors TR1 and TR2.
  • the base and emitter provide forward bias, and the two transistors TR1 and TR2 start to conduct. Since the characteristics of the two transistors are not exactly the same, one of the transistors will be turned on first, assuming that the transistor TR2 is turned on first, generating a set.
  • the electrode current I e2 the voltage of the corresponding winding of the coil NP2 is up and down.
  • the winding of the base coil NB2 also has a positive and negative induced voltage, which increases the base of the transistor TR2.
  • Current which is a positive feedback process, so that the transistor TR2 is quickly turned on; accordingly, the voltage of the winding of the coil NB1 corresponding to the transistor TR1 is up and down, and this voltage reduces the base current of the transistor TR1.
  • Transistor TR1 is quickly cut off completely.
  • the collector current of the switch tube increases sharply, the rate of increase is much larger than the increase of the base current, the transistor of the transistor TR2 is out of saturation, and the voltage drop UCE of the collector to the emitter of the transistor of the transistor TR2 increases, correspondingly, the transformer N P2
  • the voltage on the winding is reduced by the same value, and the voltage induced by the winding of the coil NB2 is reduced.
  • the base voltage of the transistor of the transistor TR2 is also lowered, causing the transistor of the transistor TR2 to change in the cut-off direction.
  • the voltage on the transformer coil will be In the reverse direction, the other transistor TR1 is turned on, and thereafter, this process is repeated to form a push-pull oscillation.
  • the waveform of the output of the winding Ns is as shown in FIG. 6.
  • the characteristics are as follows:
  • the core-saturation characteristic is used for push-pull oscillation, and the output waveform of the coupling transformer is approximate square wave, and the conversion efficiency of the circuit is high.
  • the circuit of Figure 5 is close to the sine wave because the inductor L1 is connected in series in the power supply loop and a capacitor CL is connected in parallel between the collector of the push-pull switch transistor.
  • B2 exhibits magnetic saturation, because B2 is small in size, the energy consumed by magnetic saturation is small, and the overall efficiency of the circuit is high.
  • the self-oscillation frequency of the Jensen circuit is relatively stable when the operating voltage, load, and temperature change.
  • the working voltage is poorly adaptable.
  • Table 1 shows the measured parameters of the Royer circuit. If the circuit of Figure 1_1 is used, it is a converter with input DC 5V, output DC 5V, and output current 200mA, that is, output power 1W.
  • the main parameters of the circuit are: Capacitor C is a luF capacitor, resistor R1 is 1 ⁇ ⁇ , capacitor C1 is 0. 047uF capacitor, transistors TR1 and TR2 are switching diodes with a magnification of about 200 times, and the collector maximum operating current is 1A.
  • the subsequent output of the transformer employs the circuit configuration of Fig. 8, and Fig. 8 shows a known full-wave rectification circuit.
  • the circuit adopts the circuit of Fig. 7 to make a converter with input DC 5V, output DC 5V, and output current 200mA, that is, output power 1W.
  • the main parameters of the circuit are: Capacitor C is luF capacitor, resistor R1 is 1 ⁇ ⁇ , capacitor Cla is 0. 047uF capacitor, transistor TR1 and TR2 are switching diodes with magnification of about 200 times, and the collector maximum working current is 1A.
  • the subsequent output of the transformer uses the circuit structure of Figure 8.
  • the conversion efficiency of the circuit is:
  • Vin is the operating voltage, ie the input voltage, I in is the input current; Vout is the output voltage, lout is the output current.
  • the circuit operating at 5V if it is operated at 8V, its own loss has reached 280mW. In the micropower DC/DC converter, this is barely acceptable, but at 12V. Under the working voltage, its own loss has reached 828mW, and at 20V, its own loss has reached 3600mW, that is, 3. 6W, the circuit working time is more than 3 seconds, the circuit will be damaged. Therefore, the conversion efficiency of the circuit also decreases as the operating voltage rises.
  • the Jensen circuit has the same problem. That is, the operating voltage is increased, causing the no-load operating current of the conventional self-excited push-pull converter to rise too fast, the no-load loss to rise too fast, and the conversion efficiency to decrease.
  • the technical problem to be solved by the present invention is: when the operating current of the self-excited push-pull converter rises slowly or does not rise as the operating voltage rises, and a surge occurs in the input voltage of the self-excitation push-pull converter When the self-excited push-pull converter has a certain anti-surge capability, it is not easy to damage the switching transistor.
  • the present invention provides a self-excitation push-pull converter, characterized in that a DC current loop of a base of a push-pull switch transistor is a constant current source between the active power supply terminals; that is, in a Royer or Jensen circuit. On the basis of this, cancel the bias resistance between the DC loop of the base of the push-pull switch transistor to the effective power supply terminal, and increase the constant current source instead of the original bias resistor.
  • the current direction of the constant current source should be consistent with the direction of the canceled original bias resistor current, that is, the current direction of the constant current source is the base level flowing into the NPN transistor; or the current direction of the constant current source is from the PNP transistor. The base level flows into the constant current source.
  • the constant current source can obtain a larger constant current value by parallel connection.
  • the constant current source may be any type of semiconductor device or an electronic circuit that implements a constant current.
  • the working principle of the invention is that the constant current source replaces the original bias resistor, but the current direction is consistent.
  • the two The current supplied by the base of the push-pull transistor is constant.
  • the circuit operates in a new manner to achieve push-pull oscillation, which is briefly described as follows:
  • the constant current source provides forward bias to the base and emitter of the transistor 1 and the transistor 2 through the feedback winding 1 and the feedback winding 2, and the transistor 1 and the transistor 2 start to conduct, because the characteristics of the two transistors are not It may be exactly the same, therefore, one of the transistors will be turned on first, assuming that the transistor 1 is turned on first, generating a collector current, the voltage of the corresponding coil winding 1 is positive for the power supply terminal, and the terminal connected to the collector of the triode 1 is negative, according to the same name
  • the base feedback winding 1 also exhibits a positive feedback induced voltage, which maintains and increases the base current of the transistor 1, which is a positive feedback process, so that the transistor 1 is quickly saturated and turned on; Ground, the induced voltage of the base feedback winding 2 corresponding to the transistor 2 reduces the base current of the transistor 2, and the transistor 2 is quickly turned off completely.
  • the base current reduction portion of the transistor 2 is all the base current increasing portion of the transistor 1.
  • the current of the coil winding 1 corresponding to the transistor 1 and the magnetic induction intensity generated by this current increase linearly with time.
  • the collector current of the transistor 1 approaches or reaches its base current.
  • transistor 1 will exit the saturation region and enter the amplification region.
  • the collector-to-emitter voltage drop UCE of the transistor 1 is significantly increased, correspondingly, three
  • the voltage across the coil winding 1 corresponding to the pole tube 1 is reduced by the same value, and the base feedback winding 1 also exhibits a corresponding induced voltage, which is also reduced.
  • This voltage weakens the base current of the transistor 1, which is a
  • the process of positive feedback quickly turns the transistor 1 out of the amplified state and enters the off state; accordingly, the induced voltage of the base feedback winding 2 corresponding to the transistor 2 increases the base current of the transistor 2, and the transistor 2 is completely completed. Saturated conduction.
  • the base current increase portion of the transistor 2 is entirely derived from the base current reduction portion of the transistor 1.
  • the two transistors are alternately turned on to complete the push-pull oscillation mode. Since the total input current of the base is limited by the constant current source and does not change with the fluctuation of the operating voltage, the circuit enters a new push-pull oscillation mode when the operating voltage rises.
  • the circuit can still operate with push-pull oscillation using the core saturation characteristic under conditions of proper load, proper operating voltage, and the like.
  • the circuit Since the bias resistor of the prior art can provide a larger base current after the operating voltage is increased, the circuit uses the core saturation characteristic to perform push-pull oscillation. At this time, the collector current is too large, and the triode is easily burned. After the constant current source is used, the circuit enters a new push-pull oscillation mode.
  • the maximum current of the collector of the transistor is limited by the base current. The maximum value of the current is related to the product of the constant current source output current and the number of amplifiers of the transistor. Thereby the push-pull transistor works in the safe zone.
  • the Jensen circuit works like this.
  • the synchronous rectification circuit is added at the output end, and the efficiency of the synchronous rectification is high, the voltage drop loss during rectification is small, and the working efficiency of the circuit can be improved; and the output can be realized in a wide input voltage range.
  • the voltage and input voltage are linearly synchronized.
  • the invention has the advantages that after the constant current source is used as the bias, the operating voltage is increased, and the no-load power consumption of the circuit is reduced under the same conditions, and the conversion efficiency of the circuit is significantly improved compared with the prior art.
  • the Royer circuit of Figure 1-1 is used to make a converter with an input DC of 5V, an output of 5V DC, and an output current of 200mA, that is, an output power of 1W.
  • the output of the transformer uses the circuit structure of Figure 8.
  • the main parameters of the circuit are: Capacitor C is a luF capacitor, resistor R1 is replaced by a 4. 3mA constant current source, capacitor C1 is 0. 047uF capacitor, transistors TR1 and TR2 are switching diodes with a magnification of about 200 times, and the collector is the largest.
  • the operating current is 1A.
  • the measured parameters are as follows:
  • the self-excited push-pull Royer converter of the present invention works at a higher level than the existing self-excited push-pull Royer converter.
  • the no-load input current, no-load loss, and conversion efficiency are significantly improved: 1.
  • the no-load current of the circuit using the invention is significantly reduced.
  • the prior art is 180 mA, and this current rises sharply to more than 300 mA after a few seconds as the working time is extended.
  • the present invention is 30 mA, and the current value is very stable, and the long-term operation does not rise. See Figure 9 for a comparison.
  • the conversion efficiency of the present invention is improved under the same operating voltage. 0% ⁇ The invention is 58. 6%, the invention is 86.0%. See Figure 11 for a comparison.
  • the present invention is applied to the circuit of Fig. 7 to make a converter with an input DC of 5V, an output of 5V DC, and an output current of 200mA, that is, an output power of 1W.
  • the main parameters of the circuit are: Capacitor C is a luF capacitor, resistor R1 is replaced by a 4. 3mA constant current source, and capacitor Cla is 0. 047uF capacitor, triode TR1 and TR2 are switching diodes with a magnification of about 200 times, and the collector is the largest.
  • the operating current is 1A.
  • the subsequent output of the transformer uses the circuit structure of Figure 8.
  • the measured parameters are as follows:
  • the self-excited push-pull Jensen converter has significantly improved no-load input current, no-load loss, and conversion efficiency after the operating voltage is increased:
  • the no-load current of the circuit using the invention is significantly reduced.
  • the prior art is 200mA, and this current rises sharply to more than 300mA after a few seconds as the working time is extended.
  • the present invention is 37 mA, and the current value is very stable, and the long-term operation does not rise. See Figure 12 for a comparison.
  • the conversion efficiency of the present invention is improved under the same operating voltage. For example, at 15 V, the prior art is 22.6%, and the present invention is 64.6%. See Figure 14 for a comparison.
  • the invention also has the advantages that the self-excitation push-pull Royer converter uses the constant current source as the bias, and as the operating voltage increases, the circuit is short-circuited at the output end under the same conditions, that is, the DC OUT of FIG. 8 is short-circuited.
  • the short circuit protection performance of the present invention is significantly improved.
  • the measured comparison data is shown in Table 5 below:
  • the use of the 4. 3 mA constant current source bias short circuit protection performance is significantly improved in connection with the special new push-pull oscillation mode brought by the present invention.
  • the existing Royer self-excitation push-pull converter has a short circuit protection mechanism.
  • the leakage inductance of the transformer is that the magnetic lines generated by the primary coil cannot pass through the secondary coil. Therefore, the inductance that causes leakage is called leakage inductance.
  • the secondary coil is usually used for output. When the secondary coil is directly short-circuited, the measured primary coil still has an inductance, which is usually considered to be a leakage inductance.
  • the circuit of the invention enters a new push-pull oscillation mode.
  • the circuit enters the high-frequency self-excitation push-pull oscillation, and does not rely on the collector current of the switch transistor of the magnetic core to achieve the push-pull state flipping.
  • the base current is limited, the collector current cannot rise, causing the switching transistor to enter the amplification region, and the push-pull state of the circuit is reversed.
  • Fig. 16 is an equivalent circuit schematic diagram of all known actual inductors.
  • the resistance of R0 is small, and the capacity of CO is also small.
  • the circuit of Figure 16 is a In the standard LC loop, the oscillating energy resonates in the loop.
  • the waveform of the oscillation is close to a sine wave. Due to the high frequency, the transmission efficiency of the transformer is low.
  • the approximate sine wave generated by the oscillation has its peak limited by the subsequent output short circuit.
  • the approximate sinusoidal energy generated by the oscillation resonates in the primary of the transformer, so the energy consumed is small, which is reflected at the input, that is, the total operating current decreases.
  • Figure 17 shows the approximate sinusoidal high-frequency oscillation measured by adding 2 turns in the transformer as a detection port when a short circuit occurs.
  • the top is not smooth and has an attenuated damped oscillation. This is the peak output of the subsequent output short circuit.
  • the circuit of the invention has a total operating current of 15 mA and a total power consumption of 180 mW when the operating voltage is short-circuited at a load of 12 V. See Table 5, and the total operating current of the prior art is higher than 120 mA, and the power consumption is as high as 1440 mW.
  • the present invention has no advantage at low voltages, the total power consumption does not exceed the power consumption of any of the switching transistors due to the low operating voltage, and the circuit is not damaged.
  • the present invention can achieve a better short circuit protection performance for the self-excited push-pull Royer converter.
  • the invention also has the advantage that when the output adopts a synchronous rectification circuit, linear synchronization of the output voltage and the input voltage can be realized over a wide input voltage range.
  • the output circuit of FIG. 8 is used, and the synchronous rectification circuit is used instead of the diode D21 and the diode D22.
  • the output voltage and the input voltage of the present invention are almost equal, and the same drawing is used.
  • the 1:1 Royer circuit, the measured comparison data is shown in Table 5 below:
  • pole tube D22 the number of turns of the output transformer output winding is reduced, so at 5V input, it is difficult to find a suitable number of turns so that the output is 5. 00V, and finally only find a relatively close 4.97V output voltage.
  • the current Royer circuit is operating at 20V, and the loss is up to 3600mW. See Table 1 for data. Even with synchronous rectification, the output voltage is linearly synchronized and cannot operate at 20V input voltage. However, since the present invention has a no-load current of 30 mA and a loss of 600 mW at 20 V, see Table 3, and it is still working normally. The invention solves the problem that the working voltage has poor adaptability, and the linear synchronization of the output voltage becomes a practical technology.
  • the MY65 4-digit digital multimeter has an internal resistance of 10 ⁇ when measuring voltage and an internal resistance of 1 ⁇ for 200 mA current. When the current exceeds 200mA, two ammeters are used to measure in parallel with 200mA, and the current readings of the two meters are added, which is the measured value. Parallel measurement of ammeter is a mature technology of existing electronic engineering.
  • the VI voltage meter head is the working voltage Vin, that is, the input voltage; the A1 current meter head is the input current Iin, which is the operating current; the V2 voltage meter head is the output voltage Vout, and the A2 current meter head is the output current lout; then the conversion efficiency can be formulated ( 2) Calculated.
  • DRAWINGS the working voltage Vin, that is, the input voltage
  • the A1 current meter head is the input current Iin, which is the operating current
  • the V2 voltage meter head is the output voltage Vout
  • the A2 current meter head is the output current lout
  • Figure 1-1 shows the schematic diagram of the common application circuit of Royer in the self-excitation push-pull converter
  • Figure 1-2 is a schematic diagram of another application circuit common to Royer in a self-excitation push-pull converter;
  • Figure 2 is a patented Royer circuit with a soft start function;
  • Z is a patent number: 03273278.
  • Figure 4 is a prototype of the circuit of Figure 3, a Royer application circuit that simplifies the feedback winding;
  • Figure 5 shows a Royer application circuit that outputs a nearly sinusoidal wave
  • Figure 6 is an output waveform diagram of the Ns winding in the circuit of Figure 1-1;
  • Figure 7 is a common application principle diagram of the well-known Jensen circuit in the self-excitation push-pull converter;
  • Figure 8 is a well-known full-wave rectifier circuit;
  • Figure 9 is a comparison diagram of no-load input current for different bias methods of 5V to 5V Royer circuits
  • Figure 10 is a comparison of no-load loss for different bias methods for 5V to 5V Royer circuits
  • Figure 11 is a comparison of conversion efficiency of different bias methods for 5V to 5V Royer circuits
  • Figure 12 is a comparison diagram of input currents for different bias methods for 5V to 5V Jensen circuits
  • Figure 13 is a comparison of the no-load losses of the different bias methods for the 5V to 5V Jensen circuit
  • Figure 14 is a comparison of conversion efficiency of different bias methods for 5V to 5V Jensen circuits
  • Figure 15 shows the different biasing methods for the 5V to 5V Royer circuit. When the load is short-circuited, the total input current of the circuit is compared.
  • Figure 16 is a schematic diagram of a practical equivalent circuit of a known inductor
  • Figure 17 is a high-frequency oscillation waveform detected in the transformer when the circuit is short-circuited according to the present invention
  • Figure 18 is a test schematic diagram commonly used in the present invention
  • Figure 19 is a circuit diagram of a first embodiment of the present invention.
  • Figure 20 is a circuit diagram of a second embodiment of the present invention.
  • Figure 21 is a circuit diagram of a third embodiment of the present invention.
  • Figure 22 is a circuit diagram of a fourth embodiment of the present invention.
  • Figure 23 shows the symbol of the constant current source in the circuit diagram
  • Figure 24-1 is a schematic diagram of a constant current source forming a constant current source
  • Figure 24-2 is a schematic diagram of a constant current source for a junction field effect transistor
  • Figure 24-3 is a schematic diagram of another type of junction field effect transistor forming a constant current source
  • Figure 24-4 is a schematic diagram of a bipolar PNP tube forming a constant current source
  • Figure 24-5 is a schematic diagram of another bipolar PNP tube forming a constant current source
  • Figure 24-6 is a schematic diagram of a constant current source using a TL431 precision adjustable reference integrated circuit
  • Figure 24-7 is a schematic diagram of a constant current source using a LM317 regulated integrated circuit
  • 25 is a fifth embodiment of the present invention, and the second embodiment of the present invention is implemented by using the constant current source circuit of FIG. 24-4;
  • Figure 26 is a schematic diagram of the sixth embodiment
  • Figure 27 is a schematic diagram of the seventh embodiment. detailed description
  • FIG. 19 is a first embodiment, as shown in FIG. 19, the difference from FIG. 1-1 is that: the constant current source II is used instead of the original resistor R1, and the main body of the circuit is a self-excitation push-pull converter.
  • the working principle of the circuit is:
  • the constant current source II replaces the original bias resistor R1, but the current direction is the same.
  • the current supplied by the bases of transistors TR1 and TR2 is constant. Observe the current of one of the collectors of the triode.
  • the constant current source II provides forward bias to the base and emitter of the transistor TR1 and the transistor TR2 through the feedback winding NB1 and the feedback winding NB2, and the transistor TR1 and the transistor TR2 start to conduct, due to the two transistors
  • the electrical characteristics may not be exactly the same. Therefore, one of the transistors will be turned on first. It is assumed that the transistor TR2 is turned on first to generate a collector current.
  • the voltage of the corresponding coil winding NP2 is positive at the power supply terminal, and the terminal connected to the collector of the transistor TR2 is negative. , that is, in the figure, it is up and down.
  • the base feedback winding NB2 also has a positive feedback induced voltage.
  • NB2 is also up and down in the figure. This voltage maintains and increases the base current of the transistor TR2. This is a positive feedback process. Therefore, the transistor TR2 is quickly turned on; correspondingly, the induced voltage of the base feedback winding NB1 corresponding to the transistor TR1, NB1 is also up and down in the figure, reducing the base current of the transistor TR1, and the transistor TR1 is very It is almost complete.
  • the base current reduction portion of the transistor TR1 is all the base current increasing portion of the transistor TR2.
  • the current of the coil winding NP2 corresponding to the transistor TR2, and the magnetic induction intensity generated by this current linearly increase with time.
  • the collector current of the transistor TR2 approaches or reaches its base current.
  • the transistor TR2 will exit the saturation region and enter the amplification region.
  • the collector-to-emitter voltage drop UCE of the transistor TR2 is significantly increased. Accordingly, the voltage across the coil winding NP2 corresponding to the transistor TR2 is reduced by the same value, and the base feedback winding NB2 also exhibits a corresponding induced voltage.
  • this voltage weakens the base current of the transistor TR2, and the collector current of the transistor TR2 is further reduced.
  • This is a process of positive feedback, so that the transistor TR2 is quickly taken out of the amplification state and enters the off state; accordingly,
  • the induced voltage of the base feedback winding NB1 corresponding to the transistor TR1 increases the base current of the transistor TR1 at this moment, and the transistor TR1 is fully fully saturated and turned on.
  • the base current increase portion of the transistor TR1 is all derived from the base current reduction portion of the transistor TR2.
  • the two transistors are alternately turned on to complete the push-pull oscillation mode. Since the total input current of the base is limited by the constant current source I I and does not change with the fluctuation of the operating voltage, the circuit enters a new push-pull oscillation mode when the operating voltage rises. As the working voltage rises, due to the core saturation operation mode, the operating current is not increased, and the no-load loss of the circuit is increased. This also improves the conversion efficiency and achieves the above-mentioned The benefits.
  • FIG. 20 is a second embodiment, as shown in FIG. 20, which is an alternative to the prior art of FIG. 2.
  • the difference from FIG. 19 is that: one end of the capacitor C1 is connected to the feedback winding center tap and constant current source of the transformer B. Connect the other end to the input power ground.
  • the capacitor C1 is no longer like the one shown in Fig. 19, and there is an inrush current to the base and the emitter of the transistor.
  • the circuit realizes the soft start function, that is, the terminal voltage of the capacitor C1.
  • the terminal voltage to the capacitor C1 rises sufficiently.
  • the transistors TR1 and TR2 are turned on, the circuit enters the push-pull oscillation.
  • the second embodiment operates in the same manner as the first embodiment. I won't go into details here.
  • FIG. 21 is a third embodiment, which is an alternative to the prior art FIG. 3, in which the prior art FIG. 3 and the present embodiment are generally used in the triodes TR1 and TR2 in order to better perform circuit performance.
  • a low-voltage Zener diode is connected in parallel from the base to the emitter.
  • the value of the low-voltage Zener diode is generally lower than the base-to-emitter reverse withstand voltage of the transistors TR1 and TR2.
  • the reverse withstand voltage is generally between 5V and 7V. .
  • a Zener diode of 5. 6V or less is taken.
  • the cathode of the Zener diode is connected to the base of the transistor TR1 or TR2, and the anode of the Zener diode is connected to the emitter of the transistor TR1 or TR2.
  • the main purpose of the Zener diode is to prevent the back pressure induced by the single feedback winding from penetrating the base to emitter of the transistor TR1 or TR2.
  • the circuit uses the base-to-emitter of the transistors TR1 and TR2 to operate as a 5 to 7V Zener in the reverse breakdown state.
  • the working principle of the circuit of the third embodiment is:
  • the constant current source II replaces the original bias resistor R1, but the current direction is the same.
  • the presence of the flow source provides constant current to the TR2 base of the two push-pull transistors TR1. Observe the current of one of the collectors of the triode.
  • the base current is limited to a specific value, the circuit operates in a new manner to achieve push-pull oscillation, which is briefly described as follows:
  • the constant current source II directly supplies the base of the transistor TR1, and the forward bias is provided to the base and the emitter of the transistor TR2 through the feedback winding NB. Since the feedback winding NB has a low internal resistance, it is close to 0. Ohm, triode TR1 and transistor TR2 start to conduct. Since the electrical characteristics of the two triodes cannot be exactly the same, one of the triodes will conduct first, assuming that the triode TR2 is turned on first, generating collector current, and its corresponding coil winding. The voltage of NP2 is positive at the power supply terminal, and the end connected to the collector of transistor TR2 is negative, that is, it is up and down in the figure.
  • the base feedback winding NB also has a positive feedback induced voltage.
  • the NB is also up and down in the figure. This voltage maintains and increases the base current of the transistor TR2. This is a positive feedback process.
  • the voltage of the upper end of the feedback winding NB is clamped to 0. 7V, and the induced voltage is the upper one. Positive and negative, at this time, the base voltage of the transistor TR1 must be less than 0. 7V, and in a non-conducting state. That is, the transistor TR1 is quickly cut off completely.
  • the base current reduction portion of the transistor TR1 is all the base current increase portion of the transistor TR2. If the induced voltage of the feedback winding NB exceeds 6V or above, the base and emitter of the transistor TR1 will be reversely broken down. In this case, the above parallel diode can be used.
  • the current of the coil winding NP2 corresponding to the transistor TR2, and the magnetic induction intensity generated by this current linearly increase with time.
  • the collector current of the transistor TR2 approaches or reaches its base current.
  • the transistor TR2 will exit the saturation region and enter the amplification region.
  • the collector-to-emitter voltage drop UCE of the transistor TR2 is significantly increased. Accordingly, the voltage across the coil winding NP2 corresponding to the transistor TR2 is reduced by the same value, and the base feedback winding NB also exhibits a corresponding induced voltage.
  • this voltage weakens the base current of the transistor TR2, and the collector current of the transistor TR2 is further reduced.
  • This is a process of positive feedback, so that the transistor TR2 is quickly taken out of the amplification state and enters the off state; accordingly, The base of the transistor TR1 is reversed
  • the induced voltage of the feed winding NB is reduced until it is reversed, but at this moment the base current of the transistor TR1 is increased, and the transistor TR1 is quickly fully saturated.
  • the base current increasing portion of the transistor TR1 is all from the base current reducing portion of the transistor TR2.
  • the two transistors are alternately turned on to complete the push-pull oscillation mode. Since the total input current of the base is limited by the constant current source I I and does not change with the fluctuation of the operating voltage, the circuit enters a new push-pull oscillation mode when the operating voltage rises. As the working voltage rises, due to the core saturation operation mode, the operating current is not increased, and the no-load loss of the circuit is increased. This also improves the conversion efficiency and achieves the above-mentioned The benefits.
  • Figure 22 is a fourth embodiment. Compared with Figure 21 of the third embodiment, the constant current source is changed to two paths I la and l ib , respectively, and the two transistors are directly biased, which improves the circuit of Figure 21 because The influence of the internal resistance of the feedback winding NB on the circuit is basically the same as that of the third embodiment, and will not be described here.
  • a constant current source is realized by a constant current diode.
  • the pins 1 and 2 correspond to pins 1 and 2 in Figure 23, respectively.
  • the constant current diode is abbreviated as CRD and is the abbreviation of English Current Regulative Diode.
  • Figure 24-2 uses a junction field effect transistor to connect to a constant current source to achieve a constant current source.
  • the pins 1 and 2 correspond to pins 1 and 2 in Figure 23, respectively.
  • the junction field effect transistor is abbreviated as JFET.
  • a constant current source circuit can also be realized by using P channel.
  • Figure 24-3 uses a junction field effect transistor to connect to a constant current source to achieve a constant current source. Adjust the value of the resistor R in Figure 24-4 to easily change the constant current value.
  • the pins 1 and 2 correspond to pins 1 and 2 in Figure 23, respectively.
  • the junction field effect transistor is abbreviated as JFET.
  • a constant current source circuit can also be realized by using P channel.
  • Io is the output current of pin 2 of Figure 24-4
  • UBE is the base and emitter voltage drop of transistor TR202
  • the silicon tube is generally about 0.6V
  • R201 is the resistance of resistor R201.
  • This circuit can also be implemented with an NPN type transistor.
  • the second embodiment is implemented by the constant current source circuit of Fig. 24-4, as shown in Fig. 25.
  • the R302 in the circuit can take a large value, so that the circuit can be optimized as a two-terminal device for convenient use. As shown in Figure 24-5, the constant current effect is slightly worse than the circuit in Figure 24-4. But it also meets the requirements of the circuit.
  • Figure 24-6 shows the principle of a constant current source using a TL431 precision adjustable reference IC.
  • the same constant current source can be realized. It can be realized by other precision adjustable reference integrated circuits.
  • the output current is about :
  • VREF is the reference voltage of the precision adjustable reference integrated circuit, generally 2. 50V or 2. 495V or 1. 25V, R301 is the resistance of resistor R301.
  • Figure 24-7 shows the principle of a constant current source using the LM317 regulator integrated circuit.
  • the constant current source can be realized as well.
  • Other linear regulator circuits can be used. Its output current is approximately: 1.20V
  • Io is the output current of pin 2 of Figure 24-7.
  • Molecular 1. 20V is the reference voltage of LM317. The early LM317 is about 1.25V, and then drops to about 1.20V.
  • R301 is the resistance of resistor R301.
  • Figure 24-1 to Figure 24-7 show a total of seven circuits to implement the constant current source of Figure 23, no matter which constant current source is used as the DC bias of the self-excitation push-pull converter, it should be regarded as the present invention. protected range. It is not limited to the above seven kinds of constant current source circuits.
  • FIG. 25 is a fifth embodiment of the present invention implemented by using the constant current source circuit of FIG. 24-4. As shown in FIG. 25, in addition to the constant current source circuit, other working principles are the same as those in the second embodiment.
  • Figure 26 is a sixth embodiment, which is an embodiment of the present invention applied to a classical Jensen circuit, the main body of which uses a classical Jensen circuit.
  • the bias circuit is replaced by a constant current source, and its working principle is similar to that of the first embodiment.
  • the main transformer B1 and the transformer B2 are connected in parallel through the resistor Rb, and the bases of the transistors TR1 and TR2 can also induce a positive feedback signal. , to achieve push-pull oscillation.
  • the current Jensen circuit of the prior art has a rise in the base current of the transistor as the operating voltage rises, resulting in a significant increase in the collector current. Due to the presence of the constant current source II, the current supplied to the bases of the two push-pull transistors TR1 and TR2 is constant, so that when the input voltage is increased, the corresponding effects of Table 4 can be achieved.
  • the output of the seventh embodiment adopts a well-known synchronous rectification circuit.
  • Other principles are the same as those in the second embodiment, and the output voltage and the input voltage can be linearly synchronized, and the voltage is linear in the wide voltage input range. Synchronize.
  • the synchronous rectification circuit in Figure 27 is a basic self-driving circuit.
  • the signal driving the synchronous rectification FET gate can come from other independent windings or other circuits.
  • it can also be used in synchronous rectification FETs.
  • the gate is connected to the capacitor, or a common technical means such as adding a resistor divider network is used to protect the gate of the synchronous rectification FET.

Abstract

一种自激推挽式变换器。在该变换器的推挽用三极管（TR1，TR2）基极的直流回路到有效供电端之间设置有恒流源（II），利用恒流源（II）向推挽用三极管（TR1，TR2）基极提供恒定电流。随着工作电压升高，由于推挽用三极管（TR1，TR2）的基极电流被恒流源（II）所限制，其集电极电流无法增加，电路进入非磁心磁饱和的推挽工作方式。

Description

一种自激推挽式变换器
技术领域
本发明涉及 DC-DC或 DC-AC变换器， 特别涉及工业控制与照明行业。 背景技术
现有的自激推挽式变换器， 电路结构来自 1955年美国罗耶 （G. H. Royer)发 明的自激振荡推挽晶体管单变压器直流变换器，这也是实现高频转换控制电路的 开端； 部分电路来自 1957年美国查赛 （Jen Sen, 有的地方译作 "井森"）发明 的自激式推挽双变压器电路， 后被称为自振荡 Jensen电路； 这两种电路， 后人 统称为自激推挽式变换器。 自激推挽式变换器在电子工业出版社的《开关电源的 原理与设计》 第 67页至 70页有描述， 该书 ISBN号 7-121-00211-6。 电路的主 要形式为上述著名的 Royer电路和自振荡 Jensen电路。
图 1-1为自激推挽式变换器常见应用， 电路结构为 Royer电路， 其中与偏置 电阻 R1 并联的电容 C1在很多场合可以省去， 在 ZL专利号为 03273278. 3， 公开 日期： 2004年 8月 25日， 名称为 《自激推挽式变换器》 文中， 提供了一种带软 启动功能的 Royer电路， 参见图 2， 解决了图 1_1中电容 C1在开机时对推挽用 开关三极管的冲击。
图 1-2也是一种应用方式， 电路结构仍为 Royer电路， 把原一只偏置电阻 R1拆为两只 Rlu和 Rid串联而已， 多用于较高的工作电压输入。 同样， 与偏置 电阻 Rlu 并联的电容 C1在很多场合可以省去， 故图 1-2中电容 C1以虚线绘制。
图 3也是常见的 Royer电路， 简化了反馈绕组的绕法， 其直流信号回路， 晶 体三极管 TR1和 TR2的工作点是一样的，但电路处于自激振荡状态时， 晶体三极 管 TR1和 TR2的工作有差异。 在公开号 US 2007182342 (Al)， 公开日期为 2007-08-09的 ((LCD BACKLIGHT DRIVER)) 中， 使用的就是类似图 3的 Royer电 路作为单元电路。而图 4是图 3电路的原形， 图 4的主要特征是使用两只偏置电 阻 Rla和 Rlb， 分别置于推挽用开关三极管的基极到有效供电端； 图 3把原本两 只偏置电阻 Rla和 Rlb， 简化为一只， 是在图 4基础上的一种节约成本方案。 在 公开号 US 2006250822 (Al)，公开日期为 2006- 11-09 ^((Switching power supply apparatus)) 中， 使用的就是类似图 4的电阻偏置方法作为单元电路。
图 5是常见的一种 Royer电路。 由于在供电回路中串入电感 Ll， 且在推挽 用开关三极管集电极之间并联一只电容 CL， 电路输出接近正弦波， 常见于节能 灯电子整流器等电路上， 同样， 也可以简化反馈绕组的绕法， 而采用类似图 3、 图 4的变形方法。
Royer电路的振荡频率是电源电压的函数， 在电子工业出版社的《开关电源 的原理与设计》 第 68页第 18行有描述， 该书 ISBN号 7-121-00211-6。 这里引 用如下： f = lO4 Hz 公式（l)
ABwSN
式中： f为振荡频率， BW为工作磁感应强度 (Τ)， Ν为线圈匝数， S为磁心有 效截面积。
图 1-1的电路结构为： 输入滤波电容 c连接于电压输入端与地之间， 对输入 电压进行滤波； 滤波后的输入电压接入启动电路， 启动电路由偏置电阻 R1和电 容 CI并联组成， 在较高的电源电压输入时， C1可以省去； 偏置电阻 R1的两端 分别与电压输入端以及为两个推挽晶体管 TR1 TR2基极提供正反馈的耦合变压 器 B原边线圈 NB1和 NB2的中心抽头连接； 两个推挽晶体管 TR1 TR2的发射极 共地， 两个集电极分别连接耦合变压器原边线圈 NP1和 NP2的两个端头， 基极连 接耦合变压器原边线圈 NB1和 NB2的两个端头，原边线圈 NP1和 NP2中的中心抽 头连接电压输入端； 耦合变压器 B的副边线圈 NS连接输出滤波电路至电压输出
W °
其工作原理简述为： 参见图 1-1 Royer电路是利用磁心饱和特性进行推挽 振荡，接通电源瞬间， 偏置电阻 R1和电容 C1并联回路通过线圈 NB1和 NB2绕组为 三极管 TR1和 TR2的基极、发射极提供了正向偏压， 两只三极管 TR1和 TR2开始导 通， 由于两个三极管特性不可能完全一样， 因此， 其中一只三极管会先导通， 假 设三极管 TR2先导通， 产生集电极电流 Ie2， 其对应的线圈 NP2绕组的电压为上正 下负， 根据同名端关系， 其基极线圈 NB2绕组也出现上正下负的感应电压， 这个 电压增大了三极管 TR2的基极电流， 这是一个正反馈的过程， 因而很快使三极管 TR2饱和导通； 相应地， 三极管 TR1对应的线圈 NB1绕组的电压为上正下负， 这 个电压减小了三极管 TR1的基极电流， 三极管 TR1很快完全截止。
三极管 TR2对应的线圈 NP2绕组里的电流， 以及这个电流产生的磁感应强度 随时间线性增加， 但磁感应强度增加到耦合变压器 B磁心的饱和点 Bm时， 线圈的 电感量迅速减小， 从而使三极管 TR2开关管的集电极电流急剧增加， 增加的速率 远大于基极电流的增加， 三极管 TR2开关管脱离饱和， 三极管 TR2开关管的集电 极到发射极的压降 UCE增大， 相应地， 变压器 NP2绕组上的电压就减小同一数值， 线圈 NB2绕组感应的电压减小， 结果使三极管 TR2开关管基极电压也降低， 造成 三极管 TR2开关管向截止方向变化， 此时， 变压器线圈上的电压将反向， 使另一 只三极管 TR1导通， 此后， 重复进行这一过程， 形成推挽振荡。 绕组 Ns的输出端 的波形如图 6所示。
其特点为： 利用磁心饱和特性进行推挽振荡， 耦合变压器输出波形为近似方 波， 电路的变换效率较高。 图 5的电路由于在供电回路中串入电感 Ll， 且在推 挽用开关三极管集电极之间并联一只电容 CL， 电路输出波形接近正弦波。
另一个与 Royer电路相似的结构，就是开关驱动功能与主功率变压器脱离的 电路，如图 7所示。这个电路就是著名的自振荡 Jensen电路， 中文常音译为 "井 森" 电路， 电路的自振荡频率和驱动功能， 改由磁饱和的变压器 B2来实现， 因 此， 主功率变压器 B1能工作在不饱和状态。 其中， C1或 Cla—般只保留一只， Cla是 C1的等效接法， 但是接在 Cla时， 电路可以实现软启动， 需要说明的是 C1和 Cla同时去除， 电路也是可以工作的。
虽然 B2出现磁饱和， 因为 B2体积小， 磁饱和消耗的能量小， 电路的总体效 率高。 与相同条件下的 Royer电路比较， 在工作电压、 负载、 温度发生变化时， Jensen电路的自振荡频率相对比较稳定。
当然， Jensen电路的应用较广， 其电路的形式也多样， 主要也是体现在图 7 中 R1偏置方式的变化。
上述的图 1至图 7， 不包括图 6， 都是现有的自激推挽式变换器。 它们有共 同的缺点为：
1、 工作电压适应性差。
负载空载时， 随着电路的工作电压上升， 电路的输入电流， 即相当于电路的 静态电流也随之上升， 引起电路的空载损耗增大。 表一为 Royer电路实测参数。 如使用图 1_1的电路， 做成输入直流 5V， 输 出直流 5V， 输出电流为 200mA的变换器， 即输出功率 1W。 电路的主要参数为： 电容 C为 luF电容， 电阻 R1为 1Κ Ω， 电容 C1为 0. 047uF电容， 三极管 TR1和 TR2为放大倍数在 200倍左右的开关三极管， 其集电极最大工作电流为 1A。变压 器的后续输出采用图 8的电路结构， 图 8为公知的全波整流电路。
在整个测试过程中， 没有对电路任何参数进行调整， 或器件更换。 工作电压 达 12V及以上时， 测试时间极短， 因为电路的空载损耗极大， 测试时间稍长， 电 路就会损坏。
表一
Figure imgf000004_0001
测试时， 在 5V时输出 200mA， 在其它工作电压下， 尽可能地配合改变负载， 把输出电流调大， 让其接近 200mA， 直到输出电压会跌落达到 5%时， 停止改变负 表二为 Jensen电路实测参数。 电路采用图 7的电路， 做成输入直流 5V， 输 出直流 5V， 输出电流为 200mA的变换器， 即输出功率 1W。 电路的主要参数为： 电容 C为 luF电容， 电阻 R1为 1Κ Ω， 电容 Cla为 0. 047uF电容， 三极管 TR1和 TR2为放大倍数在 200倍左右的开关三极管， 其集电极最大工作电流为 1A。变压 器的后续输出采用图 8的电路结构。
表二
Figure imgf000004_0002
电路的变换效率为：
Vout X lout
η = χ 100 -公式 (2)
Vin x Iin 式中： Vin为工作电压， 即输入电压， I in为输入电流； Vout为输出电压， lout为输出电流。
从表一可以看出， 工作在 5V的电路， 若让其工作在 8V下， 其自身的损耗已 达 280mW，在微功率 DC/DC变换器中， 这是勉强能接受的， 而在 12V的工作电压 下， 其自身损耗已达 828mW， 而在 20V下， 其自身损耗已达 3600mW， 即 3. 6W， 电路工作时间超过 3秒， 电路就会损坏。 因此， 电路的变换效率也随着工作电压 上升而下降。 Jensen电路有相同的问题。 即工作电压升高， 引起现有自激推挽 式变换器的空载工作电流上升过快、 空载损耗上升过快、 变换效率下降。
2、 抗浪涌性能差。 基于上述原因， 当输入电压出现浪涌时， 电路极易损坏， 主要都是损坏开关三极管。
3、 设计工作在其它电压值的自激推挽式变换器， 都存在这种缺点。 发明内容
有鉴如此，本发明要解决的技术问题是： 使自激推挽式变换器工作电流随工 作电压上升时， 上升缓慢或不上升， 且当自激推挽式变换器输入电压中出现浪涌 时， 自激推挽式变换器具备一定的抗浪涌能力， 不易损坏开关三极管。
为解决上述技术问题，本发明提供一种自激推挽式变换器， 其特征在于推挽 用开关三极管基极的直流回路到有效供电端之间为一恒流源； 即在 Royer或 Jensen电路的基础上， 取消推挽用开关三极管基极的直流回路到有效供电端之 间的偏置电阻， 分别增加恒流源代替原偏置电阻。
所述的恒流源的电流方向应与被取消的原偏置电阻电流方向一致，即恒流源 的电流方向为流入至 NPN晶体三极管的基级；或恒流源的电流方向从 PNP晶体三 极管的基级流入至所述的恒流源。
所述的恒流源可以通过并联获得更大的恒流值。
所述的恒流源可以是任何一种半导体器件或实现恒流的电子线路。
本发明的工作原理是， 恒流源取代了原偏置电阻， 但电流方向一致， 当自激 推挽式变换器工作电压由于某种原因升高时， 由于恒流源的存在， 向两只推挽三 极管基极提供的电流是恒定不变的。 观察其中的一只三极管集电极电流进行对 比， 使用现有技术时， 随着工作电压升高， 所需的磁心饱和电流急剧增大， 弓 I起 电路的空载损耗增加，变换效率降低。本发明由于基极电流被限制在一个特定的 值， 电路进入一种新的方式工作， 实现推挽振荡， 简述如下：
接通电源瞬间， 恒流源通过反馈绕组 1和反馈绕组 2为三极管 1和三极管 2 的基极、发射极提供了正向偏压， 三极管 1和三极管 2开始导通， 由于两个三极 管特性不可能完全一样，因此，其中一只三极管会先导通，假设三极管 1先导通， 产生集电极电流， 其对应的线圈绕组 1的电压为电源端正，与三极管 1集电极连 接的一端为负， 根据同名端关系， 其基极反馈绕组 1也出现正反馈的感应电压， 这个电压维持了、增加了三极管 1的基极电流， 这是一个正反馈的过程， 因而很 快使三极管 1饱和导通； 相应地， 三极管 2对应的基极反馈绕组 2的感应电压， 减小了三极管 2的基极电流，三极管 2很快完全截止。三极管 2的基极电流减小 部分， 全部成为三极管 1的基极电流增加部分。
三极管 1对应的线圈绕组 1的电流，以及这个电流产生的磁感应强度随时间 线性增加， 磁感应强度增加到耦合变压器 B磁心的饱和点 Bm之前时， 三极管 1 的集电极电流接近或达到其基极电流和其放大倍数乘积时，三极管 1会退出饱和 区， 进入放大区。 三极管 1的集电极到发射极的压降 UCE显著增大， 相应地， 三 极管 1对应的线圈绕组 1的两端电压就减小同一数值，其基极反馈绕组 1也出现 相应的感应电压， 也在减小， 这个电压减弱了三极管 1的基极电流， 这是一个正 反馈的过程， 因而很快使三极管 1退出放大状态， 进入截止状态； 相应地， 三极 管 2对应的基极反馈绕组 2的感应电压， 却增加了三极管 2的基极电流， 三极管 2很快完全饱和导通。 三极管 2的基极电流增加部分， 全部来自三极管 1的基极 电流减小部分。
这样， 两只三极管交替导通， 完成推挽振荡模式。 由于基极总输入电流， 被 恒流源限制， 不随工作电压波动而改变， 所以电路在工作电压升高时， 电路进入 一种新的推挽振荡模式。
当然， 在适当负载、适当工作电压等条件下时， 电路仍可工作在利用磁心饱 和特性进行推挽振荡。
由于在工作电压提高后， 使用现有技术的偏置电阻能提供更大的基极电流， 电路利用磁心饱和特性进行推挽振荡，这时的集电极电流过大，极易烧毁三极管。 而使用恒流源偏置后， 电路进入一种全新的推挽振荡模式， 三极管集电极最大电 流被基极电流限制，该电流最大值与恒流源输出电流和三极管的放大部数的乘积 有关。 从而使推挽三极管工作在安全区内。
Jensen电路的工作原理与此相似。
作为上述技术方案的进一步改进，在输出端增加同步整流电路， 由于同步整 流的效率高， 整流时压降损失小， 可以提高电路的工作效率； 同时在较宽的输入 电压范围内， 可以实现输出电压和输入电压线性同步。
有益效果
本发明的优点在于使用恒流源作为偏置后， 随着工作电压升高， 电路在同等 条件下， 空载功耗降低， 电路的变换效率与现有技术相比有明显提升， 下面以二 组实际测试数据说明。
同样使用图 1-1的 Royer电路， 做成输入直流 5V， 输出直流 5V， 输出电流 为 200mA的变换器， 即输出功率 1W。 变压器的输出采用图 8的电路结构。 电路 的主要参数为： 电容 C为 luF电容， 电阻 R1替换为 4. 3mA恒流源， 电容 C1为 0. 047uF电容， 三极管 TR1和 TR2为放大倍数在 200倍左右的开关三极管， 其集 电极最大工作电流为 1A。 实测其参数如下表三：
表三
Figure imgf000006_0001
用表三的数据与表一的数据进行对比， 可以看到， 在同等条件下， 本发明的 自激推挽式 Royer变换器和现有的自激推挽式 Royer变换器相比，在工作电压升 高后， 空载输入电流、 空载损耗、 变换效率都有显著改进： 1、 相同工作电压下， 使用本发明的电路空载电流明显下降。 如在 20V工作 电压下， 现有技术为 180mA，且该电流随着工作时间延长， 几秒后急剧升高至 300mA以上。 本发明为 30mA， 且电流值非常稳定， 长期工作不出现上升。 其对比 图参见图 9。
2、相同工作电压下，本发明的空载损耗降低。如在 20V下，现有技术为 3600mW， 本发明为 600mW。 其对比图参见图 10。
3、相同工作电压下，本发明的变换效率提升。如在 20V下，现有技术为 58. 6%, 本发明为 86. 0%。 其对比图参见图 11。
另外， 使用本发明应用于图 7电路， 做成输入直流 5V， 输出直流 5V， 输出 电流为 200mA的变换器， 即输出功率 1W。 电路的主要参数为： 电容 C为 luF电 容， 电阻 R1替换为 4. 3mA恒流源， 电容 Cla为 0. 047uF电容，三极管 TR1和 TR2 为放大倍数在 200倍左右的开关三极管， 其集电极最大工作电流为 1A。 变压器 的后续输出采用图 8的电路结构。 实测其参数如下表四：
表四
Figure imgf000007_0001
自激推挽式 Jensen变换器和现有的自激推挽式 Jensen变换器相比，在工作电压 升高后， 空载输入电流、 空载损耗、 变换效率都有显著改进：
1、 相同工作电压下， 使用本发明的电路空载电流明显下降。 如在 15V工作 电压下， 现有技术为 200mA，且该电流随着工作时间延长， 几秒后急剧升高至 300mA以上。 本发明为 37mA， 且电流值非常稳定， 长期工作不出现上升。 其对比 图参见图 12。
2、相同工作电压下，本发明的空载损耗降低。如在 15V下，现有技术为 3000mW， 本发明为 555mW。 其对比图参见图 13。
3、相同工作电压下，本发明的变换效率提升。如在 15V下，现有技术为 22. 6%, 本发明为 64. 6%。 其对比图参见图 14。
其它有益效果一：
本发明的优点还在于， 自激推挽式 Royer变换器使用恒流源作为偏置后， 随 着工作电压升高， 电路在同等条件下， 输出端出现短路时， 即图 8的 DC OUT短 路， 本发明的短路保护性能有明显提高。 同样使用图 1-1的 Royer电路， 实测对 比数据如下表五：
表五
Figure imgf000007_0002
3 9 44 13 57
4 13 52 15 56
5 18 60 17 55
8 35 80 22 59
12 69 120 29 15
15 瞬间烧毁推挽三极 14
90 30
管
20 瞬间烧毁推挽三极 12
180 30
管
现在技术的 Royer电路在出现负载短路时， 当工作电压提高时， 工作电流同 步升高， 且在 15V及 15V以上工作电压时， 瞬间烧毁推挽三极管； 更换同型号三 极管， 使用本发明的电路测试， 发现在工作电压提高、 负载短路时， 输入总电流 下降。 对比图参见图 15。
上述使用 4. 3mA恒流源偏置短路保护性能明显提高与本发明带来的特殊的 新的推挽振荡模式有关，现有的 Royer自激推挽式变换器， 其短路保护的实现机 理是通过变压器的漏感实现的， 变压器都会存在漏感， 理想的变压器并不存在， 变压器的漏感是初级线圈所产生的磁力线不能都通过次级线圈，因此产生漏磁的 电感称为漏感。 次级线圈通常作输出用， 当次级线圈直接短路时， 这时测出的初 级线圈仍存在电感量，通常近似地认为是漏感。现有的 Royer自激推挽式变换器， 参见图 1-1，当负载出现短路时，等效于 NP1和 NP2的电感量降至一个很小的值， 电路进入高频自激推挽式振荡， 参见公式（1)， 负载短路时， 相当于公式（1)中 SN的乘积变小， 工作频率上升。 随着工作电压提高， 电阻 R1提供的基极电流随 工作电压升高而线性上升， 电路要维持振荡， 磁心要实现磁饱和要消耗更多的能 量， 所以， 负载短路时， 电路的总输入电流， 即工作电流升高。
而本发明电路进入新的推挽振荡模式， 当负载出现短路时， 电路进入高频自 激推挽式振荡时，不是依靠磁心磁饱时开关三极管的集电极电流急聚上升实现推 挽状态翻转， 而是基极电流被限制， 集电极电流无法上升， 导致开关三极管进入 放大区， 实现电路的推挽状态翻转， 不存在磁心进入磁饱和而消耗更多的能量， 由于变压器的线圈， 匝与匝之间存在分布电容， 其等效电路为图 16所示， 图 16 为公知的所有实际电感的等效电路原理图。 R0的电阻较小， CO的容量也很小， 在低频工作时， C0的影响可以忽略不计， 但在负载出现短路时， 电路进入高频 自激推挽式振荡时， 图 16的电路就是一个标准的 LC回路， 振荡的能量在回路中 谐振， 振荡的波形接近正弦波， 由于频率高， 变压器的传输效率低， 振荡产生的 近似正弦波，有其峰值被后续的输出短路回路所限幅而已， 振荡产生的近似正弦 波能量在变压器的初级中谐振， 故消耗的能量小， 体现在输入端， 就是总工作电 流下降。 图 17为短路发生时， 在变压器中加绕 2匝作为检测口， 测出的近似正 弦波高频振荡， 其顶部不光滑， 且有衰减式阻尼振荡， 这是其峰值被后续的输出 短路回路所限幅而产生的特有现象。本发明的电路在负载出现短路时， 工作电压 12V时， 总工作电流为 15mA,总功耗才 180mW， 参见表五， 而现有技术总工作电 流高过 120mA， 功耗高达 1440mW。
尽管在低压时， 本发明没有优势， 但由于工作电压低， 总功耗不会超过任一 只开关三极管的功耗， 电路不会损坏。
即本发明可以让自激推挽式 Royer变换器获得更好的短路保护性能。
其它有益效果二： 本发明的优点还在于， 当输出采用同步整流电路， 可以实现在较宽的输入电 压范围内， 实现输出电压和输入电压线性同步。
自激推挽式 Royer变换器使用恒流源作为偏置后，采用图 8的输出电路， 并 使用同步整流电路取代二极管 D21、 二极管 D22， 本发明的输出电压和输入电压 几乎相等， 同样使用图 1-1的 Royer电路， 实测对比数据如下表五：
表六
Figure imgf000009_0001
为了方便对比，把输出电压与输入电压的比值单独列出， 这样很直观地反映出对 比， 请见表七：
表七
Figure imgf000009_0002
极管 D22, 输出变压器输出绕组的匝数要减少， 所以在 5V输入时， 很难找到合 适的圈数使得输出为 5. 00V, 最后只找到一个比较接近的 4. 97V输出电压。
现在技术的 Royer电路在工作电压提高至 20V, 由于损耗达 3600mW， 见表一 数据， 即使采用同步整流技术使得输出电压线性同步， 也无法工作在 20V的输入 电压下。 而本发明由于在 20V, 空载电流是 30mA， 损耗是 600mW， 参见表三， 仍 可正常工作。本发明解决了工作电压适应性差， 才使得输出电压线性同步成为实 用技术。
从表七可以看出，在 3V至 20V范围内：现有技术，输出精度在 -7. 7%至 15. 9% 之间； 而本发明输出精度稳定在 -3. 7%至 -0. 7%之间。 测试时都采用图 18的接线方式， RL为可变负载，可以有效地减小测量误差。 电流表和电压表均使用 MASTECH®品牌的 MY65型 4位半数字万用表的 200mA档和 20V档或 200V档， 同时使用了四块及四块以上的万用表。
MY65型 4位半数字万用表其测电压时， 内阻为 10Μ Ω， 200mA电流档的内阻 为 1 Ω。 当电流超过 200mA时， 采用了两块电流表置于 200mA档并联测量， 把两 块表的电流读数相加，即为测量值。电流表并联测量是现有电子工程的成熟技术。
VI电压表头为工作电压 Vin， 即输入电压； A1电流表头为输入电流 Iin， 即 为工作电流； V2电压表头为输出电压 Vout， A2电流表头为输出电流 lout; 那么 变换效率可以用公式 (2)计算得出。 附图说明
图 1-1 为自激推挽式变换器中 Royer常见应用电路原理图；
图 1-2为自激推挽式变换器中 Royer常见的另一种应用电路原理图； 图 2 Z为 L专利号为 03273278. 3， 公开的一种带软启动功能的 Royer电路； 图 3为简化反馈绕组的一种 Royer应用电路；
图 4 为图 3电路的原形， 简化反馈绕组的一种 Royer应用电路；
图 5 为一种输出接近正弦波的 Royer应用电路；
图 6 为图 1-1电路中 Ns绕组的输出波形图；
图 7 为自激推挽式变换器中著名的 Jensen电路的常见应用原理图； 图 8 为公知的全波整流电路；
图 9 为 5V转 5V Royer电路不同偏置方法空载输入电流对比图；
图 10 为 5V转 5V Royer电路不同偏置方法空载损耗对比图；
图 11 为 5V转 5V Royer电路不同偏置方法变换效率对比图；
图 12 为 5V转 5V Jensen电路不同偏置方法输入电流对比图；
图 13 为 5V转 5V Jensen电路不同偏置方法空载损耗对比图；
图 14 为 5V转 5V Jensen电路不同偏置方法变换效率对比图；
图 15 为 5V转 5V Royer电路不同偏置方法， 负载短路时， 电路输入总电流 对比图；
图 16 为公知的电感实际等效电路原理图；
图 17 为本发明电路在负载短路时， 变压器中检测到的高频振荡波形； 图 18 为本文中通用使用的测试原理图；
图 19 为本发明第一实施例的电路图；
图 20 为本发明第二实施例的电路图；
图 21 为本发明第三实施例的电路图；
图 22 为本发明第四实施例的电路图；
图 23 为恒流源在电路图中符号；
图 24-1 为恒流二极管构成恒流源的原理图；
图 24-2 为结型场效应管构成恒流源的原理图；
图 24-3 为另一种结型场效应管构成恒流源的原理图；
图 24-4 为双极性 PNP管构成恒流源的原理图；
图 24-5 为另一种双极性 PNP管构成恒流源的原理图；
图 24-6 为采用 TL431精密可调基准集成电路构成的恒流源的原理图； 图 24-7 为采用 LM317稳压集成电路构成的恒流源的原理图； 图 25 为实施例五， 为实施例二采用图 24-4的恒流源电路实现的本发明电 原理图；
图 26 为实施例六原理图；
图 27 为实施例七原理图。 具体实施方式
图 19为第一实施例， 如图 19所示， 较背景技术图 1-1中的不同处在于： 采 用恒流源 I I替代了原电阻 Rl， 电路的主体为自激推挽式变换器， 电路的工作原 理为：
恒流源 I I取代了原偏置电阻 Rl， 但电流方向一致， 当图 19的自激推挽式 变换器工作电压由于某种原因升高时， 由于恒流源的存在， 向两只推挽三极管 TR1和 TR2基极提供的电流是恒定不变的。观察其中的一只三极管集电极电流进 行对比， 使用现有技术时， 随着工作电压升高， 所需的磁心饱和电流急剧增大， 引起电路的空载损耗增加，变换效率降低。本发明由于基极电流被限制在一个特 定的值， 电路进入一种新的方式工作， 实现推挽振荡， 简述如下：
接通电源瞬间， 恒流源 I I通过反馈绕组 NB1和反馈绕组 NB2为三极管 TR1 和三极管 TR2的基极、发射极提供了正向偏压， 三极管 TR1和三极管 TR2开始导 通， 由于两个三极管的电气特性不可能完全一样， 因此， 其中一只三极管会先导 通， 假设三极管 TR2先导通， 产生集电极电流， 其对应的线圈绕组 NP2的电压为 电源端正，与三极管 TR2集电极连接的一端为负， 即在图中为上正下负。 根据同 名端关系， 其基极反馈绕组 NB2也出现正反馈的感应电压， NB2在图中也是上正 下负，这个电压维持了、增加了三极管 TR2的基极电流，这是一个正反馈的过程， 因而很快使三极管 TR2饱和导通； 相应地， 三极管 TR1对应的基极反馈绕组 NB1 的感应电压， NB1在图中也是上正下负， 减小了三极管 TR1的基极电流， 三极管 TR1很快完全截止。 三极管 TR1的基极电流减小部分， 全部成为三极管 TR2的基 极电流增加部分。
三极管 TR2对应的线圈绕组 NP2的电流，以及这个电流产生的磁感应强度随 时间线性增加， 磁感应强度增加到耦合变压器 B磁心的饱和点 Bm之前时， 三极 管 TR2的集电极电流接近或达到其基极电流和其放大倍数乘积时，三极管 TR2会 退出饱和区， 进入放大区。 三极管 TR2的集电极到发射极的压降 UCE显著增大， 相应地，三极管 TR2对应的线圈绕组 NP2的两端电压就减小同一数值， 其基极反 馈绕组 NB2也出现相应的感应电压， 也在减小， 这个电压减弱了三极管 TR2的基 极电流， 三极管 TR2的集电极电流进一步减小， 这是一个正反馈的过程， 因而很 快使三极管 TR2退出放大状态， 进入截止状态； 相应地， 三极管 TR1对应的基极 反馈绕组 NB1的感应电压，此刻却增加了三极管 TR1的基极电流， 三极管 TR1很 快完全饱和导通。三极管 TR1的基极电流增加部分， 全部来自三极管 TR2的基极 电流减小部分。
这样， 两只三极管交替导通， 完成推挽振荡模式。 由于基极总输入电流， 被 恒流源 I I限制， 不随工作电压波动而改变， 所以电路在工作电压升高时， 电路 进入一种新的推挽振荡模式。随着工作电压升高，由于脱离了磁心饱和工作方式， 所以不会引起工作电流急聚增加， 不会引起电路的空载损耗急聚增加， 同样也就 提高了变换效率， 实现了前文所说的有益效果。
图 20为第二实施例， 如图 20所示， 是对现有技术图 2的发明替代， 较图 19中的不同处在于： 电容 C1一端连接在变压器 B的反馈绕组中心抽头与恒流源 相连接点， 另一端连接在输入电源地线上。 不仅可以实现图 19的全部功能， 同 时， 由于 C1是连接在地线上， 在电路上电瞬间， 电容 C1不再像图 19那样， 对 三极管的基极与发射级存在一个冲击电流。 相反， 在本实施中， 由于电容 C1两 端电压不能突变， 电路实现了软启动功能， 即电容 C1的端电压， 随着恒流源对 电容 C1充电， 至电容 C1的端电压升高至足以让三极管 TR1和 TR2导通时， 电路 才进入推挽振荡。
第二实施例电路工作原理同第一实施例。 这里不再赘述。
图 21为第三实施例， 这是对应现有技术图 3的发明替代， 现有技术图 3和 本实施例在实际应用中， 为了更好的发挥电路性能， 一般会在三极管 TR1和 TR2 的基极至发射极分别并联一只低压稳压二极管，低压稳压二极管取值一般低于三 极管 TR1和 TR2的基极至发射极的反向耐压， 反向耐压一般在 5V至 7V之间。一 般取 5. 6V以下的稳压二极管。 稳压二极管的阴极连接在三极管 TR1或 TR2的基 极上，稳压二极管的阳极连接在三极管 TR1或 TR2的发射极上。稳压二极管的主 要作用是防止单反馈绕组引发的反压击穿三极管 TR1或 TR2的基极至发射极。
若没有并联稳压二极管，电路是利用三极管 TR1和 TR2的基极至发射极在反 向击穿状态下相当于一只 5至 7V的稳压管工作。第三实施例电路的工作原理是: 恒流源 I I取代了原偏置电阻 Rl， 但电流方向一致， 当图 21的自激推挽式 变换器工作电压由于某种原因升高时， 由于恒流源的存在， 向两只推挽三极管 TR1的 TR2基极提供的电流是恒定不变的。观察其中的一只三极管集电极电流进 行对比， 使用现有技术时， 随着工作电压升高， 所需的磁心饱和电流急剧增大， 引起电路的空载损耗增加，变换效率降低。本发明由于基极电流被限制在一个特 定的值， 电路进入一种新的方式工作， 实现推挽振荡， 简述如下：
接通电源瞬间， 恒流源 I I直接对三极管 TR1的基极供电， 同时通过反馈绕 组 NB为三极管 TR2的基极、发射极提供了正向偏压，由于反馈绕组 NB内阻很低， 接近 0欧姆，三极管 TR1和三极管 TR2开始导通， 由于两个三极管的电气特性不 可能完全一样， 因此， 其中一只三极管会先行导通， 假设三极管 TR2先导通， 产 生集电极电流， 其对应的线圈绕组 NP2的电压为电源端正，与三极管 TR2集电极 连接的一端为负， 即在图中为上正下负。 根据同名端关系， 其基极反馈绕组 NB 也出现正反馈的感应电压， NB在图中也是上正下负， 这个电压维持了、 增加了 三极管 TR2的基极电流， 这是一个正反馈的过程， 因而很快使三极管 TR2饱和导 通； 相应地， 由于三极管 TR2的基极在这时处于 0. 7V左右的电压， 即反馈绕组 NB的上端被钳位至 0. 7V， 而感应电压为上正下负， 这时三极管 TR1的基极电压 一定小于 0. 7V, 而处于不导通状态。 即三极管 TR1很快完全截止。 三极管 TR1 的基极电流减小部分， 全部成为三极管 TR2的基极电流增加部分。 若反馈绕组 NB的感应电压超过 6V或以上， 三极管 TR1的基极和发射极会被反向击穿， 这时 可以用上述并联二极管的方法解决。
三极管 TR2对应的线圈绕组 NP2的电流，以及这个电流产生的磁感应强度随 时间线性增加， 磁感应强度增加到耦合变压器 B磁心的饱和点 Bm之前时， 三极 管 TR2的集电极电流接近或达到其基极电流和其放大倍数乘积时，三极管 TR2会 退出饱和区， 进入放大区。 三极管 TR2的集电极到发射极的压降 UCE显著增大， 相应地，三极管 TR2对应的线圈绕组 NP2的两端电压就减小同一数值， 其基极反 馈绕组 NB也出现相应的感应电压， 也在减小， 这个电压减弱了三极管 TR2的基 极电流， 三极管 TR2的集电极电流进一步减小， 这是一个正反馈的过程， 因而很 快使三极管 TR2退出放大状态， 进入截止状态； 相应地， 三极管 TR1的基极因反 馈绕组 NB的感应电压减小直至反向， 此刻却增加了三极管 TR1的基极电流， 三 极管 TR1很快完全饱和导通。三极管 TR1的基极电流增加部分， 全部来自三极管 TR2的基极电流减小部分。
这样， 两只三极管交替导通， 完成推挽振荡模式。 由于基极总输入电流， 被 恒流源 I I限制， 不随工作电压波动而改变， 所以电路在工作电压升高时， 电路 进入一种新的推挽振荡模式。随着工作电压升高，由于脱离了磁心饱和工作方式， 所以不会引起工作电流急聚增加， 不会引起电路的空载损耗急聚增加， 同样也就 提高了变换效率， 实现了前文所说的有益效果。
图 22为第四实施例， 与第三实施例的图 21相比较， 恒流源改为两路 I la和 l ib, 分别对两只三极管直接提供偏置， 改善了图 21电路中， 因为反馈绕组 NB 内阻对电路引发的影响， 其工作原理基本和第三实施例的相同， 这里不再赘述。
在上述的实施例中， 恒流源直接使用了图 23的电气符号取代， 事实上， 实 现这一恒流源有多种公知的形式：
如图 24-1 使用恒流二极管实现恒流源， 其 1、 2引脚分别对应图 23中的 1、 2引脚。 恒流二极管縮写为 CRD， 为英文 Current Regulative Diode的縮写。
如图 24-2 使用结型场效应管接成恒流源实现恒流源。 其 1、 2引脚分别对 应图 23中的 1、 2引脚。 结型场效应管縮写为 JFET。 采用 P沟道同样可以实现 恒流源电路。
如图 24-3 使用结型场效应管接成恒流源实现恒流源。调节图 24-4的电阻 R 的取值， 可以方便地改变恒流电流值。 其 1、 2引脚分别对应图 23中的 1、 2引 脚。 结型场效应管縮写为 JFET。 采用 P沟道同样可以实现恒流源电路。
如图 24-4使用两只 PNP型三极管接成恒流源实现恒流源，电路是经典电路， 其输出电流约为：
U
Io - BE
公式 (3)
-201
式中 Io为图 24-4的 2脚输出电流， UBE为三极管 TR202的基极、发射极压 降， 硅管一般取 0. 6V左右， R201为电阻 R201的阻值。 该电路同样可以用 NPN 型三极管实现。实施例二采用图 24-4的恒流源电路实现的本发明，如图 25所示。
当三极管 TR201和 TR202的放大倍数较大时，电路中的 R302可以取值较大， 这样电路可以优化为二端子器件， 以方便使用。 如图 24-5所示， 其恒流效果略 差于图 24-4的电路。 但也满足电路的使用要求。
图 24-6 为采用 TL431精密可调基准集成电路构成的恒流源的原理图， 一样 可以实现恒流源，采用其它精密可调基准集成电路一样可以实现，如用 TL432, 其 输出电流约为：
Figure imgf000013_0001
Io = '公式 (4)
^301
式中 Io为图 24-6的 2脚输出电流， VREF为精密可调基准集成电路的基准 电压， 一般是 2. 50V或 2. 495V或 1. 25V， R301为电阻 R301的阻值。
图 24-7 为采用 LM317稳压集成电路构成的恒流源的原理图， 一样可以实现 恒流源， 采用其它的线性稳压集电路同样可以实现。 其输出电流约为： 1.20V
式中 Io为图 24-7的 2脚输出电流， 分子 1. 20V为 LM317的基准电压， 早 期的 LM317为 1. 25V左右， 后降为 1. 20V左右， R301为电阻 R301的阻值。
图 24-1至图 24-7共列举了 7种电路实现图 23的恒流源， 无论使用何种恒 流源作为自激推挽式变换器的直流偏置， 都应视为本发明的保护范围。不仅仅局 限于上述的种 7种恒流源电路。
图 25为 实施例五，采用图 24-4的恒流源电路实现的本发明，如图 25所示， 除了恒流源电路， 其它的工作原理同实施例二。
图 26 为实施例六， 是本发明应用于经典的 Jensen 电路的实施例， 电路的 主体采用经典的 Jensen电路。 偏置电路替换为恒流源， 其工作原理类似于实施 例一，主变压器 B1和变压器 B2之间， 是通过电阻 Rb并联的，三极管 TR1和 TR2 的基极同样也可以感应出正反馈的信号， 实现推挽振荡。 同样， 当输入工作电压 升高时， 现有技术的 Jensen电路， 其三极管基极电流随工作电压上升而上升， 导致集电极电流也大幅上升。 由于恒流源 II的存在， 向两只推挽三极管 TR1和 TR2基极提供的电流是恒定不变的， 所以， 当输入电压升高时， 可以实现表四所 对应的有益效果。
图 27实施例七， 实施例七的输出采用了公知的同步整流电路， 其它原理同 实施例二， 可以实现输出电压和输入电压线性同步， 实现在较宽的电压输入范围 内， 隔离式电压线性同步。
图 27中的同步整流电路是基本自驱动电路， 在实际应用时， 驱动同步整流 场效应管栅极的信号可以来自于其它的独立绕组或其它电路； 另外， 也可以在同 步整流场效应管的栅极串入电容，或增加电阻分压网络等常用的技术手段实现对 同步整流场效应管栅极的保护。
以上仅是本发明的优选实施方式， 应当指出的是， 上述优选实施方式不应视 为对本发明的限制，本发明的保护范围应当以权利要求所限定的范围为准。对于 本技术领域的普通技术人员来说，在不脱离本发明的精神和范围内， 还可以做出 若干改进和润饰， 这些改进和润饰也应视为本发明的保护范围。如采用公知的三 极管复合管代替相应的三极管； 用 PNP型三极管代替 NPN型三极管， 而把电 源输入电压极性反过来。

Claims

权利要求
1、 一种自激推挽式变换器， 其特征在于： 推挽用开关三极管基极的直流回路到 有效供电端之间为一恒流源。
2、根据权利要求 1所述的自激推挽式变换器， 其特征在于： 所述的推挽用开关 三极管为 NPN晶体三极管，所述的恒流源的电流方向为流入至所述的 NPN晶体三 极管的基极。
3、根据权利要求 1所述的自激推挽式变换器， 其特征在于： 所述的推挽用开关 三极管为 PNP晶体三极管，所述的恒流源的电流方向为从所述的 PNP晶体三极管 的基极流入至所述的恒流源。
4、根据权利要求 1或 2或 3所述的自激推挽式变换器， 其特征在于： 所述的推 挽变换器的输出端为一同步整流电路。
5、 一种权利要求 1中实现推挽振荡的方法， 其特征在于： 通过在推挽用开关三 极管基极的直流回路与有效供电端之间设置恒流源；利用所述的恒流源向所述的 推挽三极管基极提供恒定电流， 随着工作电压升高， 由于所述的推挽三极管基极 电流被所述的恒流源限制，其集电极电流无法增加， 电路进入非磁心磁饱和的推 挽工作方式。
6、 根据权利要求 5所述的方法， 其特征在于： 所述的推挽用开关三极管采用 NPN晶体三极管， 所述的恒流源的电流方向为流入至所述的 NPN晶体三极管的基 极。
7、 根据权利要求 6所述的方法， 其特征在于： 所述的推挽用开关三极管采用 PNP晶体三极管， 所述的恒流源的电流方向为从所述的 PNP晶体三极管的基极流 入至所述的恒流源。
8、根据权利要求 5或 6或 7所述的方法， 其特征在于： 所述的恒流源通过并联 恒流源以提供更大的三极管基极恒流值。
9、根据权利要求 8所述的方法， 其特征在于： 所述的恒流源为可实现恒流的半 导体器件或恒流电路。
10、 根据权利要求 5所述的方法， 其特征在于： 通过在推挽变换器的输出端设 置同步整流电路， 使推挽振荡输出电压和输入电压线性同步。
PCT/CN2012/070254 2011-07-18 2012-01-12 一种自激推挽式变换器 WO2013010385A1 (zh)

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