WO2001082479A2 - Procede et appareil de filtrage de signaux dans un sous-systeme comprenant un amplificateur de puissance comportant un groupe d'appareils micromecaniques vibrants - Google Patents

Procede et appareil de filtrage de signaux dans un sous-systeme comprenant un amplificateur de puissance comportant un groupe d'appareils micromecaniques vibrants Download PDF

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WO2001082479A2
WO2001082479A2 PCT/US2001/040566 US0140566W WO0182479A2 WO 2001082479 A2 WO2001082479 A2 WO 2001082479A2 US 0140566 W US0140566 W US 0140566W WO 0182479 A2 WO0182479 A2 WO 0182479A2
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filters
micromechanical
bank
filter
resonator
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PCT/US2001/040566
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WO2001082479A3 (fr
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Clark T-C Nguyen
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The Regents Of The University Of Michigan
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Priority to CA002406223A priority Critical patent/CA2406223A1/fr
Priority to JP2001579451A priority patent/JP2003532323A/ja
Priority to EP01929082A priority patent/EP1275201A2/fr
Priority to AU2001255868A priority patent/AU2001255868A1/en
Publication of WO2001082479A2 publication Critical patent/WO2001082479A2/fr
Publication of WO2001082479A3 publication Critical patent/WO2001082479A3/fr

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/462—Microelectro-mechanical filters
    • H03H9/465—Microelectro-mechanical filters in combination with other electronic elements
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/02—Details
    • H03H9/02244—Details of microelectro-mechanical resonators
    • H03H9/02393—Post-fabrication trimming of parameters, e.g. resonance frequency, Q factor
    • H03H9/02409—Post-fabrication trimming of parameters, e.g. resonance frequency, Q factor by application of a DC-bias voltage
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/462—Microelectro-mechanical filters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/462—Microelectro-mechanical filters
    • H03H9/467—Post-fabrication trimming of parameters, e.g. center frequency
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/48—Coupling means therefor
    • H03H9/50—Mechanical coupling means
    • H03H9/505—Mechanical coupling means for microelectro-mechanical filters
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01H—ELECTRIC SWITCHES; RELAYS; SELECTORS; EMERGENCY PROTECTIVE DEVICES
    • H01H59/00—Electrostatic relays; Electro-adhesion relays
    • H01H59/0009—Electrostatic relays; Electro-adhesion relays making use of micromechanics
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/02—Details
    • H03H9/02244—Details of microelectro-mechanical resonators
    • H03H2009/02488—Vibration modes
    • H03H2009/02511—Vertical, i.e. perpendicular to the substrate plane
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/02—Details
    • H03H9/02244—Details of microelectro-mechanical resonators
    • H03H2009/02488—Vibration modes
    • H03H2009/02519—Torsional
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/02—Details
    • H03H9/02244—Details of microelectro-mechanical resonators
    • H03H2009/02488—Vibration modes
    • H03H2009/02527—Combined

Definitions

  • This invention relates to methods and apparatus for filtering signals in a subsystem including a power amplifier utilizing a bank of vibrating micromechanical apparatus.
  • FIG. 2 illustrates a comparison of MEMS and SAW technologies wherein MEMS offers the same or better high- ⁇ 2 frequency selectivity with orders of magnitude smaller size.
  • a single mechanical link does not possess adequate processing power for most applications.
  • the true power of ⁇ mechanical links can be unleashed, and signal processing functions with attributes previously inaccessible to transistor circuits may become feasible.
  • the stability of the oscillator signals used for frequency translation, synchronization, or sampling is of utmost importance.
  • Oscillator frequencies must be stable against variations in temperature against aging, and against any phenomena, such as noise or microphonics, that cause instantaneous fluctuations in phase and frequency.
  • the single most important parameter that dictates oscillator stability is the Q of the frequency-setting tank (or of the effective tank for the case of ring oscillators). For a given application, and assuming a finite power budget, adequate long- and short-term stability of the oscillation frequency is insured only when the tank Q exceeds a certain threshold value.
  • Tank Q also greatly influences the ability to implement extremely selective IF and RF filters with small percent bandwidth, small shape factor, and low insertion loss. As tank Q decreases, insertion loss increases very quickly, too much even for IF filters, and quite unacceptable for RF filters. As with oscillators, high- ⁇ 2 tanks are required for RF and IF filters alike, although more so for the latter, since channel selection is done predominantly at the IF in super-heterodyne receivers. In general, the more selective the filter, the higher the resonator Q required to achieve a given level of insertion loss.
  • U.S. Patent No. 6,049,702 to Tham et al. discloses an integrated passive transceiver section wherein microelectromechanical (MEM) device fabrication techniques are used to provide low loss, high performance switches. Utilizing the MEM devices also makes possible the fabrication and use of several circuits comprising passive components, thereby enhancing the performance characteristics of the transceiver.
  • MEM microelectromechanical
  • U.S. Patent No. 5,872,489 to Chang et al. discloses an integrated tunable inductance network and method.
  • the network utilizes a plurality of MEM switches which selectively interconnect inductance devices thereby providing a selective inductance for a particular circuit.
  • U.S. Patent No. 5,963,857 to Grey wall discloses an article comprising a micromachined filter.
  • the micromachined filters are assembled as part of a radio to miniaturize the size of the radio.
  • U.S. Patent Nos. 5,976,994 and 6,169,321 to Nguyen et al. disclose a batch-compatible, post-fabrication annealing method and system to trim the resonance frequency and enhance the quality factor of micromechanical structures.
  • U.S. Patent Nos. 5,455,547; 5,589,082 and 5,537,083 to Lin et al. disclose microelectromechanical signal processors.
  • the signal processors include many individual microelectromechanical resonators which enable the processor to function as a multi-channel signal processor or a spectrum analyzer.
  • U.S. Patent No. 5,640,133 to MacDonald et al. discloses a capacitance-based, tunable, micromechanical resonator.
  • the resonators may be selectively tuned and used in mechanical oscillators, accelero meters, electromechanical filters and other electronic devices.
  • U.S. Patent Nos. 5,578,976 to Yao, 5,619,061 to Goldsmith et al. and 6,016,092 to Qiu et al. disclose various micromechanical and microelectromechanical switches used in communication apparatus.
  • U.S. Patent No. 5,839,062 to Nguyen et al. disclose a MEMS-based receiver including parallel banks of microelectromechanical filters.
  • U.S. Patent No. 5,783,973 to Weinberg et al. discloses a micromechanical, thermally insensitive silicon resonator and oscillator.
  • An object of the present invention is to provide a method and apparatus for filtering signals in a subsystem including a power amplifier utilizing a bank of vibrating micromechanical apparatus to allow the power amplifier to operate with less power consumption.
  • a method for filtering signals to obtain at least one desired passband of frequencies in a subsystem including a power amplifier to allow the power amplifier to operate with less power consumption.
  • the method includes providing a bank of micromechanical apparatus. Each of the apparatus has at least one passband of frequencies.
  • the method also includes the step of controllably switching the bank of micromechanical apparatus to select the at least one desired passband and to substantially attenuate power outside the at least one desired passband.
  • the method includes the step of vibrating the micromechanical apparatus corresponding to the at least one desired passband.
  • the subsystem may be an RF transmitter subsystem and the at least one desired passband may be at least one desired channel.
  • Each of the apparatus may be switchable and tunable.
  • Each of the devices may have a Q greater than 10 and may be greater than 5000.
  • the number of devices may number more than 10 and may even number more than 1000.
  • the apparatus may be devices, filters, circuits or signal processors.
  • the power amplifier may be a non-linear or low power amplifier.
  • a signal filtering apparatus which filters signals to obtain at least one desired passband of frequencies and also to allow the power amplifier to operate with less power comsumption.
  • the apparatus includes a parallel bank of micromechanical filters. Each of the filters has a passband and is capable of handling amplified signals.
  • the apparatus also includes a controller for selecting at least one desired filter of the bank of filters to thereby provide the at least one desired passband of frequencies.
  • Each of the filters may have an input connected to a common input and each of the filters has an output connected to a common output.
  • the controller may include a decoder for controlling application of an appropriate bias voltage to the at least one desired filter.
  • the apparatus may be a high-Q, high-power filter.
  • Each of the filters may be switchable and tunable.
  • the present invention uses MEMS resonators/filters to condition the output/operation of a power amplifier so as to allow it to function with much higher efficiency (and thus, much lower power).
  • MEMS tanks can be used to remove high-order harmonics caused by nonlinearity around a power amplifier, thus allowing it to operate in a nonlinear fashion.
  • FIGURE 1 is a prior art, schematic view of the front-end of a transceiver including off-chip, board-level implementation of SAW, ceramic and crystal resonators in a schematic perspective view;
  • FIGURE 2a is a prior art, schematic perspective view of a SAW resonator and a number of MEMS resonators formed on a silicon die to compare the two approaches;
  • FIGURE 2b is a prior art, enlarged schematic perspective view of one of the MEMS resonators as indicated at 2b in Figure 2a;
  • FIGURE 3 is a system level schematic block diagram of the front-end design for a typical wireless transceiver showing off-chip, high- ⁇ , passive components targeted for replacement via micromechanical versions of the present invention
  • FIGURE 4 is a graph of transmission [dB] versus frequency illustrating desired filter characteristics
  • FIGURE 5a is a perspective schematic view of a symmetrical two- resonator VHF ⁇ mechanical filter with typical bias, excitation and signal conditioning electronics;
  • FIGURE 5b is an electrical equivalent circuit for the filter of Figure 5a;
  • FIGURE 6 is a system level block diagram of an RF front-end receiver including an RF channel-select receiver architecture utilizing large numbers of micromechanical resonators in banks and schematically and perspectively illustrating a typical micromechanical filter of Figure 5a;
  • FIGURE 7 is a system/circuit diagram for an RF channel-select micromechanical filter bank
  • FIGURE 8 is a system level block diagram of the RF front-end receiver of Figure 6 and schematically and perspectively illustrating a micromechanical switch thereof;
  • FIGURE 9 is a system level block diagram of the RF front-end receiver of Figures 6 and 8 and schematically and perspectively illustrating a mixer- filter-gain stage thereof based on the filter of Figures 6 and 5a;
  • FIGURE 10 is a system level block diagram of the RF front-end receiver of Figures 6, 8 and 9 and schematically and perspectively illustrating a micromechanical resonator oscillator thereof;
  • FIGURE 11 is a system/circuit diagram for a switchable ⁇ mechanical resonator synthesizer
  • FIGURE 12 is a system level block diagram of the RF front-end receiver of Figures 6, 8, 9 and 10 wherein the LNA is shown eliminated by phantom lines due to the low loss channel selector, the T/R switch and the mixer- filter-gain stage;
  • FIGURE 13 is a system block diagram architecture showing the receive path of a communication device
  • FIGURE 14 is a system block diagram of an RF channel-select transmitter architecture utilizing high-power ⁇ mechanical resonators
  • FIGURE 15 is a schematic top perspective view of a 92 MHz (VHF) free-free beam polysilicon ⁇ mechanical resonator wherein support beams isolate the resonator beam element from a substrate thereby allowing higher Q operation;
  • VHF 92 MHz
  • FIGURE 16 is a graph illustrating a measured frequency characteristic for the resonator of Figure 15;
  • FIGURE 17a is a schematic perspective view of a UHF ⁇ mechanical filter utilizing free-free beam ⁇ mechanical resonators designed to operate at a second mode
  • FIGURE 17b is a partial equivalent circuit for the filter of Figure 17a, identifying the circuit functions of individual beam elements.
  • Figure 3 presents a system level schematic block diagram for a typical super-heterodyne wireless transceiver.
  • a small box is positioned in the corner of each box to represent a component that can be replaced with a micromechanical (MEMS) version.
  • MEMS micromechanical
  • several of the constituent components can already be miniaturized using integrated circuit transistor technologies. These include the low noise amplifiers (LNA's) in the receive path, the solid-state power amplifier (SSPA) in the transmit path, synthesizer phase-locked loop (PLL) electronics, mixers, and lower frequency digital circuits for baseband signal demodulation.
  • LNA's low noise amplifiers
  • SSPA solid-state power amplifier
  • PLL synthesizer phase-locked loop
  • the SSPA (and sometimes the LNA's) are often implemented using compound semiconductor technologies (i.e. , GaAs). Thus, they often occupy their own chips, separate from the other mentioned transistor-based components, which are normally realized using silicon-based bipolar and CMOS technologies. However, given the rate of improvement of silicon technologies (silicon-germanium included), all of the above functions may be integrated onto a single-chip.
  • TDMA Time Division Multiple Access
  • TDMA Time Division Multiple Access
  • the requirement for small channel bandwidths results in a requirement for extremely selective filtering devices for channel selection and extremely stable (noise-free) local oscillators for frequency translation.
  • the required selectivity and stability can only be achieved using high-Q components, such as discrete inductors, discrete tunable capacitors (i.e. , varactors), and SAW and quartz crystal resonators, all of which interface with IC components at the board level.
  • the needed performance cannot be achieved using conventional IC technologies, because such technologies lack the required Q. It is for this reason that virtually all commercially available cellular or cordless phones contain numerous passive SAW and crystal components.
  • clamped-clamped and free-free flexural-mode beams with Q's on the order of 10,000 (in vacuum) and temperature coefficients on the order of -12 ppm/°C, are available for the VHF range, while thin-film bulk acoustic resonators ( ⁇ 2 ⁇ 1,000) have so far addressed the UHF range.
  • Anchor loss mechanisms can be greatly alleviated by using "anchorless" resonator designs, such as shown in the above-noted patent application Serial No. 09/482,670 and as illustrated by the view of Figure 15.
  • This device utilizes a free-free beam (i.e. , xylophone) 60 suspended above a ground plane and sense electrode 61 and a drive electrode 63 by four torsional support beams 62 attached at flexural node points. The beams 62, in turn, are supported at anchors 64.
  • the impedance presented to the beam 60 by the supports 62 can be effectively nulled out, leaving the beam 60 virtually levitated and free to vibrate as if it had no supports.
  • UHF frequency is obtained via use of free- free beam resonators specifically designed to operate at higher modes.
  • the desired mode is selected (while suppressing other modes), by strategic placement and excitation of electrodes, and by the use of dimples under the structure 60 to force node locations corresponding to the desired mode.
  • the mode can also be specified by placing the support beams at nodal locations, as in Figure 17a.
  • Figure 16 shows a frequency characteristic for a 92.25 MHz version of this ⁇ mechanical resonator with a Q of nearly 8,000 — still plenty for channel- select RF applications.
  • Table 1 presents expected resonance frequencies for various beam dimensions, modes, and structural materials, showing a wide range of attainable frequencies, from VHF to UHF.
  • Timoshenko methods that include the effects of finite h and W,.
  • ⁇ mechanical circuits for communications are those implementing low-loss bandpass filters, capable of achieving frequency characteristics as shown in Figure 4 where a broader frequency passband than achievable by a single resonator beam is shown, with a sharper roll-off to the stopband (i.e. , smaller shape factor).
  • a number of micromechanical resonators may be coupled together by soft coupling springs.
  • a coupled resonator system is achieved that exhibits several modes of vibration.
  • the frequency of each vibration mode corresponds to a distinct peak in the force-to-displacement frequency characteristic, and to a distinct, physical mode shape of the coupled mechanical resonator system.
  • each resonator in the lowest frequency mode, all resonators vibrate in phase; in the middle frequency mode, the center resonator ideally remains motionless, while the end resonators vibrate 180° out of phase; and finally, in the highest frequency mode, each resonator is phase- shifted 180° from its adjacent neighbor.
  • the complete mechanical filter exhibits a jagged passband.
  • termination resistors designed to lower the ⁇ 2's of the input and output resonators by specific amounts are required to flatten the passband and achieve a more recognizable filter characteristic, such as in Figure 4.
  • the filters use a number of high-O, micromechanical beam elements connected in a network that achieves the specified bandpass frequency response. If effect, a micromechanical filter is another example of a micromechanical circuit, similar to that of Figure 15, but in this case using a plurality of beam elements to achieve a frequency shaping response not achievable by a single beam element.
  • the constituent resonators in ⁇ mechanical filters are normally designed to be identical, with identical dimensions and resonance frequencies.
  • the center frequency of the overall filter is equal to the resonance frequency f 0 of the resonators, while the filter passband (i.e. , the bandwidth) is determined by the spacings between the mode peaks.
  • the relative placement of the vibration peaks in the frequency characteristic — and thus, the passband of the eventual filter — is determined primarily by the stiffnesses of the coupling springs (k sij ) and of the constituent resonators at the coupling locations (fe r ). Specifically, for a filter with center frequency f 0 and bandwidth B, these stiffnesses must satisfy the expression:
  • k ⁇ j is a normalized coupling coefficient found in filter cookbooks.
  • the filter bandwidth is not dependent on the absolute values of resonator and coupling beam stiffness; rather, their ratio k si k r dictates bandwidth.
  • the procedure for designing a mechanical filter involves two main steps (not necessarily in this order): first, design of a mechanical resonator with resonance frequency/-, and adjustable stiffness k r ; and second, design of coupling springs with appropriate values of stiffness k slJ to enable a desired bandwidth within the adjustment range of resonator
  • Figure 5a shows a perspective view schematic of a practical two- resonator micromechanical filter capable of operation in the HF to VHF range.
  • the filter consists of two ⁇ mechanical clamped-clamped beam resonators with anchors 18 at their opposite ends, coupled mechanically by a soft coupling spring or beam 19, all suspended above a substrate (not shown).
  • Conductive (polysilicon) strips 20, 22, 24, and 26 underlie each resonator by approximately 1000 A (as also in Figures 6, 9 and 17a), a center one 20 serving as a capacitive transducer input electrode positioned to induce resonator vibration in a direction perpendicular to the substrate, a center one 24 serving as an output electrode and the flanking ones 22 and 26 serving as tuning or frequency pulling electrodes capable of voltage-controlled tuning of resonator frequencies.
  • the resonator-to-electrode gaps are determined by the thickness of a sacrificial oxide spacer during fabrication and can thus be made quite small (e.g. , 0.1 ⁇ m or less) to maximize - electromechanical coupling.
  • the filter is excited with a DC-bias voltage V P applied to the conductive mechanical network, and an AC signal applied to the input electrode, but this time through an appropriately valued source resistance R ⁇ that loads the Q of the input resonator to flatten the passband.
  • the output resonator of the filter must also see a matched impedance to avoid passband distortion, and the output voltage v 0 is generally taken across this impedance.
  • the required value of I/O port termination resistance can be tailored for different applications, and this can be advantageous when designing low noise transistor circuits succeeding the filter, since such circuits can then be driven by optimum values of source resistance to minimize noise.
  • An electrical input signal is applied to the input port and converted to an input force by the electromechanical transducer (which for the case of Figure 5a is capacitive) that can then induce mechanical vibration in the x direction;
  • Mechanical vibration comprises a mechanical signal that is processed in the mechanical domain — specifically, the signal is rejected if outside the passband of the filter, and passed if within the passband; and (3) The mechanically processed signal appears as motion of the output resonator and is reconverted to electrical energy at the output transducer, ready for processing by subsequent transceiver stages.
  • micromechanical signal processor clearly suits this device. Details of the design procedure for micromechanical filters now follow.
  • the network topologies for the mechanical filters of this work differ very little from those of their purely electronic counterparts, and in principal, can be designed at the system-level via a procedure derived from well-known, coupled resonator ladder filter synthesis techniques.
  • a procedure derived from well-known, coupled resonator ladder filter synthesis techniques Given the equivalent LCR element values for a prototype ⁇ mechanical resonator, it is possible to synthesize a mechanical filter entirely in the electrical domain, converting to the mechanical domain only as the last step.
  • such a procedure is not recommended, since knowledge and ease of design in both electrical and mechanical domains can greatly reduce the effort required.
  • the coupling beam is recognized as an acoustic transmission line that can be made transparent to the filter when designed with quarter-wavelength dimensions as described in the prior art.
  • the filter bandwidth B is determined not by absolute values of stiffness, but rather by a ratio of stiffnesses (k sl2 /k rc ), where the subscript c denotes the value at the coupling location; and second, the value of resonator stiffness k rc varies with location (in particular, with location velocity) and so can be set to a desired value by simply choosing an appropriate coupling beam attachment point.
  • the location is easily determined as described in the prior art.
  • (4) Generate a complete equivalent circuit for the overall filter and verify the design using a circuit simulator.
  • Figure 5b presents the equivalent circuit for the two-resonator micromechanical filter of Figure 5a.
  • Each of the outside resonators are modeled via circuits.
  • the coupling beam actually operates as an acoustic transmission line, and thus, is modeled by a -T-network of energy storage elements.
  • Transformers are used between the resonator and coupling beam circuits of Figure 5b to model the velocity transformations that arise when attaching the coupling beams at locations offset from the center of the resonator beam.
  • the whole circuit structure of Figure 5b can be recognized as that of the LC ladder network for a bandpass filter.
  • Figure 17a is similar to Figure 5a and presents the schematic of a filter structure, along with a partial equivalent electrical circuit of Figure 17b (obtained via electromechanical analogy) that identifies the mechanical network as a bandpass filter.
  • the filter structure is seen to be comprised of a number of mechanical resonators (modeled by LCR tanks) connected by acoustic transmission lines (modeled by T-networks of energy storage elements) — a structure similar to other resonator-based filters, but using micromechanical elements with orders of magnitude higher Q, giving it the ability to perform with much lower insertion loss than other technologies. (Not to mention orders of magnitude smaller size.)
  • UHF frequency is obtained via use of a second mode free-free beam resonator including beams 70 and 72 coupled together and to an output beam 74 by coupling beams 73.
  • the resonator is specifically designed to operate at higher modes.
  • the desired mode is selected (while suppressing other modes), by strategic placement and excitation of balanced input electrodes 76 and 78, and by the use of dimples 80 under the beams 70, 72 and 74 to force node locations corresponding to the desired mode.
  • Balanced output electrodes 81 are positioned under the output beam 74.
  • the beams 70 and 74 are supported by non-intrusive supports or beams 82 and 84, respectively, which, in turn, are supported by anchors 86 and 88, respectively.
  • the filter of Figure 17a constitutes the first attempt of its kind to implement filter circuits using free-free beam micromechanical resonators. Unlike previous filters using clamped-clamped beams (that could only attain VHF frequency), both transversal and torsional motions are considered for the coupling beams in Figure 17a.
  • Figure 9 presents the schematic for a symmetrical ⁇ mechanical mixer-filter, showing the bias and input scheme required for down-conversion with a gain stage.
  • this device provides filtering as part of its function, the overall mechanical structure is exactly that of a ⁇ mechanical filter. The only differences are the applied inputs and the use of a non-conductive coupling beam to isolate the IF port from the LO. If the source providing V P to the second resonator is ideal (with zero source resistance) and the series resistance in the second resonator is small, LO signals feeding across the coupling beam capacitance are shunted to AC ground before reaching the IF port. In reality, finite resistivity in the resonator material allows some amount of LO-to-IF leakage.
  • the mixer conversion gain/loss in this device is determined primarily by the relative magnitudes of the DC-bias V P applied to the resonator and the local oscillator amplitude V LO .
  • the mixer-filter device described above is one example of a ⁇ mechanical circuit that harnesses non-linear device properties to provide a useful function.
  • Another very useful mode of operation that further utilizes the non-linear nature of the device is a ⁇ mechanical switch.
  • Figure 8 presents an operational schematic for a ⁇ mechanical switch.
  • a conductive or actuation plate 30 is suspended above a pair of actuation electrodes 32 by suspension beams 34 having anchors 36.
  • a switch conductor portion 38 of the plate 30 is suspended over a pair of grounds 40 and a sense electrode or conductor 42. When the switch is in the "on-state" here, the conductor 42 is shorted to the grounds 40.
  • switches In general, to minimize insertion loss, the majority of switches use metals as their structural materials. It is their metal construction that makes ⁇ mechanical switches so attractive, allowing them to achieve "on-state" insertion losses down to 0.1 dB — much lower than FET transistor counterparts, which normally exhibit ⁇ 2 dB of insertion loss. In addition to exhibiting such low insertion loss, ⁇ mechanical switches are extremely linear, with IIP3's greater than 66 dBm, and can be designed to consume no DC power (as opposed to FET switches, which sink a finite current when activated).
  • Figure 3 shows the use of other MEMS-based passive components, such as medium- ⁇ 2 micromachined inductors and tunable capacitors used in VCO's and matching networks, as well as low-loss (-0.1 dB) ⁇ mechanical switches that not only provide enhanced antenna diversity, but that can also yield power savings by making TDD (rather than FDD) more practical in future transceivers.
  • MEMS-based passive components such as medium- ⁇ 2 micromachined inductors and tunable capacitors used in VCO's and matching networks, as well as low-loss (-0.1 dB) ⁇ mechanical switches that not only provide enhanced antenna diversity, but that can also yield power savings by making TDD (rather than FDD) more practical in future transceivers.
  • ⁇ mechanical circuits offer the same system complexity advantages over off-chip discrete components that planar IC circuits offer over discrete transistor circuits.
  • ⁇ mechanical circuits should be utilized on a massive scale, or at least as much as possible.
  • Figure 6 presents the system-level block diagram for a possible receiver front-end architecture that takes full advantage of the complexity achievable via ⁇ mechanical circuits, such as the micromechanical filter of Figure 5a.
  • the main driving force behind this architecture is power reduction, attained in several of the blocks by trading power for high selectivity (i. e. , high- .
  • the key power saving blocks in Figure 6 are now described.
  • channel selection (rather than pre-selection) were possible at RF frequencies (rather than just an IF)
  • succeeding electronic blocks in the receive path e.g. , LNA, mixer
  • LNA local oscillator
  • Figure 7 presents one fairly simple rendition of the key system block that realizes the desired RF channel selection.
  • this block consists of a bank of ⁇ mechanical filters with all filter inputs connected to a common block input and all outputs to a common block output, and where each filter passband corresponds to a single channel in the standard of interest.
  • each filter passband corresponds to a single channel in the standard of interest.
  • it does not have to be limited to a single channel. It could also be several channels as well (e.g. , 3 channels) and this could still be very advantageous, depending upon the communications standard.
  • a given filter is switched on (with all others off) by decoder-controlled application of an appropriate DC-bias voltage to the desired filter.
  • the desired force input and output current are generated in a ⁇ mechanical resonator only when a DC-bias V P is applied (i.e. , without V P , the input and output electrodes are effectively open-circuited).
  • this RF channel selector can be quantified by assessing its impact on the LNA linearity specification imposed by the IS-98-A interim standard for CDMA cellular mobile stations.
  • the required IIP3 of the LNA is set mainly to avoid desensitization in the presence of a single tone (generated by AMPS) spaced 900 kHz away from the CDMA signal center frequency.
  • AMPS AMPS-generated single tone spaced 900 kHz away from the CDMA signal center frequency.
  • reciprocal mixing of the local oscillator phase noise with the 900 kHz offset single tone and cross-modulation of the single tone with leaked transmitter power outputs dictate that the LNA IIP3 exceeds +7.6 dBm.
  • the power requirement in the sustaining amplifier might be dictated solely by loop gain needs (rather than by phase noise needs), which for a ⁇ mechanical resonator-based VCO with R X ⁇ 40 ⁇ , L x ⁇ 84 ⁇ H, and C x ⁇ 0.5fF, might be less than 1 mW.
  • a switchable bank similar to that of Figure 7 but using ⁇ mechanical resonators 46, not filters, each corresponding to one of the needed LO frequencies, and each switchable into or out of an oscillator sustaining circuit 48 by transistor switches 49 at their electrodes 51 is illustrated in Figure 11 and is preferred over the tuning-fork- resonator oscillator 50 illustrated in Figure 10. Because ⁇ mechanical resonators are now used in this implementation, the Q and thermal stability of the oscillator may now be sufficient to operate without the need for locking to a lower frequency crystal reference. The power savings attained upon removing the PLL and prescaler electronics needed in past synthesizers can obviously be quite substantial.
  • synthesizer power consumption can be reduced from the ⁇ 90 mW dissipated by present-day implementations using medium-Q L and C components, to something in the range of only 1-4 mW. Again, all this is attained using a circuit topology that would seem unreasonable if only macroscopic high-Q resonators were available, but that becomes plausible in the micromechanical arena.
  • Figure 11 presents the basic topology of the LO synthesizer.
  • a bank of numerous switchable high-O, (Q> 5,000) micromechanical resonators is utilized, where each resonator corresponds to one of the needed frequencies in a given communications network.
  • Q> 5,000 micromechanical resonators is utilized, where each resonator corresponds to one of the needed frequencies in a given communications network.
  • no phase-locking circuit is required, since the local oscillator merely switches the appropriate resonator into the oscillator feedback loop to generate the needed output frequency.
  • the Q of the micromechanical resonators are orders of magnitude higher than before ( ⁇ 2 ⁇ 5,000, as opposed to 40),the phase noise performance of this oscillator should be orders of magnitude better than present-day VCO's.
  • the l// 2 -to-white noise corner frequency of the standard phase noise vs. frequency plot may be so close to the carrier, that l// 2 noise may no longer be a major consideration, and only white noise is present at the important frequency offsets.
  • power could be traded for Q to such an extent that it may become possible to operate the oscillator with less than 1 mW of power consumption. Combined with the fact that phase-locking is no longer required in the scheme of Figure 11, this constitutes more than 90 mW of power savings, while improving the performance of the oscillator.
  • a single-chip LO synthesizer, using the architecture of Figure 11 can achieve impressive phase noise performance at UHF frequencies (-800 MHz). By using higher-mode, free-free beam resonator designs previously described, UHF (and possibly S-Band or higher) frequency synthesizers are feasible. K- or Ka-Band applications are not unreasonable, and should benefit from the architecture of Figure 11 either directly as shown, or indirectly, since a UHF high- ⁇ reference oscillator should greatly improve the stability of K- or Ka-band synthesizers.
  • the use of a ⁇ mechanical mixer-filter in the receive path as illustrated in Figure 9 eliminates the DC power consumption associated with the active mixer normally used in present-day receive architectures.
  • the mixer-filter-gain stage includes a pair of n-type resonators 21 ,23 coupled together by a p-type or undoped beam or spring 19 similar to filters of Figures 5a and 6.
  • FIG. 13 depicts a receive path comprised of a relatively wideband image reject ⁇ mechanical RF filter followed immediately by a narrowband IF mixer-filter that then feeds subsequent IF electronics.
  • the only active electronics operating at RF in this system are those associated with the local oscillator, which if it uses a bank of ⁇ mechanical resonators as previously described, may be able to operate at less than 1 mW.
  • Figure 14 depicts one rendition, in which an RF channel selector is placed after the power amplifier (PA) in the transmit path.
  • This channel selector uses a similar circuit as that of Figure 7, but using ⁇ mechanical resonators with sufficient power handling capability.
  • This transmit topology provides enormous power savings.
  • the high-O,, high-power filter with less than 1 dB of insertion loss follows the PA, cleaning all spurious outputs, including those arising from spectral regrowth. Consequently, more efficient PA designs can be utilized, despite their non-linearity.
  • a PA previously restricted by linearity considerations to 30% efficiency in present-day transmitter architectures may now be operable closer to its maximum efficiency, perhaps 50% . For a typical transmit power of 600 mW, this efficiency increase corresponds to 800 mW of power savings.
  • a more efficient PA topology could be used, such as Class E, with theoretical efficiencies approaching 100%, the power savings could be much larger.
  • the architecture of Figure 14 also features a micromechanical upconverter that uses a mixer-filter device, such as previously described, to upconvert and filter the information signal before directing it to the power amplifier.
  • the baseband signal at frequency ⁇ IF to be upconverted is applied to the input electrode 20 and the upconverting carrier signal at frequency ⁇ LO is applied to the input resonator device 21.
  • Upconversion occurs through mixing around the nonlinear capacitive transducer between the input electrode 20 and the resonator 21.
  • the electrical baseband information signal at frequency ⁇ IF is upconverted to a force at frequency ⁇ L0 + ⁇ IF .
  • This force is then filtered by the filter structure of Figure 9, which is now designed to have a passband around ⁇ LO + ⁇ IF .
  • this passband is made small enough, channel selection to the point of removing not only distortion harmonics, but also spectrally regrown components, is possible.
  • V p dc-bias V p
  • the displacement of this resonator is converted to an electrical output voltage or current, depending upon the output load. It should be noted, also, that gain is possible in upconverting the baseband signal to the RF signal, so this stage also serves as a gain stage, as well.
  • the second method for upconversion involves filtering the based band signal first, then upconverting.
  • the baseband signal is again applied to the input electrode 20, but the dc-bias is applied to the input resonator, and the carrier signal to the output resonator.
  • the baseband signal is first filtered by the structure, then upconverted at the output via electromechanical amplitude modulation. Again, gain is possible in this configuration.
  • High power handling micromechanical resonators may use alternative geometries (e.g. , no longer flexural mode) and the use of alternative transduction (e.g. , piezoelectric, magneto strictive).
  • Vibrating ⁇ mechanical resonators constitute the building blocks for a new integrated mechanical circuit technology in which high Q serves as a principal design parameter that enables more complex circuits.
  • high Q serves as a principal design parameter that enables more complex circuits.

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  • Physics & Mathematics (AREA)
  • Acoustics & Sound (AREA)
  • Piezo-Electric Or Mechanical Vibrators, Or Delay Or Filter Circuits (AREA)
  • Micromachines (AREA)
  • Transceivers (AREA)
  • Transmitters (AREA)
  • Oscillators With Electromechanical Resonators (AREA)
  • Amplifiers (AREA)
  • Measurement Of Velocity Or Position Using Acoustic Or Ultrasonic Waves (AREA)
  • Input Circuits Of Receivers And Coupling Of Receivers And Audio Equipment (AREA)
  • Filters And Equalizers (AREA)
  • Networks Using Active Elements (AREA)

Abstract

La présente invention concerne plusieurs procédés et architectures fondés sur des systèmes microélectromécaniques (MEMS) dans lesquels on utilise des résonateurs micromécaniques vibrants dans des circuits pour effectuer des fonctions de filtrage, de mélange, de référence de fréquence et d'amplification. Un appareil est prévu pour sélectionner au moins une bande passante ou une voie désirée dans un sous-système d'émetteur de fréquence radioélectrique utilisant un groupe de dispositifs micromécaniques vibrants. Un des avantages principaux résultant de l'utilisation de telles architectures est lié à une diminution de la consommation d'énergie qui entraîne une sélectivité élevée (c'est-à-dire un facteur de qualité (Q) élevé). Par conséquent, la présente invention repose sur l'utilisation d'un grand nombre de liaisons micromécaniques dans des réseaux SSI pour assurer des fonctions de traitement du signal avec une consommation de puissance CC pratiquement nulle.
PCT/US2001/040566 2000-04-20 2001-04-20 Procede et appareil de filtrage de signaux dans un sous-systeme comprenant un amplificateur de puissance comportant un groupe d'appareils micromecaniques vibrants WO2001082479A2 (fr)

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Application Number Priority Date Filing Date Title
CA002406223A CA2406223A1 (fr) 2000-04-20 2001-04-20 Procede et appareil de filtrage de signaux dans un sous-systeme comprenant un amplificateur de puissance comportant un groupe d'appareils micromecaniques vibrants
JP2001579451A JP2003532323A (ja) 2000-04-20 2001-04-20 振動マイクロメカニカル装置列を利用して電力増幅器を有するサブシステムの信号を濾過するための方法及び装置
EP01929082A EP1275201A2 (fr) 2000-04-20 2001-04-20 Procede et appareil de filtrage de signaux dans un sous-systeme comprenant un amplificateur de puissance comportant un groupe d'appareils micromecaniques vibrants
AU2001255868A AU2001255868A1 (en) 2000-04-20 2001-04-20 Method and apparatus for filtering signals in a subsystem including a power amplifier utilizing a bank of vibrating micromechanical apparatus

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US19906300P 2000-04-20 2000-04-20
US60/199,063 2000-04-20

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PCT/US2001/040566 WO2001082479A2 (fr) 2000-04-20 2001-04-20 Procede et appareil de filtrage de signaux dans un sous-systeme comprenant un amplificateur de puissance comportant un groupe d'appareils micromecaniques vibrants
PCT/US2001/012806 WO2001082475A2 (fr) 2000-04-20 2001-04-20 Procede et appareil pour la conversion-elevation et le filtrage d'un signal d'information au moyen d'un dispositif micromecanique vibrant
PCT/US2001/040562 WO2001082476A2 (fr) 2000-04-20 2001-04-20 Procede et appareil pour selectionner au moins un canal desire en utilisant une banque d'un appareil micromecanique vibratoire
PCT/US2001/040564 WO2001082477A2 (fr) 2000-04-20 2001-04-20 Procede et dispositif de filtrage de signaux utilisant un resonateur micromecanique vibrant
PCT/US2001/040565 WO2001082478A2 (fr) 2000-04-20 2001-04-20 Procede et sous-systeme de traitement de signaux dans lesquels on utilise une pluralite de dispositifs micromecaniques vibrants
PCT/US2001/040563 WO2001082467A2 (fr) 2000-04-20 2001-04-20 Procede et appareil employes pour generer un signal ayant au moins une frequence de sortie desiree au moyen d'un groupe de dispositifs micromecaniques vibrants

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PCT/US2001/040562 WO2001082476A2 (fr) 2000-04-20 2001-04-20 Procede et appareil pour selectionner au moins un canal desire en utilisant une banque d'un appareil micromecanique vibratoire
PCT/US2001/040564 WO2001082477A2 (fr) 2000-04-20 2001-04-20 Procede et dispositif de filtrage de signaux utilisant un resonateur micromecanique vibrant
PCT/US2001/040565 WO2001082478A2 (fr) 2000-04-20 2001-04-20 Procede et sous-systeme de traitement de signaux dans lesquels on utilise une pluralite de dispositifs micromecaniques vibrants
PCT/US2001/040563 WO2001082467A2 (fr) 2000-04-20 2001-04-20 Procede et appareil employes pour generer un signal ayant au moins une frequence de sortie desiree au moyen d'un groupe de dispositifs micromecaniques vibrants

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EP1475856A1 (fr) * 2002-02-13 2004-11-10 Matsushita Electric Industrial Co., Ltd. Filtre contre les vibrations d'une micromachine
WO2005029700A1 (fr) * 2003-09-19 2005-03-31 Sony Corporation Resonateur de microsysteme electromecanique, procede de commande de ce resonateur, procede de fabrication de ce dernier et filtre
JPWO2004032320A1 (ja) * 2002-10-03 2006-02-02 シャープ株式会社 マイクロ共振装置、マイクロフィルタ装置、マイクロ発振器および無線通信機器
EP1706942A4 (fr) * 2004-01-09 2008-05-21 Bosch Gmbh Robert Oscillateur microelectromecanique a compensation de frequence et/ou de phase
DE10252828B4 (de) * 2002-11-13 2018-07-05 Epcos Ag Mit akustischen Wellen arbeitendes Bauelement und Verfahren zur Herstellung eines Bauelements

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JP4586404B2 (ja) 2004-04-28 2010-11-24 ソニー株式会社 フィルタ装置及び送受信機
JP4470606B2 (ja) * 2004-06-18 2010-06-02 ソニー株式会社 高周波素子、並びに通信装置
JP4608984B2 (ja) * 2004-07-21 2011-01-12 ソニー株式会社 微小共振器およびその製造方法、ならびに電子機器
JP2006041911A (ja) * 2004-07-27 2006-02-09 Matsushita Electric Ind Co Ltd Ｍｅｍｓフィルタ装置およびその製造方法
JP4635619B2 (ja) * 2005-01-20 2011-02-23 ソニー株式会社 微小共振器及び通信装置
JP4604730B2 (ja) * 2005-01-20 2011-01-05 ソニー株式会社 微小振動子、半導体装置及び通信装置
JP4617904B2 (ja) * 2005-02-01 2011-01-26 ソニー株式会社 微小振動子、半導体装置及び通信装置
JP4645227B2 (ja) * 2005-02-28 2011-03-09 セイコーエプソン株式会社 振動子構造体及びその製造方法
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JP2007150736A (ja) 2005-11-28 2007-06-14 Sony Corp 微小電気機械デバイス
JP2007174438A (ja) * 2005-12-23 2007-07-05 Toshiba Corp フィルタ回路及びフィルタを備えた無線通信システム
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EP1475856A1 (fr) * 2002-02-13 2004-11-10 Matsushita Electric Industrial Co., Ltd. Filtre contre les vibrations d'une micromachine
EP1475856A4 (fr) * 2002-02-13 2005-03-23 Matsushita Electric Ind Co Ltd Filtre contre les vibrations d'une micromachine
US6995633B2 (en) 2002-02-13 2006-02-07 Matsushita Electric Industrial Co., Ltd. Micromachine vibration filter
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JPWO2004032320A1 (ja) * 2002-10-03 2006-02-02 シャープ株式会社 マイクロ共振装置、マイクロフィルタ装置、マイクロ発振器および無線通信機器
DE10252828B4 (de) * 2002-11-13 2018-07-05 Epcos Ag Mit akustischen Wellen arbeitendes Bauelement und Verfahren zur Herstellung eines Bauelements
WO2005029700A1 (fr) * 2003-09-19 2005-03-31 Sony Corporation Resonateur de microsysteme electromecanique, procede de commande de ce resonateur, procede de fabrication de ce dernier et filtre
EP1706942A4 (fr) * 2004-01-09 2008-05-21 Bosch Gmbh Robert Oscillateur microelectromecanique a compensation de frequence et/ou de phase
US7453324B2 (en) 2004-01-09 2008-11-18 Robert Bosch Gmbh Frequency and/or phase compensated microelectromechanical oscillator
US7532081B2 (en) 2004-01-09 2009-05-12 Robert Bosch Gmbh Frequency and/or phase compensated microelectromechanical oscillator
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WO2001082478A3 (fr) 2002-04-11
WO2001082477A3 (fr) 2002-04-11
WO2001082467A2 (fr) 2001-11-01
AU2001255868A1 (en) 2001-11-07
AU2001255866A1 (en) 2001-11-07
EP1290788A2 (fr) 2003-03-12
WO2001082476A3 (fr) 2002-08-15
WO2001082478A2 (fr) 2001-11-01
EP1285491A2 (fr) 2003-02-26
WO2001082475A3 (fr) 2002-02-21
CA2406223A1 (fr) 2001-11-01
JP2003532323A (ja) 2003-10-28
AU2001255865A1 (en) 2001-11-07
WO2001082467A3 (fr) 2002-04-11
AU2001255867A1 (en) 2001-11-07
WO2001082476A2 (fr) 2001-11-01
AU2001257612A1 (en) 2001-11-07
JP2003532320A (ja) 2003-10-28
WO2001082475A2 (fr) 2001-11-01
AU2001261036A1 (en) 2001-11-07
CA2406176A1 (fr) 2001-11-01
EP1277277A2 (fr) 2003-01-22
WO2001082479A3 (fr) 2002-04-11
JP2004515089A (ja) 2004-05-20
JP2003532322A (ja) 2003-10-28
WO2001082477A2 (fr) 2001-11-01
CA2406518A1 (fr) 2001-11-01
CA2406543A1 (fr) 2001-11-01
EP1275201A2 (fr) 2003-01-15

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