EP0010502A1 - Variable Induktivität - Google Patents

Variable Induktivität Download PDF

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Publication number
EP0010502A1
EP0010502A1 EP79400766A EP79400766A EP0010502A1 EP 0010502 A1 EP0010502 A1 EP 0010502A1 EP 79400766 A EP79400766 A EP 79400766A EP 79400766 A EP79400766 A EP 79400766A EP 0010502 A1 EP0010502 A1 EP 0010502A1
Authority
EP
European Patent Office
Prior art keywords
magnetic
phase
variable
magnetic field
control
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Granted
Application number
EP79400766A
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English (en)
French (fr)
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EP0010502B1 (de
Inventor
Gérald Roberge
André Doyon
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Hydro Quebec
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Hydro Quebec
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Filing date
Publication date
Application filed by Hydro Quebec filed Critical Hydro Quebec
Priority to DE8383111475T priority Critical patent/DE2967595D1/de
Priority to DE8383111087T priority patent/DE2967589D1/de
Publication of EP0010502A1 publication Critical patent/EP0010502A1/de
Application granted granted Critical
Publication of EP0010502B1 publication Critical patent/EP0010502B1/de
Expired legal-status Critical Current

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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
    • H01F21/00—Variable inductances or transformers of the signal type
    • H01F21/02—Variable inductances or transformers of the signal type continuously variable, e.g. variometers
    • H01F21/08—Variable inductances or transformers of the signal type continuously variable, e.g. variometers by varying the permeability of the core, e.g. by varying magnetic bias
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
    • H01F29/00—Variable transformers or inductances not covered by group H01F21/00
    • H01F29/14—Variable transformers or inductances not covered by group H01F21/00 with variable magnetic bias
    • H01F29/146—Constructional details
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
    • H01F29/00—Variable transformers or inductances not covered by group H01F21/00
    • H01F29/14—Variable transformers or inductances not covered by group H01F21/00 with variable magnetic bias
    • H01F2029/143—Variable transformers or inductances not covered by group H01F21/00 with variable magnetic bias with control winding for generating magnetic bias

Definitions

  • the present invention relates to a variable inductance device and more specifically relates to a device, the effective permeability of which is controlled by a closed magnetic circuit through which a magnetic flux with constant and adjustable current flows.
  • variable inductance device or “variable inductance” will be used interchangeably.
  • One of the aims of the present invention is to avoid the drawbacks mentioned above, relating to known devices, and aims to provide an inductance with a low level of harmonics by appropriate control of its permeability or reluctance.
  • the present invention relates to a variable inductance which comprises a first magnetic circuit closed tick, formed of an anisotropic material through which an alternating magnetic field circulates; a second closed magnetic circuit, also formed of an anisotropic material, through which a magnetic flow with adjustable direct current flows; the first and second magnetic circuits being arranged with respect to each other so as to define at least two common magnetic spaces in which the respective alternating and continuous magnetic fields are superposed orthogonally to orient the magnetic dipoles of said common spaces in a direction determined by the intensity of said magnetic flux from the second circuit and thereby controlling the permeability of said first magnetic circuit to said alternating field.
  • FIG. 1 illustrates an embodiment of a single-phase variable inductor made up of two magnetic circuits M and N arranged orthogonally.
  • the magnetic circuit M is formed of a core in two parts Ml and M2 connected by junction zones Dl and D2 belonging to the magnetic core N and subsequently called "common magnetic spaces".
  • This magnetic circuit M is excited by an alternating current winding disposed between the terminals Pl, P2 which extends over the two parts Ml and M2 of the magnetic core M.
  • the magnetic circuit N consists of a core single through which circulates a magnetic field excited by a direct current winding disposed between the terminals C1, C2.
  • the orthogonal arrangement of the two magnetic circuits has the effect of producing in the common magnetic spaces D1 and D2 a magnetic torque proportional to the value, in the core N, of the direct current magnetic field, which polarizes the dipoles of these common magnetic spaces. Because of this orthogonal arrangement, the respective magnetic fluxes of the two nuclei cannot take the same path; the direct current magnetic field orients, by polarizing them, the magnetic dipoles of the common magnetic spaces so as to act on the permeability of the magnetic circuit excited by the alternating current winding. native as desired.
  • the cores M and N are made of ferro-magnetic materials of the same cross section, either ferrite or in rolled iron, and therefore have a property inherent.
  • the dipoles of the common spaces Dl and D2 in the absence of a DC polarizing field N, tend to orient in the direction of the alternating magnetic field, the permeability of the nucleus M then being a measure of the ease with which the magnetic dipoles are oriented in the direction of this exciting field.
  • the nucleus M becomes saturated when its dipoles are completely oriented in the direction of this magnetic field.
  • the application of a direct current magnetic field in the core N in a direction transverse to the alternating magnetic field of the core M has the effect of acting on the dipoles of the common magnetic spaces Dl D2, by polarizing them, for move them away from their equilibrium position, so that the alternating magnetic field of the nucleus M must grow in modulus so that each dipole maintains its same equilibrium position in the common magnetic spaces D1 and D2.
  • This process does not affect the leakage inductance in any way, but only the magnetization inductance of the variable inductance core.
  • the magnetic saturation induction is increased and the magnetization curves become more linear with the increase in the direct current magnetic field in the common spaces D1 and D2. Consequently, the application of a direct current magnetic field perpendicular to an alternating magnetic field produces a variable gap effect for the alternating magnetic circuit.
  • the contact surfaces between the magnetic circuits M and N are machined and mechanically clamped one on the other or are produced according to any other equivalent mounting method, while the DC winding Cl and C2 is supplied by an auxiliary source with constant and adjustable direct current.
  • a secondary winding Sl, S2 superimposed on the primary winding Pl, P2 makes it possible to filter the harmonics of zero sequence components and, moreover, to connect this core with variable inductance to a circuit of useu
  • This single-phase variable inductance device therefore essentially consists in producing in common magnetic spaces a direct current magnetic field which has the effect of opposing rotation dipoles of these common spaces for an adequate control of the effective permeability of the alternating magnetic circuit. It is clear that the common magnetic spaces can be established as well in the nucleus of phase M as in the nucleus of control N, as Ci- above described.
  • FIG. 2 illustrates a self-checking connection of the single-phase device of FIG. 1 by insertion of a diode bridge R at full alternation between the alternating winding Pl, P2 and the continuous winding Cl, C2 of the device.
  • This arrangement makes it possible to continuously vary the permeability of the core M as a function of sudden variations in the alternating magnetic flux.
  • FIG. 2 allows a three-phase use of the variable inductance of FIG. 1.
  • the secondary winding S1, S2 is connected in delta with the other two phases so as to filter the third components and ninth harmonics of the alternating magnetic flux.
  • the primary windings Pl and P2 are then connected in a star with floating neutral. In this case, the excitation windings of the three phases can be connected either in series or in parallel.
  • the number of turns of the alternating excitation coil can be modified using thyristors T slaved to a voltage setpoint, which has the effect of shifting the curve of the operating point of the inductor.
  • the response time of the variable inductance circuit when it is in self-control, is almost instantaneous, that is to say that the response time will be less than a period.
  • the regulation control time it may vary depending on the control mode used and reach one or two periods (based on 60 Hertz) depending on the needs of the user.
  • the eddy current and hysteris losses are considerably reduced by using ferrite to form the magnetic DC circuit N.
  • the geometry of the circuit, the type of core used; the length of the magnetic circuit are all factors that reduce losses.
  • FIG. 3 gives the ranges and operating points of the single-phase variable inductor when used in self-control, as illustrated in FIG. 2.
  • the current I in the inductor is indicated on the abscissa (c ' that is to say in the circuit Pl-P2) and on the ordinate the phase-> neutral voltage U ⁇ + N (one of the terminals Pl-P2 being neutral).
  • curve 1 in dotted line is a magnetization curve of the alternating current core M in closed circuit and in the absence of any control core N whereas the dashed line curve 2 corresponds to the magnetization obtained when the common magnetic spaces are replaced by a piece of wood of equivalent thickness.
  • a voltage rise range for an AC voltage across the inductor varying from 0 to a little beyond the knee of the curve, range in which the the slope of each of the operating point curves is particularly large
  • FIG 4 presents a three-phase model of the variable inductance.
  • Each of the phases, PA, PB, and PC are respectively connected to the cores MA, MB and MC of the same cross section through each of which circulates an alternating magnetic field of corresponding phase.
  • Each core has a branch mounted orthogonally to the control core N, the winding El-E2 of which is excited by a source of constant but adjustable direct current.
  • the control circuit being common to the three phases, it is noted that there is cancellation of the voltages induced at 120 Hz in the DC control coil N, as in the previous single-phase model, and that there is no alternative flow in this continuous flow core, except in the regions of the common spaces D3, D4 and D5.
  • the phases of the cores MA, MB and MC are not arranged symmetrically so that this circuit is not optimal as regards the length of the phase cores, their junctions and their geometric arrangement with respect to to the control nucleus N.
  • FIG. 5 illustrates a symmetrical arrangement of the three-phase variable inductance in which the phase cores MA, MB and MC form an angle of 120 ° relative to each other and are mechanically mounted on the control core N which is hexagonal in shape.
  • This arrangement of FIG. 4 allows a range of variations of the impedance in the same order of magnitude as in the previous case e an appreciable reduction in the relative losses, therefore an increase in the quality factor of the inductance.
  • This type of construction does not show magnetic legs for the return of the flow in transient regime.
  • FIGS. 4 and 5 allows elimination of the third and ninth harmonic currents by means of a star connection of the three phases PA, PB and PC, with floating neutral, not connected to ground, and the elimination of fluxes.
  • third and ninth harmonics using a superimposed secondary winding, PSA, PSB and PSC, connected in a triangle.
  • the losses in the control core N are considerably reduced due to the fact that no bidirectional reaction remains between the control core and the phase nuclei, since there is no alternating magnetic flux in the core of control N, the sum of the effects of the three phases being zero.
  • the neutral of the star connection being possible with homo- components establish a transitional regime.
  • variable inductor of Figures 4 and 5 When used in three-phase, the arrangement of the variable inductor of Figures 4 and 5 has an increased advantage over the use of three single-phase inductors of the type shown in Figure 2 due to the fact that the same amount control energy is required for all three phases than that which would be required for a single phase, so that the losses of control are less and distributed over the three phases.
  • control of the direct current magnetic flux can be carried out by self-control, using diode bridges, as in the case of the single-phase inductor of FIG. 2, or even by reverse control using a constant and adjustable direct current winding, superimposed on the self-checking winding, on the control core N.
  • FIG. 6 shows the variations in impedance of the three-phase inductance as a function of the increase in ampere-turns injected into the control core N.
  • the impedances V / I of each phase vary in a ratio of up to 11/1 for a direct current magnetic field varying from 0 to 4,848 ampere-turns .
  • impedances varying in a ratio of 20/1 for a rolled steel model and 25/1 for a ferrite model have been obtained.
  • the dotted line 1 shows the behavior of the variable inductor for a voltage of 80 volts rms measured phase-neutral.
  • the dotted line 2 shows the behavior of the variable inductance when it is connected in series with a capacitor and the resultant of which is inductive.
  • the value of the capacity used was 200 uF and the three-phase source was kept fixed at 120 volts rms across the circuit.
  • the increase in volts-a.amperes of the variable inductance for a displacement from A to B on the curves is 360 volts-amperes. three-phase for 4,848 ampere-turns. This increase in power is approximately 1.78 times greater than for the case of the inductor alone for the same voltage.
  • Figure 7 shows a family of saturation curves.
  • the alternating current IcA has been plotted on the ordinate as an effective value, the ampere-turns of the direct current control as abscissa, and the phase-neutral voltages as an effective value as a curve parameter.
  • This figure 7 provides information on the behavior of dipoles in the magnetic space common to the two magnetic circuits.
  • Figures 8, 9, 10 and 11 respectively show the harmonics ratio of the third, fifth, seventh and ninth harmonics current as a function of the ampere-turns with direct current. These harmonic rates are calculated between the harmonic considered and the Fundamental for a full load alternating current which corresponds to 5.0 (X 606) ampere-turns with direct current.
  • the harmonic rates calculated for only one phase of the three-phase inductance in Figure 4, are very low and even negligible for some harmonics.
  • curves 1, 2, 3 and 4 corresponding to tests carried out under voltages, in effective values, of 80 volts, 160 volts, 200 volts, and 280 volts, respectively.
  • the asymmetrical arrangement of the magnetic circuits in Figure 4 plays an important role in this phenomenon. Indeed, the control nucleus N is oval and the phase nuclei are not arranged at 120 ° relative to each other on this control nucleus. Improved results are obtained with the three-phase inductors of Figures 15 and 16 in which the phase cores are well arranged at 120 ° relative to each other and where the control core is cylindrical.
  • FIG. 12 presents curves of distortion of the phase-neutral voltage of 180 volts in rms value as a function of the harmonics generated by a phase of the three-phase inductance.
  • Curve 1 gives results measured for the network alone while curves 2 and 3 illustrate the results obtained when the variable inductor is connected to the network and where the control flow is respectively zero and equal to 1.212 ampere-turns dc. It can then be seen that the rate of distortion of the phase voltage is at all times below 1%.
  • FIG. 13 presents curves obtained by carrying on the abscissa a ratio of impedance ZO / Z, on the ordinate the voltage U ⁇ N phase-neutral at the terminals Pl-P2 of the inductance and in parameter of curves the number of ampere-turns of the direct current magnetic circuit, Zo corresponding to the impedance of a phase, when the direct current magnetic field is zero, and Z to the impedance of this phase for the indicated values of direct current ampere-turns.
  • the impedance ratios decrease with increasing saturation of the alternating current nuclei and that when there is complete saturation the impedance ratio is equal to unity, because then the dipoles of the common magnetic space make a zero angle with the vector of the alternating magnetic field.
  • the saturation occurs at a higher level the higher the transverse direct current magnetic field, as in the case of the control currents of 4848 ampere-turns ce.
  • FIGS. 14a to 14e respectively give the three-phase power curves of the variable inductance for phase-neutral voltages respectively of 80, 160, 200, 240 and 280 volts in rms value.
  • the curve marked "VA” gives the total power (active and reactive) supplied by the industrial tance expressed in volts-amperes and the curve marked watts gives the losses of the inductance in the form of active power expressed in watts
  • the solid lines indicate the volts-amps and the three-phase watts of the variable inductance.
  • the characteristic relating to curve 14a it can be said that these losses decrease under the effect of the increase in the transverse direct current magnetic field.
  • the increase in watts is related to an increase in the components of third and ninth harmonics, as indicated previously.
  • This phenomenon of decreasing losses in the core with the increase in the reactive energy of the variable inductor contributes to increasing the efficiency of the inductor around 96% when the direct current magnetic field reaches a value of 3030 amperes. -tours.
  • FIGS. 15 and 16 illustrate another arrangement of three-phase inductance according to a stack of cylindrical cores of identical cross section.
  • This arrangement allows a symmetrical distribution of the PA, PB and PC phase windings around the legs 1-1 ', 2-2' and 3-3 'of the cores M' and M ", respectively.
  • the control core N whose l winding is supplied with adjustable direct current by terminals El and E2, also comprises legs Nl, N2 and N3 which are mounted opposite legs 1, 2 e; 3 of the core M ', on the one hand , and legs N'l, N'2 and N'3 shown opposite the legs 1 ', 2 and 3' of the core M ", of other art.
  • the operation and characteristics of this three-phase inductor are improved compared to those of the three-phase inductor in Figure 4.
  • the excitation mode proposed in FIG. 17 comprises two superimposed control systems similar to the arrangement described above for FIG. 2, that is to say: a control supplied directly by the high voltage power circuit and a reverse control low power connected to the direct current source V constant, but adjustable.
  • the three-phase current is rectified using diode bridges T and crosses the excitation winding El, E2 before completing its return circuit.
  • a second- the winding is superimposed on the first in the core trole and powered by a low-power constant-current source V.
  • This latter winding is arranged so that the direct current magnetic field generated in the control core N opposes the main direct current magnetic field generated by the self-monitoring winding.
  • the magnetic field resulting in the control core will then be a function of the magnetic field generated by the three-phase alternating current, rectified by T, which flows in the winding in self-control and, consequently, a function of the voltage level at the terminals variable inductance.
  • This control is simple and does not require any feedback loop to correct the desired magnetic torque on the dipoles in the common magnetic space N.
  • This magnetic torque is generated directly by the resulting direct current magnetic field injected into the nucleus of control and the choice of the number of turns of the self-checking winding plays a very important role.
  • the attached table shows the harmonic distortion rates of the phase current obtained when the three-phase inductor in Figure 17 is used either in self-control or in self-control with reverse control.
  • the figures in parentheses refer to the operating points indicated in Figure 18.
  • FIG. 18 presents the characteristic curves of the three-phase cylindrical inductance of FIG. 17 as a function of the ampere-turns of direct current control and as a function of a self-control. More particularly, the curve “X" is that obtained for the operation in self-control only of the inductor while the curve “Y" represents the operating characteristic of the three-phase inductor in self-control with current supply continuous reverse of the control core.
  • variable permeability inductor described above lends itself particularly well to an application as a static compensator when used in parallel with a capacitor bank for power transmission networks. Indeed, as indicated previously, the response time of the variable inductance is about or less than one cycle for a network voltage of 60 Hertz and the tran- sition is done without deformation of the current. In addition, the harmonic distortion of the inductor being very low, no filter other than the connection of the secondary delta is necessary, which contributes to very significantly reducing the cost and increasing the reliability of the static compensator. It should also be noted that this variable inductance can be connected directly to the high voltage of the network and that its losses of iron and copper are comparable to those of a transformer.
  • control mode proposed for the inductor with variable permeability of the cylindrical type illustrated in FIG. 17, is particularly advantageous in an application to the static compensator.
  • This three-phase inductor comprises a self-control circuit coming from the rectification of the current of the inductor and a reverse control of low power coming from an independent direct current source.
  • the inductance thus controlled offers an ideal element for controlling the energy conveyed by an energy transmission line, because the operating range of this inductor is threefold (voltage rise, regulation and overvoltage, the saturation level of the inductance is never reached, the response to a voltage disturbance on the transmission line is instantaneous and its reliability is considerable mainly due to the simplicity of this control.
  • this inductance three-phase becomes the variable element for a static compensator whose performance meets the present needs of energy transmission networks.
  • the phase currents pass from the capacitive state to inductive state in an interval of about 0.5 cycles on a basis of 60 Hertz.
  • This transition from the capacitive state, where I is less than zero, to the inductive state is particularly t well shown in Figure 19 whose curves illustrate the operating points of the static compensator using a variable inductance with reverse control ranging from 0 to 500 negative ampere-turns.
  • the variable inductance described above therefore allows transmission without deformation of the current wave, except for the adjustment of the angle from + 90 ° to - 90 ° relative to the supply voltage of the compensator. ; as for the phase current distortion, it remains negligible.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Coils Or Transformers For Communication (AREA)
  • Supply And Distribution Of Alternating Current (AREA)
  • Control Of Electrical Variables (AREA)
  • Ac-Ac Conversion (AREA)
EP79400766A 1978-10-20 1979-10-19 Variable Induktivität Expired EP0010502B1 (de)

Priority Applications (2)

Application Number Priority Date Filing Date Title
DE8383111475T DE2967595D1 (en) 1978-10-20 1979-10-19 Variable inductance device
DE8383111087T DE2967589D1 (en) 1978-10-20 1979-10-19 Variable inductance for a three-phase circuit

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
CA000313821A CA1118509A (fr) 1978-10-20 1978-10-20 Variable inductance
CA313821 1978-10-20

Related Child Applications (2)

Application Number Title Priority Date Filing Date
EP83111087.9 Division-Into 1979-10-19
EP83111475.6 Division-Into 1979-10-19

Publications (2)

Publication Number Publication Date
EP0010502A1 true EP0010502A1 (de) 1980-04-30
EP0010502B1 EP0010502B1 (de) 1985-07-10

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ID=4112642

Family Applications (3)

Application Number Title Priority Date Filing Date
EP79400766A Expired EP0010502B1 (de) 1978-10-20 1979-10-19 Variable Induktivität
EP83111475A Expired EP0109096B1 (de) 1978-10-20 1979-10-19 Anordnung mit variabler Induktivität
EP83111087A Expired EP0106371B1 (de) 1978-10-20 1979-10-19 Variable Induktivität für Dreiphasenkreis

Family Applications After (2)

Application Number Title Priority Date Filing Date
EP83111475A Expired EP0109096B1 (de) 1978-10-20 1979-10-19 Anordnung mit variabler Induktivität
EP83111087A Expired EP0106371B1 (de) 1978-10-20 1979-10-19 Variable Induktivität für Dreiphasenkreis

Country Status (6)

Country Link
US (1) US4393157A (de)
EP (3) EP0010502B1 (de)
JP (1) JPS6040171B2 (de)
BR (1) BR7906797A (de)
CA (1) CA1118509A (de)
DE (1) DE2967481D1 (de)

Cited By (1)

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EP0106371A3 (en) 1984-05-30
JPS6040171B2 (ja) 1985-09-10
US4393157A (en) 1983-07-12
EP0106371A2 (de) 1984-04-25
JPS5556608A (en) 1980-04-25
EP0010502B1 (de) 1985-07-10
BR7906797A (pt) 1980-06-17
EP0109096B1 (de) 1986-04-30
EP0109096A1 (de) 1984-05-23
CA1118509A (fr) 1982-02-16
DE2967481D1 (en) 1985-08-14
EP0106371B1 (de) 1986-03-26

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