CN1691527A - 直接转换接收机 - Google Patents
直接转换接收机 Download PDFInfo
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- CN1691527A CN1691527A CNA2005100739438A CN200510073943A CN1691527A CN 1691527 A CN1691527 A CN 1691527A CN A2005100739438 A CNA2005100739438 A CN A2005100739438A CN 200510073943 A CN200510073943 A CN 200510073943A CN 1691527 A CN1691527 A CN 1691527A
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D1/00—Demodulation of amplitude-modulated oscillations
- H03D1/22—Homodyne or synchrodyne circuits
- H03D1/2245—Homodyne or synchrodyne circuits using two quadrature channels
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D1/00—Demodulation of amplitude-modulated oscillations
- H03D1/22—Homodyne or synchrodyne circuits
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- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/06—Receivers
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Abstract
根据本发明的零差拍接收机包括一个信号输入端和与该信号输入端相耦合的第一和第二混频器。本机振荡器提供一个在由输入信号调制的频率上的基准信号。由本机振荡器输出的基准信号通过可开关相变元件,其输出直接与第一混频器耦合并通过第二相变元件与第二混频器间接耦合。可开关相变元件根据控制输入的状态将输入信号的相位改变一预定的相位,这样,在一种状态相变元件的输出与在另一状态时相变元件的输出相差π弧度。第二相变元件将输入信号的相位改变π/2弧度。混频器的输出耦合到一对可开关反相器和低通滤波器上。可开关相变元件和可开关反相器的开关是由开关振荡器控制的。
Description
本申请是申请号为99807661.9的专利申请的分案申请。
技术领域
本发明总地涉及直接转换接收机(或零差接收机),它将所接收的信号分成同相(I)和正交(Q)分量,同时也减少由于混频器的非线性而产生的二次互调和调谐的本振上的反馈。
背景技术
超外差接收机是现代通讯装置中应用最广的接收机。这种接收机可在家庭、办公室或汽车的电视机、电话或收音机上找到。超外差接收机将(载波频率为f1的)输入射频(RF)信号与本机振荡器所产生的(在频率f2)正弦信号混频(或相乘)。所生成的输出信号包括两个对输入信号频移的中心在组合频率的和以及差上的频率(f1+f2和f1-f2)。最为典型的是,最高频率分量(在以f1+f2为中心)被用带通滤波器滤波且输出信号仅含中频(IF)分量(以f1-f2为中心)。在高性能的超外差接收机中,这种处理要重复几次。
尽管超外差接收机用得很广,但它们却用了诸如带通滤波器之类昂贵和不能集成的RF和IF元件。结果,超外差接收机不能理想地用在诸如移动电话、寻呼机、无绳电话等体积小成本低的移动通信系统中。
诸如直接转换接收机(或零差接收机)的另类接收机是已知的技术并相对于超外差接收机具有明显的优点。图1传统的直接转换接收机将输入信号直接转换成同相基带分量和正交分量而不必在途中转成中频信号。此传统直接转换接收机的工作很简单。(可被数学地表示为g(t)=gi(t)cos(2πf1t)-gqsin(2πf1t)的)输入带通信号g(t)可在RF输入上接收并通过预选滤波器1和低噪声放大器(LNA)2。预选滤波器1就是一个带通滤波器,使所要的信号g(t)通过并将带外假信号阻掉。在大多数的应用中,预选滤波器的带宽远大于所要信号的带宽。此外,除所要的信号外,该预选滤波器可让不需要的信号通过。
在通过了预选滤波器1之后,信号g(t)被分离,并通过两个混频器3,3′发送出去。在上混频器3′中,信号g(t)与正弦波混频,调谐到与诸如cos(2πf1t)的载频相同的频率上。在下混频器3中,信号g(t)与在上混频器3′中的同样正弦波混频,但其相位差π/2(例如sin(2πf1t))。混频器3,3′以基带为中心产生所要信号g(t)的同相分量gi(t)和正交分量gq(t)以及两倍的载频(2fc)。由低通滤波器6,6′消除高频分量,并由放大器7,7′最终放大同相和正交信号。
直接转换接收机相比于最普通的超外差接收机有几个优点。第一,直接转换接收机直接将输入信号转换成其基带信号并省去了最初将RF信号转成IF信号的步骤。因此,可省去所有的中频滤波器、混频器和放大器,并使电路得以简化。第二,除预选滤波器外,直接转换接收机仅采用低通滤波器而不是带通滤波器。一般来讲,与带通滤波器相比低通滤波器易于集成在一单个芯片上。因此,可在单个集成电路上大量地构造直接转换接收机,这使得它比超外差接收机体积小价格低。
在直接转换接收机相对于超外差接收机具有多个优点的同时,也应看到传统的直接转换接收机也有些缺点。传统直接转换接收机的一个问题是混频器中出现的二次失真。二次失真的引起是由于混频器是固有的非线性器件的缘故。当测出在所要信号上的频道外RF信号时,混频器的非线性在基带加直流偏置上产生所要信号的二次谐波。由于直接转换接收机也将所要的信号移到基带上,由混频器产生的此二次失真就可大大降低接收机的特性。此外,混频器可象“平方律”检波器那样工作,并将强干扰的包络转换成基带。如果干扰的包络在时间上恒定,则在基带上出现直流偏置。在此情况下,有几种已知的方法来抑制此不希望的直流偏置。例如,对混频器输出的基带高通滤波而衰减该直流偏置。尽管此方法有效地消除了直流分量,但由于干扰的非恒定的包络而对失真不起作用。因此,需要一种零差接收机,能衰减由干扰的恒定或非恒定包络引起的失真。
直接转换接收机的另一问题是假发射问题。在直接转换接收机中的假发射的主源是本振泄漏。在普通的超外差接收机中,到天线上的本振泄漏是由第一接收机带通滤波器衰减的。但在直接转换接收机中,本振频率是在预选滤波器的通带之内的。因此,本振泄漏在传统的直接转换接收机中不被抑制。
发明内容
根据本发明的零差接收机包括信号输入端和与此信号输入端相连的第一和第二混频器。本振在输入信号的调制频率上提供一个基准信号。本振输出的基准信号通过可切换相变元件,其输出直接耦合到第一混频器并间接地通过第二相变元件耦合到第二混频器。可切换相变元件根据控制输入的状态以预定相位改变输入信号的相位。在一种状态下相变元件的输出与在另一状态下相变元件的输出差π弧度。第二相变元件将一输入信号的相位改变π/2弧度。混频器的输出与一对可切换反相器和低通滤波器相耦合。可切换相变元件和可切换反相器的切换是由开关振荡器控制的。
本发明的目的在于构造一个无线接收机,它因具有增加了的失真抑制而具有改进的接收质量。
本发明的再一目的在于构造一个无线接收机,它减少了从本振上的假发射。
附图说明
图1为传统的直接转换接收机;
图2是本发明的优选的实施例;
图3是用于实现本发明的反相器元件的电路;
图4和5是本发明的另一个实施例。
具体实施方式
图2示出本发明优选实施例的方框图。在RF输入端上接收一个射频信号并送到预选滤波器1然后至低噪声(LN)放大器2。输入信号随后被分解并馈送到上混频器3′和下混频器3。本机振荡器4产生一个调谐在与输入RF信号的载波同频的频率上的正弦波。本振4所产生的正弦波馈到可切换相变元件5的一个输入上,根据控制输入20的状态将输入正弦波的相位改变-π/2或π/2弧度(-90°或90°)。尽管相变并非必须是-π/2和π/2,但第一相变量和第二相变量之差必须等于π弧度(或180°)。
开关振荡器7产生一个开关信号,它具有对应于“开”态和“关”态的两个状态(例如方波脉冲串)。此开关信号馈入相变元件5的控制输入端20并进入可开关反相器元件9,9′的控制输入端22,22′。当开关信号处于“开”态时,相变元件5和反相器元件9,9′被分别设为-π/2和-1。当开关信号为“关”态时,相变元件5和反相器元件9,9′分别被设为π/2和1。
相变元件5的输出直接连到下混频器3的第二输入上,还连到第二相变元件8上。第二相变元件8连续地将输入的相位改变π/2。第二相变元件8的输出包括对上混频器3′的第二输入。
上混频器3′的输出被馈至上反相器元件9′,且下混频器3的输出被馈至下反相器元件9上。每个反相器元件9,9′的输出随后通过通道滤波器10,10′,并最后通过一个放大器11,11′。
本发明用于抑制在两个混频器中所生成的任何互调,它在本振上未被经历180°的相位改变。由于混频器所产生的所有二次互调未被本振的180°的相移所影响,本发明就可有效地减少由于通道外RF信号的二次谐波、直流偏置和强干扰的包络幅度的影响,进而不会大大影响所要信号的接收质量。
为了示意本发明的优选实施例,假定以已知频率f1传播的所要信号g(t)以图2所示的RF输入上的形式表示出来。在g(t)通过预选滤波器1和LN放大器2之后,可被分解并送到上3′和下3混频器上。本振4产生一正弦波,例如余弦型波形,其频率与g(t)的载频相同(即cos(2πf1t))。因此,本振4所产生的正弦曲线从cos(2πf1t)改变到cos(2πf1t-π/2)=-sin(2πf1t)或cos(2πf1t+π/2)=sin(2πf1t),这要取决于相变元件5的状态。此正弦波随后被分解,一支直接馈入下混频器3的输入端,而另一支则经第二相变元件8馈送。第二相变元件8在±sin(2πf1t)时改变输入相位π/2。因此在此例中,对上混频器3′的输入变为±sin(2πf1t+π/2)=±cos(2πf1t)。
混频器3,3′将g(t)乘以由本振4所产生的正弦波,以在上变频器3′中生成±g(t)*sin(2πf1t)且在下混频器3中生成±g(t)*cos(2πf1t)。混频器3,3′的输出随后被馈到反相器9,9′,它与本振的相变同步地改变开关状态。换言之,当相变元件5相移-π/2时,反相器9,9′则设定为-1。与之相似,当相变元件相移π/2时,反相器9,9′则设为1。反相器9,9′有效地“重构”了所要信号的I和Q分量并产生g(t)*sin(2πf1t)和g(t)*cos(2πf1t)的输出。因此,所要信号的第一次Q和I分量并不受与反相器相耦合的180°相移的影响。
尽管所要信号并未受相变元件5和反相器9,9′的影响,二次互调乘积可明显被衰减。这是因为,所有二次互调乘积不被本振4上180°相移的影响,但却受反相器9,9′中符号改变的影响。因此,反相器9,9′通过以开关振荡器7的频率改换信号的极性而有效地“截止了”二次乘积。这将基带的任何二次失真转到低通滤波器10,10′的止带上。当然,系统还可将开关振荡器7频率周围出现的任何二次失真下转换。为了减少任何二次失真的下转换,开关振荡器7的频率应大于预选滤波器1的带宽。
在衰减基带的二次失真的同时,本发明还得益于它可减少本振4所产生的RF输入中的假发射。假发射在已有技术的零差接收机中是一个问题,因为由本振所产生的信号可通过混频器3,3′传播回来。
再见图2,在本发明中,本振4所产生的信号在加到混频器5,5′之前通过可切换相变元件5。这具有调制本振4输出信号的频谱的效果。如果选定了开关频率f0,则它大于预选滤波器1的带宽,预选滤波器的止带可防止信号耦合回RF输入端,从而减少从本振上泄漏假发射。因此,在优选实施例中,选择开关振荡器的频率使其大于预选滤波器1的带宽。
尽管开关振荡器其频率应大于预选滤波器的带宽,但并不总是要求如此。在优选实施例中,开关振荡器的频率由频率控制器25依伪随机格式改变。用于控制开关振荡器7的开关频率fs的频率控制器25可以以锁相环或其它频率控制装置的方式实现,其设计在已有技术中是已知的。这影响功率谱的扩展,并进而在扩展的信号的带宽大过用于测漏的谱分析仪的分辨带宽时进一步减少本振的泄漏。
在优选实施例中,本振和开关振荡器都锁定在一单一源上,且图2虚线内的所有元件都以集成电路方式实现。
图4和5示出本发明附加的实施例。在图4中,在每个混频器3,3′和反相器9,9′之间添加一低通滤波器12,12′。另外,馈到每个反相器的由开关振荡器7产生的信号在延时元件13上延时τ。延时(τ)被设定为等于低通滤波器12,12′的群延时。这些低通滤波器12,12′被用于限制在反相器9,9′输入上信号的带宽。总地讲,低通滤波器12,12′的角频率远大于开关频率f0。
图5示出本发明的另一实施例。在此电路中,所接收的信号被分解成四路,且每路被分别馈入混频器3,3′,3″,3。本振4将正弦波馈入每个混频器3,3′,3″,3上。在最上混频器3中,本振信号的相位改变π/2(即sin(2πf1t))。在中上的混频器3′中,本振信号的相位改变-π/2(即-sin(2πf1t))。在中下的混频器3″中,本振信号的相位改变π(即-cos(2πf1t))。在最下的混频率3中,本振信号的相位不改变(即cos(2πf1t))。混频器3,3′,3″,3输出的每个都经低通滤波器8,8′,8″,8馈送,并被放大器9,9′,9″,9放大。最后,最上两路被减法器10相减,以在产生所要信号正交分量的同时消除二次互调乘积。与之相似,下两路由减法器10′相减,以在产生所要信号的同相分量的同时消除二次互调乘积。
本发明所有实施例中的每个元件都是已有技术中已知的,并可根据特定应用以各种方式实现。例如,反相器元件可由图3所示的各电压开关来实现。当反相器元件3,3′接收的开关信号处在第一状态时,反相器切换到-1,并将信号的电压反相。当由反相器元件3,3′接收的开关信号处于第二状态时,反相器则切换到+1状态并不加改变地使信号通过。另外,反相器3,3′可通过切换电流而不是电压来控制。
本发明消除了由混频器的非线性而造成的互调乘积引起的偶次失真(二次失真最严重)。
本领域的技术人员将认识到,在不脱离本发明的所附权利要求的精神实质的情况下可做出各种修改和改变。
Claims (7)
1.一种用于接收以已知频率传播的射频信号的装置,包括:
用于接收RF信号的输入装置;
第一混频器,具有第一输入、第二输入和输出,这样,第一输入与所述输入装置电耦合;
调谐的本机振荡器,产生其频率与所要的接收信号的频率相同的输出信号,并与所述第四混频器耦合;
具有输入和输出的第一相变元件,这样,该输入端与所述本机振荡器电耦合,且输出与所述第一混频器电耦合,这样,所述第一相位就是将输入信号的相位改变90°;
具有输入和输出的第二相变元件,这样,该输入端与所述本机振荡器电耦合,且输出与所述第二混频器电耦合,这样,所述第二相位就是将输入信号改变-90°;
与所述第一混频器的输出电耦合的第一低通滤波器;
与所述第二混频器的输出电耦合的第二低通滤波器;
具有第一输入端和第二输入端以及一个输出端的减法器元件,这样,第一输入端与所述第一低通滤波器的输出电耦合,第二输入端与所述第二低通滤波器的输出电耦合,从而使所述减法器元件从在其第一输入上接收的信号中减去在其第二输入中接收的信号。
2.如权利要求1的装置,还包括:
第一放大器,耦合在第一低通滤波器和所述减法器元件第一输入端之间;以及
第二放大器,耦合在第二低通滤波器和所述减法器元件第二输入端之间。
3.如权利要求1的装置,还包括:
具有第一输入端、第二输入端和一输出端的第三混频器,这样,第一输入端与所述输入装置电耦合且所述第二输入端直接与所述本机振荡器的输出相连;
具有输入和输出的第三相变元件,这样,其输入端即与所述本机振荡器电耦合,其输出端与所述混频器的第二输入端电耦合,这样,所述相位就是将输入信号改变180°;
与所述第三混频器的输出电耦合的第三低通滤波器;
与所述第四混频器的输出电耦合的第四低通滤波器;
具有第一输入端和第二输入端以及一个输出端的第二减法器元件,这样,第一输入端与所述第三低通滤波器的输出端电耦合,第二输入端与所述第四低通滤波器的输出端电耦合,这样,所述第二减法器就从其第一输入上接收的信号中减去从其第二输入端上接收的信号。
4.如权利要求3的装置,还包括:
第一放大器,耦合在第一低通滤波器和所述减法器元件第一输入端之间;
第二放大器,耦合在第二低通滤波器和所述减法器元件第二输入端之间;
第三放大器,耦合在第三低通滤波器和第二减法器元件第一输入端之间;以及
第四放大器,耦合在第四低通滤波器和第二减法器元件第二输入端之间。
5.如权利要求1-4中任何一个的装置,还包括一个预选滤波器,耦合在所述混频器和射频输入之间,并且具有一个下截止频率和上截止频率。
6.如权利要求5的装置,其中所述低通滤波器中的每一个都有一个至少等于所述预选滤波器的上下截止频率之间的差的角频率。
7.如权利要求5的装置,还包括一个电串联在所述预选滤波器和所述混频器之间的放大器。
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US09/064930 | 1998-04-22 | ||
US09/064,930 US6192225B1 (en) | 1998-04-22 | 1998-04-22 | Direct conversion receiver |
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CNB998076619A Expired - Fee Related CN1254924C (zh) | 1998-04-22 | 1999-04-20 | 用于接收射频信号的装置及减少失真的方法 |
CNA2005100739438A Pending CN1691527A (zh) | 1998-04-22 | 1999-04-20 | 直接转换接收机 |
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US (1) | US6192225B1 (zh) |
EP (2) | EP1367735A1 (zh) |
JP (2) | JP2002512471A (zh) |
KR (1) | KR100665463B1 (zh) |
CN (2) | CN1254924C (zh) |
AU (1) | AU755670B2 (zh) |
BR (1) | BR9909818A (zh) |
EE (1) | EE200000609A (zh) |
HK (1) | HK1039223B (zh) |
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IL139157A0 (en) | 2001-11-25 |
US6192225B1 (en) | 2001-02-20 |
KR20010034820A (ko) | 2001-04-25 |
MY123208A (en) | 2006-05-31 |
EE200000609A (et) | 2002-02-15 |
HK1039223A1 (en) | 2002-04-12 |
HK1039223B (zh) | 2006-12-15 |
CN1254924C (zh) | 2006-05-03 |
KR100665463B1 (ko) | 2007-01-04 |
EP1074093A1 (en) | 2001-02-07 |
EP1367735A1 (en) | 2003-12-03 |
BR9909818A (pt) | 2000-12-19 |
JP2005168035A (ja) | 2005-06-23 |
WO1999055015A1 (en) | 1999-10-28 |
CN1306699A (zh) | 2001-08-01 |
AU3571099A (en) | 1999-11-08 |
JP2002512471A (ja) | 2002-04-23 |
AU755670B2 (en) | 2002-12-19 |
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