CN108353053B - 高速通信系统 - Google Patents
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L1/00—Arrangements for detecting or preventing errors in the information received
- H04L1/004—Arrangements for detecting or preventing errors in the information received by using forward error control
- H04L1/0041—Arrangements at the transmitter end
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L1/00—Arrangements for detecting or preventing errors in the information received
- H04L1/004—Arrangements for detecting or preventing errors in the information received by using forward error control
- H04L1/0056—Systems characterized by the type of code used
- H04L1/0057—Block codes
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
- H04L25/03006—Arrangements for removing intersymbol interference
- H04L25/03012—Arrangements for removing intersymbol interference operating in the time domain
- H04L25/03019—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception
- H04L25/03057—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception with a recursive structure
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
- H04L25/03878—Line equalisers; line build-out devices
- H04L25/03885—Line equalisers; line build-out devices adaptive
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
- H04L25/03891—Spatial equalizers
- H04L25/03898—Spatial equalizers codebook-based design
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/38—Synchronous or start-stop systems, e.g. for Baudot code
- H04L25/40—Transmitting circuits; Receiving circuits
- H04L25/49—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems
Abstract
对基带和载波调制向量码字进行传送,包括使用多个编码器,其中每个编码器均用于接收信息比特并生成表示向量码字的一组基带编码符号;一个或多个调制电路,每个调制电路均用于对一组相应的基带编码符进行操作,并使用相应的独有的载波频率来产生一组载波调制编码符号;以及,求和电路,该求和电路用于生成一组线路特定输出,每个线路特定输出均代表所述载波调制编码符号当中的相应符号与至少一组基带编码符号之和。
Description
相关申请的交叉引用
本申请为PCT申请,并根据PCT第8条,要求申请号为62/189953,申请日为2015年7月8日,发明人为Ali Hormati,Armin Tajalli和Amin Shokrollahi,名称为“高速通信系统”的美国临时申请,以及申请号为62/185403,申请日为2015年6月26日,发明人为AliHormati,Armin Tajalli和Amin Shokrollahi,名称为“高速通信系统”的美国临时申请的优先权,并通过引用将此两者的内容整体并入本文,以供所有目的之用。
参考文献
以下参考文献通过引用整体并入本文,以供所有目的之用:
公开号为2011/0268225,申请号为12/784414,申请日为2010年5月20日,发明人为Harm Cronie和Amin Shokrollahi,名称为“正交差分向量信令”的美国专利申请,下称《Cronie 1》;
申请号为13/030027,申请日为2011年2月17日,发明人为Harm Cronie、AminShokrollahi和Armin Tajalli,名称为“利用稀疏信令码进行抗噪声干扰、高引脚利用率、低功耗通讯的方法和系统”的美国专利申请,下称《Cronie 2》;
申请号为14/158452,申请日为2014年1月17日,发明人为John Fox、BrianHolden、Peter Hunt、John D Keay、Amin Shokrollahi、Richard Simpson、Anant Singh、Andrew Kevin John Stewart和Giuseppe Surace,名称为“具有低同步开关噪声的芯片间通信的方法和系统”的美国专利申请,下称《Fox1》;
申请号为13/842740,申请日为2013年3月15日,发明人为Brian Holden、AminShokrollahi和Anant Singh,名称为“芯片间通信用向量信令码中偏斜耐受方法以及用于芯片间通信用的向量信令码的高级检测器”的美国专利申请,下称《Holden 1》;
申请号为61/934804,申请日为2014年2月2日,发明人为Ali Hormati和AminShokrollahi,名称为“利用符号间干扰比进行代码评价的方法”的美国临时专利申请,下称《Hormati 1》;
申请号为62/026860,申请日为2014年7月21日,发明人为Ali Hormati和AminShokrollahi,名称为“多点数据传输”的美国临时专利申请,下称《Hormati 2》;
申请号为61/934807,申请日为2014年2月2日,发明人为Amin Shokrollahi,名称为“高引脚利用率的向量信令码及其在芯片间通信及存储中的应用”的美国临时专利申请,下称《Shokrollahi 1》;
申请号为61/839360,申请日为2013年6月23日,发明人为Amin Shokrollahi,名称为“具有低接收器复杂度的向量信令”的美国临时专利申请,下称《Shokrollahi 2》;
申请号为61/946574,申请日为2014年2月28日,发明人为Amin Shokrollahi,Brian Holden和Richard Simpson,名称为“内嵌时钟的向量信令码”的美国临时专利申请,下称《Shokrollahi 3》;
申请号为62/015172,申请日为2014年7月10日,发明人为Amin Shokrollahi和Roger Ulrich,名称为“高信噪比特性的向量信令码”的美国临时专利申请,下称《Shokrollahi 4》;
申请号为13/895206,申请日为2013年5月15日,发明人为Roger Ulrich和PeterHunt,名称为“用于通过差和高效检测芯片间通信用向量信令码的电路”的美国专利申请,下称《Ulrich 1》;
“数据速率为20Gbps以上的常规互连系统受控符号间干扰设计技术”,Wendemagegnehu T.Beyene及Amir Amirkhany,IEEE Transactions on AdvancedPackaging,第31卷,第4期,731~740页,2008年11月,下称《Beyene》。
技术领域
本发明总体涉及通信领域,尤其涉及能够进行信息传递的信号的发送以及在有线通信中对此类信号的检测。
背景技术
通信系统的一个目的在于将信息从一个物理位置传输至另一物理位置。一般而言,此类信息传输的目标在于,可靠、快速且消耗最少的资源。宽泛地说,信息传输方法分为物理通信信道由一种传输方法专属使用的“基带”方法,以及通过分割频域内的物理通信信道而生成两个或多个可供传输方法使用的独立频率信道的“宽带”方法。
进一步地,还可根据物理介质对基带方法进行分类。一种常见的信息传输介质为串行通信链路,此种链路可基于将地面或其他常用基准作为比较对象的单个有线电路,或者基于将地面或其他常用基准作为比较对象的多个此类电路,或者基于相互之间作为比较对象的多个此类电路。后者的常见的一例为使用差分信令。差分信令的工作原理在于,在一条线路中发送信号,并在配对线路中发送所述信号的相反信号。其中,该信号的信息由上述两线路之间的差值,而非由其相对于地面或其他固定基准值的绝对值表示。
并行数据传输也是一种增大互连带宽的常用方法,其总线数从16或16条以下增加至32条,64条以及更多条。由于并行信号线上引发的串扰和噪声会产生接收错误,因此增加奇偶校验对错误检测进行改善,并通过有源总线端接方法解决信号异常问题。然而,此类较宽的数据传输宽度不可避免地导致数据偏斜,从而成为提高总线数据传输吞吐量的限制因素。已开发的替代方法采用更窄的总线宽度以及更快的工作时钟速度,这其中大量的工作投入于通过采用阻抗控制的连接器和微带线布线等方式优化互连介质的传输线特性上。即使如此,其不可避免的路径不完善性要求采用主动均衡及符号间干扰(ISI)消除技术,此类技术包括发送器所用的主动预加重补偿以及接收器所用的连续时间线性均衡(CTLE)及判定反馈均衡(DFE),所有这些均可导致通信接口复杂性和功耗的增大。
与差分信令相比,有多种信令方法可在增加引脚利用率的同时,保持相同的有益特性。此类方法当中的一种为向量信令。通过向量信令,多条线路中的多个信号在保持每个信号的独立性的同时可视为一个整体。因此,向量信令码可融合单电路差分信令的稳健性以及并行数据传输因高线路数而实现的高数据传输吞吐量。承载向量信令码字的传输介质中的整体信号当中的每一个信号均称为分量,而所述多条线路的数目称为码字的“维数”(有时也称为“向量”)。在二元向量信令中,向量的每个分量(或称“符号”)的取值为两个可能取值当中的一值。在非二元向量信令中,每个符号的取值为从由两个以上可能取值所组成的集合中选出的一值。向量符号可取值的集合称为向量信令码的“符集”。在本文中,向量信令码为由长度均为N的称作码字的向量组成的集合C。向量信令码的任何适用子集均为该码的“子码”。此类子码可本身为一种向量信令码。在操作过程中,码字的坐标为有界坐标,我们选用-1和1之间的实数对它们进行表示。集合C大小的二进制对数与长度N之间的比值称为该向量信令码的引脚利用率。当向量信令码的所有码字的坐标之和恒为零时,该向量信令码称为“平衡”码。向量信令方法的其他示例见《Cronie 1》,《Cronie 2》,《Cronie 3》,《Cronie 4》,《Fox 1》,《Fox 2》,《Fox 3》,《Holden 1》,《Shokrollahi 1》,《Shokrollahi 2》及《Hormati I》。
如上所述,宽带信令方法对频域内的可用信息传输介质进行分隔,以生成两个或多个频域“信道”,之后,这些信道便可利用将基带信息转换为频域信道信号的已知载波调制方法,以与基带电路类似的方式传输信息。由于此类信道可在幅度、调制方式和信息编码方式方面独立控制,因此可使得该组信道适应包括信号损失、失真以及噪声随时间和频率的变化等的广范围变化的信息传输介质特性。
非对称数字用户线路(ADSL)是一种广泛用于在传统铜制电话电路上传输数字数据的宽带信令方法。在ADSL方法中,数量可达数百个的频域信道当中的每一个均根据用于传输的铜制电路的具体噪声和损失特性,针对幅度、调制方法及数字载波能力进行独立配置。
发明内容
描述了利用基带和宽带两种技术在多条线路上进行的数字信息通信。在所提供的实施例中,在37.5GHz下具有35dB衰减度的四线通信信道用作与本文所述的系统和方法联用的典型传输介质。在一种实施方式中,所述传输介质上生成两个基于频率的信道,每个信道均使用向量信令码及双二进制编码以每线路56Gb/秒的有效速率在四条线路上传输三数据比特组。
附图说明
图1所示为本文所使用的传输信道模型的频域和时域特性。
图2所示为在两对(四条)传输信道上使用ENRZ信令的第一实施方式的CTLE增益和发送频谱的模拟结果。
图3所示为第一实施方式的接收眼开度的模拟结果。
图4所示为在两对(四条线路)传输信道上使用ENRZ信令及结合双二进制编码的第二实施方式的CTLE增益和发送频谱的模拟结果。
图5所示为第二实施方式的接收眼开度的模拟结果。
图6所示为第三实施方式的宽带和载波信道的频谱。
图7所示为第三实施方式的脉冲响应和跨信道ICI的模拟结果。
图8所示为第三实施方式的接收眼图的模拟结果。
图9为融合了基带和载波频段信令的发送器实施方式的框图。
图10为融合了基带和载波频段信令的另一发送器实施方式的框图。
图11为对基带和载波频段信号进行检测的接收器实施方式的框图。
图12所示为采用基带频段及工作速率为224Gb/秒/线路对的单个载波频段的第四实施方式的眼开度的模拟结果。
图13所示为采用基带频段及工作速率为112Gb/秒/线路对的单个载波频段的第六实施方式的眼开度的模拟结果。
图14所示为采用基带频段及工作速率为224Gb/秒/线路对的单个载波频段的第七实施方式的眼开度的模拟结果。
图15所示为本发明第九和第十实施方式所涉及的数据比特和高冗余度比特在六条子信道和多个相继发送单位间隔的分布。
图16为针对本发明第九和第十实施方式所示的“++--”载波子信道所加入的纠错处理的框图。
具体实施方式
互连一直是大型数字系统设计中的限制因素。无论在是由母板互连的模块层面,还是在大型印刷电路板内互连的功能子系统层面,人们对于无差错的可靠高速数字互连的需求总是使得现有技术捉襟见肘。
本文所述的系统和方法可实现每条互联线路以至少50Gb/秒的数据速率在至少一个发送装置和至少一个接收装置之间进行稳健可靠的数据传输,其中,采用具有图1所示频域和时域特性的例示信道模型。对于熟悉本领域的人员而言显而易见的是,此类传输信道与简单NRZ信令等的常规通信信令方法不兼容,简单NRZ信令例如在112Gb/秒下具有56GHz的奈奎斯特(Nyquist)频率,因此在本发明物理传输信道中对应于46dB的固有衰减水平。
本发明数据速率还使得所附发送和接收装置内的集成电路数据处理能力疲于应付。因此,此类装置内处理的高速数据设定为分布于多个并行处理“阶段”。举例而言,其并非以单一数据路径以100Gb/秒(即比特间仅10皮秒)的速率处理数据,而是可将同一数据流分布于16个处理阶段,从而使得每个处理阶段具有更加合理的160皮秒/比特的处理时间。然而,该处理时间的增加的代价在于额外的处理元件复杂性的显著增大。此外,该分布式处理还导致给定的数字比特结果变的可用之前的延迟时间的增加,从而限制了利用该结果预测后续比特结果的能力,而该能力恰好是DFE方法得以实施的基础。
此外,增大的数据传输速率还因互连设备上传播信号的波长的缩短而导致物理问题。举例而言,在印刷电路微带线上的频率为56GHz的传播信号的波长约为4毫米,因此仅具有部分波长尺寸的周期性异常(甚至包括含所述电路板的浸渍织物的纹理)都可能对信号完整性造成较大干扰,从而彰显了现有均衡和补偿方法的重要性。
采用阿达玛(Hadamard)变换编码信息
如《Cronie 1》中所述,沃尔什-阿达玛(Walsh-Hadamard)变换为由+1和-1组成的所有行和所有列均相互正交的方形矩阵。阿达玛矩阵以其所有大小为2N的形式以及所选取的其他大小的形式著称。本文描述尤其以利用4×4的阿达玛矩阵作为示例编码器。
本文实施例中所采用的4阶阿达玛矩阵为:
通过将三个信息比特A,B,C与阿达玛矩阵H4的第2,第3和第4行相乘,可对这些信息比特进行信息编码,以得到四个输出值,下文称之为“符号值”。按照惯例,这些结果通过适当的常数因子进行缩放,以使该符号值处于+1~-1的范围内。可注意到的是,H4的第一行对应于共模信令,本申请中并不使用该信令,而是利用剩下的三个向量将比特A,B和C分别编码为输出值W,X,Y,Z,这些向量也称为阿达玛码的“模”或“子信道”。由于所编码的输出值同时携带从对A,B和C进行编码获得的信息,因此该输出值为各模的叠加或相加结果,即所述向量信令码的子信道代码向量之和。
熟悉本领域的技术人员将注意到的是,按此方式对A,B,C编码后所获的所有可能值为W,X,Y,Z的模相加值,这些值均为平衡值,也就是说,其和恒常为零。如果W,X,Y,Z的模相加值缩放至使得其最大绝对值为1(即为了便于描述,该信号处于+1~-1范围),则应该注意的是,所有可实现的值均为向量(+1,-1/3,-1/3,-1/3)或向量(-1,1/3,1/3,1/3)的排列组合。这些排列组合称为向量信令码H4的码字。在下文中,该H4码称为整体NRZ码(ENRZ),并用作为后续各例的向量信令码的代表例,但这并不构成任何限制。
ENRZ
《Hormati I》指出ENRZ具有最佳的符号间干扰(ISI)特性,而《Holden1》和《Ulrich1》指出其可进行高效检测。如上所述,ENRZ将三个二进制数据比特编码为四符码字,以例如用于在传输介质的四条线路上进行传输。当在本发明信道的四条线路上使用ENRZ信令时,可实现的数据传输速率仅为75千兆个符号/秒的信令速率,相当于两对传输信道中每对线路的速率为112Gbps。
采用75千兆个符号/秒速率的ENRZ信令及参考信道模型结合的第一实施方式的模拟结果显示,可在发送器内采用2个前馈均衡(FFE)抽头,并在接收器内使用连续时间线性均衡(CTLE)及12个判定反馈均衡(DFE)抽头,其性能如图2各曲线所示。图3的接收眼图模拟结果显示,其垂直眼开度为93mV,边沿间的水平眼开度为14.5皮秒。
双二进制编码
双二进制编码为本领域中的一种已知的解决方案,其中,通过对串行传输数据流中前后相继的比特进行处理而实现对所得发送数据频谱的定形和约束。众所周知,由传输介质的扰动等造成的符号间干扰(ISI)可使得某个单位间隔内所接收的信号幅度受到先前单位间隔的残余能量的干扰。举例而言,传输介质扰动所造成的反向脉冲反射可使得所接收的信号因受到先前发送的信号的剩余影响而被削弱。因此,得悉这一效应的发送器可将当前发送信号值与前一发送信号值相结合,以试图针对该符号间干扰效应提前做好准备或进行预补偿。因此,采用双二进制等的部分响应代码经常被描述为一种特定预均衡滤波形式,其旨在产生建设性而非仅字面意义上的ISI数据编码手段。
如《Beyene》中所述,其他已知的部分响应代码也具有类似的ISI管理能力。作为参考目的,表1中列出了描述此类编码或滤波方法的特征方程。
部分响应系统 | 特征方程 |
双二进制 | x<sub>n</sub>+x<sub>n-1</sub> |
双码 | x<sub>n</sub>-x<sub>n-1</sub> |
变型双二进制 | x<sub>n</sub>-x<sub>n-2</sub> |
2类 | x<sub>n</sub>+2x<sub>n-1</sub>+x<sub>n-2</sub> |
除非另外说明,本申请所实施的双二进制处理假定为将当前单位间隔内发送的信号与前一单位间隔内发送的信号在通过0.5的因子缩放后相加。可选地,其中还可利用发送低通滤波器对发送频谱进行进一步的控制。在其他实施方式中,将ISI控制编码与阿达玛编码以任何顺序结合,其中,该ISI控制编码为如下所述的双二进制,变型双二进制,双码,2类及汉明(Hamming)滤波当中的任何一种。在此类实施方式中,所述ISI控制编码还可描述为由采用上述部分响应编码或滤波方法当中任何一种的部分响应编码器实施。
在极其了解通信信道特性的情况下,可将发送器的ISI控制操作配置为无需接收器实施明确的相反操作,其中,该信道特性本身即可有效地起到实施相反操作的作用。在其他实施方式中,可例如先对二进制数据的双二进制编码所生成的三进制信号进行明确检测,然后实施明确的双二进制至二进制的解码操作。或者,当在接收器中使用DFE等常用的ISI消除技术时,也可有效地实现此类发送器的ISI补偿效应。由于本文中的每一实施例的接收器均已采用DFE,因此不再进一步示出的接收器的双二进制(或其他部分响应代码)处理。
在第二实施方式中,先进行75千兆个符号/秒速率的ENRZ编码,然后对每个线路信号进行双二进制处理,其中,利用上述参考信道模型对采用2个FFE抽头,CTLE及12个DFE抽头的情形进行了模拟,所产生的CTLE增益及频谱结果如图4所述。图5接收眼图的模拟结果显示,其垂直眼开度为75mV,边沿间水平眼开度为13.7皮秒。
虽然上述结果比简单NRZ数据发送的结果有着显著的改善,但是其表明需要额外的工作。
信道化
如果单纯采用基带通信解决方案无法满足要求时,那么宽带方法是否较为有益?历史上,在通过电话网络的传统铜制线路基础设施上提供高速数字服务技术的开发过程中,已发现了存在此类严重程度的物理传输信道限制问题,并且以极低数据速率的代价将其解决。在DSL所需的3Mb数据速率下,传播信号波长为数百米,这与线路端头,接线处以及现场发现的绝缘损伤处之间的典型间隔距离密切相关。对于典型铜制电话信号路径而言,如果不对频率响应进行补偿的话,则将呈现出此类异常之间的反射干扰所导致的许多陷波及斜坡,因导线及绝缘质量下降而发生的损耗性衰减,以及来自例如AM无线电发送器等的噪声源的侵扰性噪声。
针对上述传统传输问题,最终采用多信道频域信道化限制其影响。在一种常用的非对称数字用户线路(ADSL)解决方案中,例如将约1MHz的可用传输介质带宽分割为4.3125kHz的信道。在此之后,对每个信道的衰减度及信噪比进行独立测试,并根据该测试结果为每个信道分配不同的数据吞吐率。如此,可放弃使用遭遇频率响应陷波或严重的外部噪声源的信道频率,并同时满负荷使用其他不存在此类问题的信道。然而,此类高信道数协议的生成和检测依赖于低成本数字信号处理解决方案的可用性,且该技术的性能随时间的变化倍数约为10,而本申请的数据速率的增长倍数约为10万。
因此,虽然上述信道衰减问题表明宽带方法可能较有助益,但是本领域已知的常规高信道数实施方式方法与所期望的数据速率不兼容。因此,需要设计一种专用于高速处理的新方法。
宽带双二进制ENRZ
第三实施方式通过融合ENRZ、双二进制及双频域信道方法,解决上述提议的问题。第一频率信道处于基带上,即类似于上述实施方式的单信道。第二频率信道同样由对正弦载波进行调制的ENRZ信令和双二进制信令构成,其选择为使得基带及载波信道的频谱分量之间的频率重叠最小化。
在以下实施例中,将采用37.5GHz的载波频率,但这并不构成任何限制。如下例所示,在采用30GHz的载波频率的模拟中,获得了类似结果,而且当使用较低频率时,可改善信道的衰减特性,但是信道间干扰会有一定程度的升高。
两个频率信道均采用37.5千兆个符号/秒的信令速率,其中,三个数据比特在基带信道的四条线路上传输,另外三个数据比特采用载波信道在相同的四条线路上传输,以生成与上述实施方式相等的总吞吐量。通过在两个信道分布相同的数据吞吐量,可将所需的每信道信令速率减半,从而可能允许更宽的水平眼开度。
图6所示为通过对运行于上述参考信道模型上的本实施方式进行模拟所产生的宽带和载波信道的频谱以及该两信道信号相应的脉冲形状。
在本实施方式中,先将两个信道当中的每个信道的数据分别进行ENRZ编码,然后通过将当前和前一单位间隔的值乘以因子0.5后相加的方式对承载ENRZ码字的四个信令流当中的每个信令流进行双二进制编码(或者,也可在将各值相加后乘以相同因子,或将该缩放纳入后续的放大和/或滤波功能中)。所产生的两个双二进制编码流(下文也称为基带编码符集)当中的每一个均先利用2个FFE抽头进行预加重,然后经由截止频率为9.37GHz的二阶巴特沃斯(Butterworth)低通滤波器进行频谱整形和ICI削减。载波信道的滤波流对37.5GHz的正弦载波进行调制,其结果与基带信道的滤波流线性组合,以供在传输信道上传输。
由于ENRZ等的阿达玛码的子信道为线性信道,也就是说,其对非二进制及二进制信号进行透明传送,因此双二进制及ENRZ编码的执行顺序可颠倒。在至少一种此类替代实施方式中,针对基带和载波信道当中的每一个,先对三个数据比特当中的每一个分别进行双二进制编码,然后在进行ENRZ编码,而不是将ENRZ码输出值进行双二进制编码。
发送器
图9为宽带双二进制ENRZ发送器的一种实施方式的框图。数据以224Gb/秒的总速率进入MUX 910,并被其分成两个独立的数据流915和918,这两个数据流均为112Gb/秒,且分别作为基带和载波信道的数据输入值。
对基带信道数据进行ENRZ编码920,其中,每三个输入数据比特生成一个由四个符号值组成的码字。随后,每个基带符号值均被独立处理,并最终通过其自身的线路传输(随其相应载波信道处理的符号值一同传输)。每个基带符号值的处理可包括由部分响应信令编码器940实施的双二进制编码以及为了满足系统信号电平标准而由放大器960实施的低通滤波和放大,从而生成处理后的基带输出值。在一些实施方式中,所述部分响应信令编码器可由两组模拟电压生成器实现,其中,每一组均由码字输入交替驱动,以生成一组表示码字符号的电压。然而,所述生成器将其输出维持两个信令间隔的时长。所述各组电压由信号求和电路相加。虽然每组电压以1/2符号速率变化,但是由于其在时间上相互交错,因此所述求和电路的输出也以该符号速率变化,并代表当前符号及前一符号之和。在一些实施方式中,如ENRZ编码器920等的编码器可包括两个同样以1/2速率运行的编码器,每个编码器用于对相应的一组模拟电压生成器进行驱动。
载波信道的处理除了载波调制这一点外其余与基带信道的处理类似,其中,对载波信道数据918进行ENRZ编码930,每三个输入数据比特生成一个由四个符号值组成的码字。随后,每个载波符号值被独立处理后与相应已处理的基带符号值混合,以供线路传输。每个载波符号值的处理包括双二进制编码950,为了满足系统信号电平标准而实施的放大和低通滤波970,以及为了生成已处理调制载波输出而实施的37.5GHz载波的调制980。
四个已处理的基带输出值当中每一个均与其相应已处理的调制的载波输出相加990,从而生成图9标示为线路A,线路B,线路C,和线路D的线路输出。
图10所示为另一发送器的实施方式,其中,先进行双二进制编码1020和1030,然后进行ENRZ编码1040和1050。除了这些操作的顺序之外,该发送器与图9实施方式的发送器完全相同。
接收器
图11为相应宽带双二进制ENRZ接收器的一种实施方式的框图。通过连续时间线性均衡器(CTLE)1110对每个来自传输介质的线路A,线路B,线路C和线路D的每个线路信号实施放大和频率均衡,然后四个经放大和均衡的接收信号输入至三个线性ENRZ混合器1120内。在一些实施方式中,CTLE 1110可包括模拟延迟电路,而且所述接收器可包括用于向每个CTLE1110提供偏斜控制信号的偏斜控制电路1112。在一些实施方式中,所述模拟延迟电路可以为用于对各个线路A~D的模拟延迟进行调节的全通滤波器(例如包括开关电容器组)。在一些实施方式中,偏斜控制电路1112可用于处理对通带多输入比较器(MIC)输出进行处理的采样器1180的输出,以确定用于调整每条线路的模拟延迟值的偏斜控制信号,然而这不应视为构成任何限制。在一种实施方式中,可通过调节判定阈值后测定相应有效眼开度的方式对每个子信道MIC进行评价,如此即可对各条线路的偏斜进行调节,以增大有效眼开度。在一些实施方式中,首先对有效眼开度最窄的子信道MIC进行调节。此外,还可采用本领域技术人员已知的其他模拟延迟电路。
如《Holden 1》所述,上述ENRZ接收混合方法通常由基带的所谓的多输入比较器(MIC)用于对ENRZ码字进行检测。其中,此类MIC中实施的ENRZ混合操作生成包括基带和宽带的线性叠加或两个ENRZ编码流当中每一个的载波调制结果在内的三个线性信号“子信道”。所述混合操作定义为:
R0=(A+C)-(B+D) (式2)
R1=(C+D)-(A+B) (式3)
R2=(A+D)-(B+C) (式4)
其中,R0,R1,R2为从ENRZ混合器1120所得的三个线性信号信道输出,且A,B,C,D为接收自CTLE 1110的四个线路信号输出。此外,通过将线路标号以不同方式排序,还可利用上述各等式的其他代数排列组合获得等效的混合结果。举例而言,当将上述线路以相反顺序标号时,R1=(A+B)-(C+D)相当于式3。采用此类混合结果的MIC还可以通过其定义等式中的线路项的符号来标示(例如,在本实施例中为++--)。
采用截止频率为18.75GHz的四极Butterworth低通滤波器1130从每个线性信号子信道中提取基带分量。作为本领域中的常见做法,采样器1140在特定的时刻或以特定的时间间隔以37.5千兆个样本/秒的速率对每个线性信号子信道的信号幅度进行测量或采集,以112Gb/秒的总数据速率生成三个解码基带数据输出比特。与此同时,每个解码比特均进行DFE计算1150,以生成用于对该比特的采样器阈值进行调节的DFE校正信号。数字反馈均衡为本领域的公知技术,因此此处不再赘述,但是需要注意的是,每次DFE计算1150均为独立计算,且同时实现对传输信道所引起的ISI的校正以及对有意生成的发送器ISI的补偿。
需要注意的是,与例如对接收线路信号进行DFE校正的常规技术不同,上述DFE校正作用于向量信令码的子信道。由于DFE所保有的历史信息必须准确表示以往每个单位间隔的值,因此为了表示具有3个、4个或更多个可能符号值的向量信令码,常规DFE不得不保有三进制,四进制或更高阶的历史值。与此相比,通过采用上述DFE校正,在向量信令码子信道上传输的二进制数据仅要求保有二进制历史信息。
与此同时,截止频率为37.5GHz的二阶Butterworth高通滤波器1150从所述三个线性信号子信道中提取载波信道信息。具有37.5GHz载波信号的平衡混合器1160将这些调制信号转换回基带,其中,与上述基带信道信号一样,先采用截止频率为18.75GHz的四极Butterworth低通滤波器1070进行处理,然后以37.5千兆个样本/秒的速率对每个子信道进行采样1080,以112Gb/秒的总数据速率生成三个解码载波数据输出比特。此外,与上述基带数据一样,每个解码载波数据输出比特均经DFE计算1190生成用于对该比特的采样器阈值进行调节的DFE校正信号。其中,每次DFE计算1190均为独立计算,且同时实现对传输信道所引起的ISI的校正以及对有意生成的发送器ISI的补偿。
由于传输信道具有显著的频率相关的损耗特性,因而将接收基带信道的增益设定为14dB,而将载波信道的增益设定为26dB。类似地,为了实现预加重,载波信道的发送器增益设定为基带信道增益的三倍。
图7所示为本实施方式的脉冲响应和跨信道ICI的模拟结果,其中,假设采用2个发送FFE抽头及15个接收DFE抽头。图8所示为基带和载波(通带)信道的接收眼图。基带的垂直眼开度为54mV,水平眼开度为24.1皮秒,通带的垂直眼开度为56mV,水平眼开度为38.7皮秒,大大优于前述实施方式。
偏斜方面的考虑
与所有向量信令码解决方案一样,由于码字只有在呈现为连贯一体的形式才能为接收器的检波器正确识别,因此必须对在携带该码字符号的传输路径上的偏斜进行约束。粗略地说,各传输路径上的传播延迟必须匹配至小于预期眼宽度的一半才能实现检波,而且必须匹配至优于该值的值才能避免眼宽度性能下降。已知的解决方法包括引入以供路径补偿的可变延迟线路和/或FIFO缓冲器,各线路的独立CDR和采样定时以及发送侧预偏斜补偿。然而,由于这些技术也可能导致符号间干扰、发送时切换噪声以及所测接收共模信号的增大,因此必须慎重使用。
由于基带和载波频带信道携带有通过独立的ENRZ编码的数据,而且此两信道经受独立的接收采样处理,因此其数据流可视为相互独立的数据流,并因而无需绝对时间对准。这一点较为有利,这是因为该两信道间滤波特性差异所导致的不同时间延迟将在基带频段上所接收的一组数据比特与载波频段上所接收的一组数据比特之间引入固有的时间差。对于熟悉本领域的人员而言容易理解的是,此两组比特可由重定时锁存器,FIFO缓冲器或其他已知装置处理,以使其与同一定时基准对齐。
替代实施方式
前述实施方式还存在多种可设想到的变形,而且所有这些变形均属于本发明范围内。经ENRZ符号值生成的发送信号,或者其ISI控制编码,或者此两者可通过具有合适比特数的数模转换器实现。可通过数字方式实现发送器内的宽带和载波信号的混合。
为了满足上述所描述的垂直眼开度,或者为了对与所述参考信道模型的信道特性不同的信道特性进行补偿,各发送器和接收器实施方式还可包括与额外增益和/或频率相关的其他滤波级。为了说明而非限制目的,此类滤波级可提供特定的幅度、增益、衰减等特性。
在至少一种实施方式中,为了免于在接收器内部署大量的DFE抽头,通过在发送器内对信号进行额外的预先滤波,将信道的前几个前导码清零。
在上述例示的宽带接收器实施方式中,将基于载波的信道转换为基带信道,以供后续检波。这其中假定接收器的可用本地载波与发送器的载波信号相一致,并因而利用锁相环或其他已知方法生成。此外,在替换和等同的实施方式中,还可采用本领域内广为人知的已知接收器方法。
在一种接收器实施方式中,可在进行模数转换采样之后,利用数字信号处理方法完成上述滤波、混合及采样操作当中的所有或部分。
扩展至更高的数据速率
本文所述实施方式可扩展至支持每对线路224Gb/秒的数据速率。
在采用该扩展形式的第四实施方式中,通过在发送器内对数据进行预先滤波而进行更为严格的ISI控制。举例而言,其中使用具有如下系数的7阶汉明滤波器:
H=[0.02,0.09,0.23,0.30,0.23,0.09,0.02] (式5)
与此相对,上述实施例中的双二进制编码对应于具有以下系数的发送滤波器:
H=[0.5,0.5] (式6)
在该第四实施方式中,所述基带和载波信道的数据速率均倍增为75千兆个符号/秒,从而使得每条线路的总数据吞吐量为112Gb/秒,或者四条互连线路的总数据吞吐量为448Gb/秒。图12所示为眼开度的模拟结果,其中,假定采用3个发送均衡前导抽头以及15个接收DFE抽头,基带信道的垂直眼开度为93mV,水平眼开度为8.3皮秒,载波信道的垂直眼开度为42mV,水平眼开度为16.6皮秒。
在一种替代实施方式中,可采用额外的载波信道。举例而言,可采用一个基带信道加三个载波信道的组合,所述三个载波信道的工作载波频率选择为使得各信道频谱分量之间的频率重叠最小化,其中,每个信道均承载ENRZ编码和ISI控制编码的数据流,而且每个信道的工作速率均为如上所述的37.5千兆个符号/秒。
扩展至其他基本信令方案
如上所述,本文所述的实施方式可用于上述实施例中为了描述而非限制目的所使用的ENRZ之外的基本向量信令码。此外,本领域普通技术人员可理解的是,还可将上述ISI管理及信道化技术与其他多线路信令方案相结合。
举例而言,本发明的第五实施方式除了在每两对线路上使用信令速率为75Gb/秒/线路对的差分信令而非在所有的四条线路上采用ENRZ之外,与上述第四实施方式完全相同。其中,为了实现更为严格的ISI控制,发送器采用具有如下系数的7阶汉明滤波器对每条信道上的数据进行预先滤波:
H=[0.02,0.09,0.23,0.30,0.23,0.09,0.02] (式7)
如此,该第五实施方式的总吞吐量为300Gb/秒,而且对于所述两个信道当中每个信道,两个线路对的信令速率均为75Gb/秒/线路对。
更低载波频率的使用
如上所述,可以以信道间干扰的增大为代价,通过采用更低的载波频率使所述载波调制信道工作于所述传输信道模型的更低衰减区域。
在第六实施方式中,通过一条基带信道和一条由19.5GHz的载波频率调制的载波信道进行操作。其中,基带和载波信道均采用如上所述的信令速率为37.5千兆波特(GBaud)(对应于26.66皮秒的单位间隔)的ENRZ编码和双二进制滤波。所得信号频谱的基带信道损失为15dB,载波信道损失为30dB。模拟结果示于图13,并在表2中进行了总结,其中,采用600mV的发送幅度,200uV的RMS信道噪声,1:7的基带/载波信道功率比,一个前导发送FIR和一个后导发送FIR,达12dB的接收CTLE,以及12个接收DFE抽头。所观察到的眼开度足以获得至少10-6的比特差错率(BER)。
表2
为了便于描述,载波和基带频率当中每种频率的三个ENRZ子信道由包含其相应多输入混合器的定义方程的逻辑线路组合标示。因此,举例而言,与用于实施(A+B)-(C+D)运算的混合器对应的线路A,B,C,D的混合组合在表2中标示为“++--”。
从图13和表2中可看出,“++--”载波子信道的眼开度远小于其他眼图的眼开度,因此其成为性能的限制因素。具体而言,水平眼开度的减小表示传输信道的线路偏斜对该子信道的影响可能极大。
纠错码的采用
在第七实施方式中,通过一条基带信道和由18.5GHz的载波频率调制的一条载波信道进行操作。其中,基带和载波信道均采用信令速率为75千兆波特(GBaud)(对应于13.33皮秒的单位间隔)的ENRZ编码和11阶汉明滤波。所得信号频谱的基带信道损失为14dB,载波信道损失为22dB。模拟结果示于图14,并在表3中进行了总结,其中,采用800mV的发送幅度,200uV的RMS信道噪声,260飞秒的随机抖动(Rj),1:7的基带/载波信道功率比,一个前导发送FIR和一个后导发送FIR,达12dB的接收CTLE,以及25个接收DFE抽头。
表3
与上述实施例一样,所观察到的眼开度足以获得至少10-6的BER,而且“++--”载波子信道同样为限制总体性能,尤其当传输信道线路偏斜情况的因素存在。
此处已设想出各种方法,用于减轻该子信道限制性能,以实现系统BER的改进。
第八实施方式除了不将上述低性能的“++--”载波子信道用于发送数据之外,与上述第七实施方式完全相同。如此,四条线路传输介质的总吞吐量为5×75=375Gbps,相当于187.25Gbps/线路对的有效速率。
第九实施方式除了在经所述低性能的“++--”载波子信道发送的数据上施加额外的可靠性协议之外,与上述第七实施方式完全相同。作为一种例示但非限制性实施方式,可在该子信道上使用“发送三次”的可靠性协议,以在三个相继单位间隔内发送同一数据比特,并利用接收器处的择多检波器对所接受的数据比特进行识别。因此,在本实施方式中,三个单位间隔内共发送16个比特(而非第七实施方式的18个比特)。如此,四条线路传输介质的总吞吐量为6×75×(16/18)=400Gbps,相当于200Gbps/线路对的有效速率。当所述基本子信道的BER至少为5.7×10-4时,所述可靠性协议的使用可使得有效BER为10-6,相当于增加6dB的垂直眼开度,以及将水平眼开度翻倍。
第十实施方式除了在经所述低性能的“++--”载波子信道发送的数据上施加前向纠错协议之外,与上述第七实施方式完全相同。作为一种例示但非限制性实施方式,可利用[7,4,3]汉明码对四个相继数据比特进行编码,以生成待在所述子信道上以七个单位间隔相继发送的七个汉明编码比特,而且接收器内设置用于恢复所接受数据比特的相应汉明解码器。如此,本实施方式在七个相继单位间隔内供发送39个数据比特(而非第七实施方式的42个比特),从而实现6×75×(39/42)=417.86Gbps的总吞吐量,其相当于208.93Gbps/线路对的有效速率。当所述基本子信道的BER至少为3.6×10-3时,该前向纠错编码的使用可使得有效BER为10-6,相当于增加7dB的垂直眼开度,以及将水平眼开度增大至2.5倍。
本发明第九和第十实施方式所涉及的数据比特和高冗余度比特在六条子信道和多个相继发送单位间隔的分布示于图15。
图16为在已编码的传输子信道中添加有纠错功能和在接收器中识别已纠错数据的功能的框图。在发送器中,如以上参考图9和图10所示,输入数据分布910于载波子信道和基带子信道当中。流向“++--”载波子信道的部分数据比特经纠错功能1510处理后提高其冗余度。与此相比,第九实施方式通过重复获得该冗余度,而第十实施方式通过汉明码编码器获得此冗余度。在此之后,按图9或图10所示方法,对与载波子信道915相关的数据比特及与基带子信道918相关的数据比特进行处理。在接收器中,来自与“++--”混合器载波信道相关的采样器的数据由纠错功能1520进行处理,以识别出原始数据比特。与此相比,第九实施方式采用择多检波器,而第十实施方式采用汉明码解码器实现此目的。来自1520的原始数据比特可与来自其他子信道的采样器输出相结合1530,以产生与呈现于发送器的数据流相同的总接收数据流。
对于本领域技术人员容易理解的是,所述冗余和/或前向纠错操作也可应用于一个以上的子信道,从而使得该子信道的有效眼开度获得相应的改善,然而与此同时,由于这一做法不可避免地导致额外开销,因此其也将使得所得数据速率下降。因此,上述将此类解决方案应用于单个子信道的实施例不应被视为构成任何限制,而是应视为该实施例参数的优选实施例。
Claims (14)
1.一种高速通信装置,其特征在于,包括:
多个编码器,每个编码器均用于接收一组相应的信息比特并生成表示向量码字的一组基带编码符号以用来在多条线路上进行传输,其中线路的数目对应于所述一组基带编码符号中的符号数目;
一个或多个调制电路,每个调制电路均用于对一组相应的基带编码符号进行操作,并利用相应的独有的载波频率生成一组载波调制编码符号,其中各个调制电路具有相应的调制器,以用来对相应的基带编码符号进行调制,从而生成相应的载波调制编码符号以用来在所述多条线路的相应线路上传输;以及
求和电路,所述求和电路用于生成一组线路特定输出,每个线路特定输出均代表所述基带编码符号当中的相应符号与至少一组载波调制符号当中的相应符号之和,其中所述求和电路用于提供在所述多条线路上进行传输的所述线路特定输出。
2.如权利要求1所述的装置,其特征在于,所述多个编码器当中的每个编码器均包括用于生成部分响应基带编码符号的部分响应编码器。
3.如权利要求2所述的装置,其特征在于,所述部分响应编码器选自:双二进制部分响应编码器、变型双二进制部分响应编码器、双码部分响应编码器、2类部分响应编码器以及汉明滤波器部分响应编码器。
4.如权利要求2所述的装置,其特征在于,所述部分响应编码器用于生成多个整体不归零向量码字的符号级加权求和结果。
5.如权利要求4所述的装置,其特征在于,所述多个整体不归零向量码字包括相继的整体不归零码字对。
6.如权利要求1~5当中任何一项所述的装置,其特征在于,所述多个编码器包括生成整体不归零向量码字的多个整体不归零编码器。
7.如权利要求6所述的装置,其特征在于,每个整体不归零码字表示子信道代码向量之和,其中每个子信道代码向量的加权取决于接收信息比特当中的相应信息比特所确定的相应对跖权重。
8.如权利要求1~5当中任何一项所述的装置,其特征在于,还包括纠错编码器,所述纠错编码器用于提供接收信息比特的一部分,其中所述接收信息比特的所述一部分具有更高冗余度。
9.如权利要求8所述的装置,其特征在于,由所述纠错编码器提供的所述接收信息比特的所述一部分通过比特重复实现更高冗余度。
10.如权利要求8所述的装置,其特征在于,所述纠错编码器包括用来生成所述接收信息比特的所述一部分的前向纠错编码。
11.如权利要求10所述的装置,其特征在于,所述前向纠错编码为[7,4,3]的汉明码。
12.一种高速通信方法,其特征在于,包括:
接收信息比特并相应地生成至少两组基带符号,每组基带符号均表示用于在多条线路上进行传输的相应的基带向量码字,其中线路的数目对应于各组基带符号中的符号数目;
通过采用至少一个载波对至少一组基带符号进行调制,以生成至少一个载波调制码字,其中所述至少一组基带符号当中的各个符号通过相应的调制器进行调制;以及
在所述多条线路上并行发送表示一个基带码字和至少一个载波调制码字之和的码字叠加信号。
13.如权利要求12所述的方法,其特征在于,所述多条线路当中的每条线路均由来自基带码字的一个基带符号以及载波调制码字的载波调制符号进行驱动。
14.如权利要求12或13所述的方法,其特征在于,所发送的码字叠加信号包括部分响应编码的基带码字和部分响应编码的载波调制码字。
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KR20180030987A (ko) | 2018-03-27 |
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US10819541B2 (en) | 2020-10-27 |
KR101978470B1 (ko) | 2019-05-14 |
CN113193938B (zh) | 2023-10-27 |
US20190363912A1 (en) | 2019-11-28 |
US10608850B2 (en) | 2020-03-31 |
EP3314835B1 (en) | 2020-04-08 |
US20160380787A1 (en) | 2016-12-29 |
KR102517583B1 (ko) | 2023-04-03 |
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WO2016210445A1 (en) | 2016-12-29 |
CN108353053A (zh) | 2018-07-31 |
US9832046B2 (en) | 2017-11-28 |
US11863358B2 (en) | 2024-01-02 |
US20210111931A1 (en) | 2021-04-15 |
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