WO2021077757A1 - Wide gain control method for variable topology llc resonant converter - Google Patents

Wide gain control method for variable topology llc resonant converter Download PDF

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WO2021077757A1
WO2021077757A1 PCT/CN2020/094484 CN2020094484W WO2021077757A1 WO 2021077757 A1 WO2021077757 A1 WO 2021077757A1 CN 2020094484 W CN2020094484 W CN 2020094484W WO 2021077757 A1 WO2021077757 A1 WO 2021077757A1
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switching tube
switching
duty cycle
switch
turned
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PCT/CN2020/094484
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French (fr)
Chinese (zh)
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杜鹃
李思远
李永昌
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广州金升阳科技有限公司
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to the technical field of switching converters, in particular to a wide gain control method of a variable topology LLC resonant converter.
  • LLC resonant converter has many advantages such as low noise, low stress and low switching loss. LLC resonant converter generally adopts frequency conversion and fixed frequency control.
  • the operating frequency of the LLC resonant converter that uses frequency conversion control alone has a wide range of changes, which makes the magnetic components in the circuit
  • the design is difficult, and when the voltage gain is wide, the efficiency of the traditional frequency conversion control LLC resonant converter is significantly reduced; while the LLC that uses the fixed frequency and phase shift control alone, because the operating frequency is fixed, the magnetic components are easy to design, but in order to make the input voltage and The load ensures the output voltage remains unchanged in a wide range, and the circuit needs to work at a large phase shift angle, which will also make it difficult for the lagging bridge arm in the phase shift circuit to achieve soft switching.
  • the lagging bridge arm in order to meet the Under the maximum phase shift angle, the lagging bridge arm can realize the requirement of soft switching, and the phase shift angle will be limited, which in turn leads to the limitation of the gain range of the traditional fixed frequency phase shift control LLC converter.
  • the circuit In short, when LLC resonant converter is used in ultra-wide input occasions, the circuit cannot take into account the characteristics of high efficiency and high gain.
  • the technical problem to be solved by the present invention is to provide a wide gain control method of a variable topology LLC resonant converter to meet the requirement of a wider input voltage.
  • the control method of the present invention adopts the technical scheme as follows: a wide gain control method of a variable topology LLC resonant converter, which is applied to a variable topology LLC resonant converter composed of an inverter circuit, an LLC resonant cavity, a transformer and a secondary rectifier filter output circuit ,
  • the inverter circuit includes switch tubes S1, S2, S3, S4, which can switch from full bridge to half bridge structure;
  • the LLC resonant cavity includes switch tubes S5, S6, resonance inductance Lr, transformer magnetizing inductance Lm, and resonance Capacitor Cr;
  • the drain of the switching tube S1 is connected to the drain of the switching tube S2 and the positive terminal of the input power supply Vin, the source of the switching tube S1 is connected to the drain of the switching tube S3 and one end of the resonant capacitor Cr, the resonant capacitor Cr The other end is connected to one end of the resonant inductor Lr and the drain of the switch S5.
  • the other end of the resonant inductor Lr is connected to one end of the magnetizing inductance Lm and the first end of the transformer T primary winding Np, and the second end of the transformer T primary winding Np Connected to the other end of the magnetizing inductance Lm, the source of the switching tube S2, the drain of the switching tube S4, and the drain of the switching tube S6.
  • the source of the switching tube S4 is connected to the source of the switching tube S3 and the negative electrode of the input power Vin
  • the source of the switching tube S6 is connected to the source of the switching tube S5; it is characterized in that the input voltage range is divided into three voltage segments, low, medium, and high, respectively, corresponding to three different modes,
  • variable-topology LLC resonant converter adopts variable-frequency PFM control of the FBLLC structure in the low-voltage section of the input voltage, and changes the output voltage gain by changing the switching frequency;
  • variable-topology LLC resonant converter adopts fixed-frequency PWM control of the FBLLC structure in the medium voltage section of the input voltage, and changes the output voltage gain by changing the duty cycle of the switch tube (S1) of the inverter circuit;
  • variable-topology LLC resonant converter adopts the fixed frequency PWM control of the HBLLC structure in the high input voltage section, and changes the output voltage gain by changing the duty cycle of the switch tube (S1) of the inverter circuit.
  • the inverter circuit works in the variable frequency PFM control mode of the FBLLC structure (FBLLC variable frequency mode), the switching tubes S5 to S6 are continuously turned off, and the switching tubes S1 to S4 maintain a duty cycle of 0.5 and Fixed, the switching tube S1 and the switching tube S2 are turned on complementarily, the switching tube S1 and the switching tube S4 are turned on and turned off at the same time, the switching tube S2 and the switching tube S3 are turned on and turned off at the same time, by adjusting the switching tubes S1 ⁇ S4
  • the output voltage V 0 can be controlled by the switching frequency of, the smaller the switching frequency, the greater the output voltage gain;
  • the inverter circuit works in the fixed frequency PWM control mode of the FBLLC structure (FBLLC variable duty cycle mode), the switching frequencies of the switching tubes S1 ⁇ S6 are equal and fixed, and the switching tube S1 and the switching tube S5 is complementarily turned on, the switching tube S2 and the switching tube S6 are complementarily turned on, the switching tube S1 and the switching tube S4 are turned on and turned off at the same time, and the switching tube S2 and the switching tube S3 are turned on and turned off at the same time.
  • the duty cycle is equal to the duty cycle of the switching tube S2, neither is greater than 0.5 and the phase difference between the two is 180°.
  • the duty cycle of the switching tube S5 is equal to the duty cycle of the switching tube S6, neither is less than 0.5 and both phases The difference is 180°.
  • the output voltage V 0 is controlled by adjusting the duty cycle of the switching tube S1 (the duty cycle of the switching tube S1 changes, and the conduction time of the bidirectional switch changes synchronously). The larger the duty cycle of the switching tube S1 is , The greater the output voltage gain;
  • the inverter circuit works in the fixed frequency PWM control mode of the HBLLC structure (HBLLC variable duty cycle mode), the switching frequencies of the switching tubes S1 ⁇ S6 are equal and fixed, and the switching tube S1 and the switching tube S5 Complementary conduction, the switching tube S3 and the switching tube S6 are complementarily turned on, the switching tube S4 is continuously turned on, and the switching tube S2 is continuously turned off.
  • HBLLC variable duty cycle mode HBLLC variable duty cycle mode
  • the duty cycle of the switching tube S1 is equal to that of the switching tube S3, and both are not greater than 0.5 And the phase difference between the two is 180°, the duty cycle of the switching tube S5 is equal to the duty cycle of the switching tube S6, both are not less than 0.5 and the phase difference between the two is 180°, the output voltage is realized by adjusting the duty cycle of the switching tube S1 For the control of V 0 , the greater the duty cycle of the switch S1, the greater the output voltage gain.
  • the invention adopts the combination of variable frequency PFM control and fixed frequency PWM control, and through the switching of full-bridge and half-bridge topological structures, thereby cooperating with the circuit to achieve a wider voltage gain range and higher efficiency, so that the converter can be applied to more Where a wide gain range is required.
  • the control method of the present invention has a small frequency conversion range, low requirements on magnetic components such as transformers, inductors, and the like. There are no leading bridge arms and lagging bridge arms. Even if the variable topology is not combined, the control method is compared with single frequency conversion PFM control. Wide gain range and high efficiency.
  • Figure 1 shows the equivalent circuit diagram of the resonant converter when the HBLLC topology is used
  • FIG. 1 shows the comparison of HBLLC and FBLLC gain curves
  • Fig. 3 is a circuit schematic diagram of a variable topology LLC resonant converter according to a preferred embodiment of the present invention
  • Fig. 4 is a gain curve of a variable topology LLC resonant converter according to a preferred embodiment of the present invention.
  • Fig. 5 is the main working waveform of the variable topology LLC resonant converter in the FBLLC frequency conversion mode of the preferred embodiment of the present invention
  • 6-11 are equivalent circuit diagrams of each switch mode when the variable topology LLC resonant converter of the preferred embodiment of the present invention works in the FBLLC variable frequency mode;
  • FIG. 12 is the main working waveform of the variable topology LLC resonant converter in the FBLLC variable duty cycle mode of the preferred embodiment of the present invention.
  • 13-18 are equivalent circuit diagrams of various switching modes when the variable topology LLC resonant converter of the preferred embodiment of the present invention works in the FBLLC variable duty cycle mode;
  • FIG. 19 is the main working waveform of the variable topology LLC resonant converter in the HBLLC variable duty cycle mode of the preferred embodiment of the present invention.
  • variable topology LLC resonant converter of the preferred embodiment of the present invention works in the HBLLC variable duty cycle mode.
  • variable topology LLC resonant converter of this embodiment includes an inverter circuit 10, an LLC resonant cavity 20, a transformer T, and a rectifier network 30 that are sequentially connected from input to output.
  • Vin is the input power of the converter
  • Ro is the output load R 0 of the converter.
  • the inverter circuit 10 is a full-bridge/half-bridge combined variable topology circuit, which is composed of a switching tube S1, a switching tube S2, a switching tube S3, and a switching tube S4.
  • the LLC resonant cavity 20 includes a resonant inductance Lr, an excitation inductance Lm and a resonant capacitor Cr, and is additionally provided with a bidirectional switch composed of a switch tube S5 and a switch tube S6.
  • the rectifier network 30 is composed of a full-bridge rectifier circuit composed of four diodes D1-D4 in parallel with an output filter capacitor C 0 .
  • the drain of the switch S1 is connected to the drain of the switch S2 and the positive terminal of the input power Vin
  • the source of the switch S1 is connected to the drain of the switch S3 and one end of the resonant capacitor Cr
  • the other end of the resonant capacitor Cr is connected
  • One end of the resonant inductor Lr and the drain of the switch S5 the other end of the resonant inductor Lr is connected to one end of the excitation inductance Lm and one end of the transformer T primary winding Np
  • the second end of the transformer T primary winding Np is connected to the excitation
  • the source of the switching tube S4 is connected to the source of the switching tube S3 and the negative electrode of the input power supply Vin.
  • the source of S6 is connected to the source of the switch S5; the 1 end of the secondary winding Ns of the transformer T is connected to the anode of the secondary rectifier diode D1 and the cathode of the secondary rectifier diode D3, and the cathode of the secondary rectifier diode D1 is connected to The cathode of the secondary side rectifier diode D2, one end of the secondary side output filter capacitor Co and one end of the output load Ro, the other end of the output load Ro is connected to the other end of the secondary side output filter capacitor Co and the anode of the secondary side rectifier diode D3 And the anode of the secondary rectifier diode D4, the cathode of the secondary rectifier diode D4 is connected to the anode of the secondary rectifier diode D2 and the 2 ends of the secondary winding Ns of the transformer T.
  • Terminal 1 of the primary winding and secondary winding of the transformer are mutually homonymous terminals
  • terminals 2 of the primary winding and secondary winding of the transformer are mutually homonymous terminals.
  • the above-mentioned ultra-wide gain range variable topology LLC resonant converter can achieve a gain range of 8:1, and the gain range of the half bridge is half of the full bridge; a bidirectional switch is added to the resonant cavity of the traditional LLC resonant circuit, and the circuit topology is unchanged Under the premise of controlling the on-time of the bidirectional switches S5 and S6, the fixed-frequency PWM control can reduce the output voltage gain of the circuit by at least half.
  • the full-bridge topology is variable-frequency switching
  • the circuit gain range can achieve 2-0.25, as shown in Figure 4.
  • variable topology LLC resonant converter with ultra-wide gain range can adopt the following variable frequency control methods:
  • the inverter circuit works in the variable frequency PFM control mode of the FBLLC structure, the switching tubes S5 ⁇ S6 are continuously turned off, the switching tubes S1 ⁇ S4 keep the duty cycle at 0.5 and fixed, the switching tube S1 and the switching tube S2 is complementarily turned on, the switching tube S1 and the switching tube S4 are turned on and off at the same time, and the switching tube S2 and the switching tube S3 are turned on and off at the same time.
  • the output voltage V is achieved by adjusting the switching frequency of the switching tubes S1 ⁇ S4. 0 control, the smaller the switching frequency, the greater the output voltage gain;
  • variable topology LLC resonant converter with ultra-wide gain range can adopt the following fixed frequency PWM control method:
  • the inverter circuit works in the fixed frequency PWM control mode of the FBLLC structure (FBLLC variable duty cycle mode), the switching frequencies of the switching tubes S1 ⁇ S6 are equal and fixed, and the switching tube S1 and the switching tube S5 is complementarily turned on, the switching tube S2 and the switching tube S6 are complementarily turned on, the switching tube S1 and the switching tube S4 are turned on and turned off at the same time, and the switching tube S2 and the switching tube S3 are turned on and turned off at the same time.
  • the duty cycle is equal to the duty cycle of the switch S2, neither is greater than 0.5 and the phase difference between the two is 180°.
  • the duty cycle of the switch S5 is equal to the duty cycle of the switch S6, neither is less than 0.5 and both phases The difference is 180°, the output voltage V 0 is controlled by adjusting the duty ratio of the switching tube S1. The larger the duty ratio of the switching tube S1, the greater the output voltage gain;
  • the inverter circuit works in the fixed frequency PWM control mode of the HBLLC structure (FBLLC variable duty cycle mode), the switching frequencies of the switching tubes S1 ⁇ S6 are equal and fixed, and the switching tube S1 and the switching tube S5 Complementary conduction, the switching tube S3 and the switching tube S6 are complementarily turned on, the switching tube S4 is continuously turned on, and the switching tube S2 is continuously turned off.
  • FBLLC variable duty cycle mode the switching frequencies of the switching tubes S1 ⁇ S6 are equal and fixed, and the switching tube S1 and the switching tube S5 Complementary conduction, the switching tube S3 and the switching tube S6 are complementarily turned on, the switching tube S4 is continuously turned on, and the switching tube S2 is continuously turned off.
  • the duty cycle of the switching tube S1 is equal to that of the switching tube S3, and both are not greater than 0.5 And the phase difference between the two is 180°, the duty cycle of the switching tube S5 is equal to the duty cycle of the switching tube S6, both are not less than 0.5 and the phase difference between the two is 180°, the output voltage is realized by adjusting the duty cycle of the switching tube S1 For the control of V 0 , the greater the duty cycle of the switch S1, the greater the output voltage gain.
  • a reasonable dead time must be set between the switching signals of the switching tube S1 and the switching tube S5 to realize the soft switching of the switching tube S1, the switching tube S4, and the switching tube S5; the switching of the switching tube S2 and the switching tube S6
  • a reasonable dead time must be set between the signals to realize the soft switching of the switching tube S2, the switching tube S3, and the switching tube S6.
  • Coss1 to Coss6 respectively represent the output capacitance to the six switching tubes S1 to S6.
  • variable-topology LLC resonant converter using variable-frequency PFM control and fixed-frequency PWM control will be described in detail below.
  • FIG. 5 shows the main features of this resonant converter when the FBLLC frequency conversion mode is used.
  • Working waveform diagram Vgs1/4 is the driving signal of the switching tubes S1 and S4, Vgs2/3 is the driving signal of the switching tubes S2 and S3, Vc, iLr, iLm, and i 0 respectively represent the voltage across Cr, the current through Lr, The current through Lm and the current through resistor R 0 . It can be seen from FIG. 5 that the output current I 0 of the present invention changes smoothly, and the device stress is small.
  • the converter has six switching modes in a half cycle, as shown in Figures 6 to 11 (the working modes of the second half cycle and the first half cycle of the LLC resonant converter are symmetrical. It can also be seen from the waveform diagram. Generally speaking, the LLC resonant converter has six switching modes. The description of the resonant converter only describes half a cycle).
  • Switch mode 1 [t 0 , t 1 ]: As shown in Figure 6, at t0, the switching tube S1 and the switching tube S4 are turned on at zero voltage; the rectifier diode D1 and the rectifier diode D4 are turned on, and the current flowing through the diode is in resonance The current is proportional to the difference between the excitation current; the voltage across the excitation inductance Lm is output clamped to nV O (n is the transformer turns ratio); the primary resonant inductance Lr and the resonant capacitor Cr participate in resonance, and the resonant current iLr is a standard sine wave If it is a negative value, the magnetizing inductance current iLm increases linearly, but it is less than the resonant current iLr;
  • Switching mode 2 [t 1 , t 2 ]: As shown in Figure 7, at t 1 , the resonant current iLr crosses the zero point; the rectifier diode D1 and the rectifier diode D4 continue to conduct; the voltage across the excitation inductor Lm is clamped by the output Bit to nVo; the primary side resonant inductance Lr and resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and is positive, and the magnetizing inductor current iLm increases linearly and is still less than the resonant current iLr;
  • Switching mode 3 [t 2 , t 3 ]: As shown in Figure 8, at t 2 the switching tube S1 and the switching tube S4 remain conductive, and the zero-crossing point of the excitation current ILm changes to the positive direction; the rectifier diode D1, the rectifier diode D4 continues to conduct; the voltage across the magnetizing inductance Lm is output clamped to nVo; the primary resonant inductance Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and a positive value, and the magnetizing inductance current iLm increases linearly, but still Less than the resonant current iLr;
  • Switch mode 4 [t 3 , t 4 ]: As shown in Figure 9, at t3, the excitation current ILm is equal to the resonant current ILr, and the excitation inductance is no longer clamped (the transformer has no energy transmission); it flows through the rectifier diode The current of D1 naturally crosses 0, and the secondary side rectifier diodes D1 and D4 are turned off with zero current to avoid the diode reverse recovery problem; the primary side resonant inductance Lr, resonant capacitor Cr, and magnetizing inductance Lm participate in resonance together, and the load energy is completely driven by the output capacitor Co provide;
  • Switching mode 5 [t 4 , t 5 ]: As shown in Figure 10, this period is the dead time, the excitation current and the resonant current iLr are equal and remain unchanged, and the secondary side rectifier diode is still in the reverse cut-off state; all The power tube is turned off; the primary resonance inductance Lr, the resonance capacitor Cr, and the excitation inductance Lm participate in resonance together; the resonance current iLr charges the output capacitors C oss1 and C oss4 of the switching tube S1 and the switching tube S4, and charging the switching tube S2 and the switching tube The output capacitors C oss2 and C oss3 of S3 are discharged, and the load energy is completely provided by the output capacitor Co;
  • Switch mode 6 [t 5 , t 6 ]: As shown in Figure 11, this period is still dead time, when the voltage across the output capacitors C oss2 and C oss3 of the switch S2 and S3 drops to 0, The body diodes of the switching tube S2 and the switching tube S3 are turned on to provide conditions for the switching tube S2 and the switching tube S3 to realize zero voltage turn-on; at t6, the switching tube S2 and the switching tube S3 realize ZVS, and the circuit enters the second half cycle.
  • FIG. 12 shows the main operating waveforms of this resonant converter when fixed-frequency PWM control is used.
  • Vgs1/4 is the switch The driving signal of the tubes S1 and S4,
  • Vgs2/3 is the driving signal of the switching tubes S2 and S3,
  • Vgs5 is the driving signal of the switching tube S5,
  • Vgs6 is the driving signal of the switching tube S6,
  • Vc, iLr, iLm, and i 0 respectively represent Cr
  • the voltage across the terminals, the current through Lr, the current through Lm, and the current through the resistor R 0 It can be seen from Fig.
  • Switch mode 1 [t 0 , t 1 ]: As shown in Figure 13, before time t 0 , the switch S6 has been turned on, the switch S5 is turned off, and its body diode bears a reverse voltage and reversely cuts off; At t0, the switching tube S1 and the switching tube S4 are turned on at zero voltage; the rectifier diode D1 and the rectifier diode D4 are turned on, and the current flowing through the diode is proportional to the difference between the resonance current and the excitation current; the voltage across the excitation inductance Lm is output clamped Bit to nV O (n is the transformer turns ratio); the primary side resonant inductor Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and is negative, and the magnetizing inductor current iLm increases linearly, but is less than the resonant current iLr;
  • Switching mode 2 [t 1 , t 2 ]: As shown in Figure 14, at t 1 , the resonant current iLr crosses the zero point; the rectifier diode D1 and the rectifier diode D4 continue to conduct; the voltage across the excitation inductor Lm is clamped by the output Bit to nVO; the primary side resonant inductor Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and is positive, and the excitation inductor current iLm increases linearly, but is less than the resonant current iLr;
  • Switch mode 3 [t 2 , t 3 ]: As shown in Figure 15, at t 2 the switch S1 and the switch S4 are turned off, the resonance current iLr is still greater than the excitation inductance current iLm, the rectifier diode D1 and the rectifier diode D4 continues to conduct; resonant current iLr to switch Sl, the switch output capacitance C oss1 S4 of, C oss4 charge, to the switch S2, the switch S3 output capacitor C oss2, C oss3 discharge, a switch S5, the output capacitor C Oss5 discharges; when the voltage across the capacitor C oss5 drops to zero, the body diode of the switching tube S5 is turned on, which provides conditions for the switching tube S5 to realize zero voltage turn-on;
  • Switch Mode 4 [t 3, t 4] : As shown in FIG. 16, the switch S5 ZVS at time t 3, the switch S6 and the rectifying diode D1, a rectifier diode D4 is still turned on; magnetizing inductance Lm is still The output voltage is clamped, the excitation current iLm continues to increase linearly, and the resonant current iLr decreases linearly;
  • Switching mode 5 [t 4 , t 5 ]: As shown in Figure 17, at t 4 , the resonant current iLr is equal to the excitation current iLm, the current flowing through the rectifier diode D1 naturally passes through 0, and the secondary side rectifier diode D1 is rectified Diode D4 is turned off at zero current to avoid diode reverse recovery problems; switch S5 and switch S6 continue to conduct, and the excitation current and resonance current iLr are equal and remain unchanged;
  • Switching mode 6 [t 5 , t 6 ]: As shown in Figure 18, at t 5 , the switch S6 is turned off while the switch S5 continues to conduct; the resonant current iLr is equal to the excitation current iLm, and the secondary rectifier diode is still in reverse off-state; resonant current iLr to switch Sl, switch the output capacitor C oss1 S4, C oss4 charge, to the switch S2, the switch of the output capacitor C oss2 S3, C oss3 discharge, to the output of switch S6
  • the capacitor C oss6 is charged; when the voltage across the capacitors C oss2 and C oss3 drops to 0, the body diodes of the switch S2 and S3 are turned on, providing conditions for the switch S2 and S3 to achieve zero voltage turn-on; t6 At the moment, the switch tube S2 and the switch tube S3 realize ZVS, and the circuit enters the second half cycle.
  • Vgs1 ⁇ Vgs6 are the driving signals of the switching tubes S1 ⁇ S6, and Vc, iLr, iLm, and i 0 respectively represent the voltage across Cr, the current through Lr, the current through Lm, and the current through Lm.
  • the variator also has six switching modes in this half cycle, as shown in Figures 20-25.
  • Switch mode 1 [t 0 , t 1 ]: As shown in Figure 20, before time t 0 , the switching tube S6 is turned on, the switching tube S5 is turned off, and its body diode bears the reverse voltage and reversely stops; At t 0 , the switching tube S1 is turned on with zero voltage; the rectifier diode D1 and the rectifier diode D4 are turned on, and the current flowing through the diode is proportional to the difference between the resonant current and the excitation current; the voltage across the excitation inductance Lm is output clamped to nV O ; The primary side resonant inductance Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and is negative, and the excitation inductance current iLm increases linearly, but is less than the resonant current iLr;
  • Switching mode 2 [t 1 , t 2 ]: As shown in Figure 21, at time t 1 , the resonant current iLr crosses the zero point; the rectifier diode D1 continues to conduct; the voltage across the magnetizing inductance Lm is output clamped to nV O ; The primary side resonant inductance Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and a positive value, and the excitation inductance current iLm increases linearly, but is less than the resonant current iLr;
  • Switching mode 3 [t 2 , t 3 ]: As shown in Figure 22, at t 2 the switch S1 and S4 are turned off, the resonance current iLr is still greater than the magnetizing inductance current iLm, the rectifier diode D1, and the rectifier diode D4 continues to conduct; iLr resonant current to the output capacitor C oss1 charge switch S1, the switch S3 to the output capacitor C oss3 discharge the output capacitance C oss5 discharge switch S5; when the voltage across the capacitor drops to zero C oss5 , The body diode of the switching tube S5 is turned on, which provides conditions for the switching tube S5 to realize zero voltage turn-on.
  • Switching mode 4 [t 3 , t 4 ]: As shown in Figure 23, at t 3, the switch S5 is turned on at zero voltage, and the switch S6, the rectifier diode D1, and the rectifier diode D4 continue to conduct; the magnetizing inductance Lm is still The output voltage is clamped, the excitation current iLm continues to increase linearly, and the resonant current iLr decreases linearly;
  • Switching mode 5 [t 4 , t 5 ]: As shown in Figure 24, at time t 4 , the resonant current iLr is equal to the excitation current iLm, and the current flowing through the rectifier diode D1 naturally crosses 0, and the secondary side rectifier diode D1, rectifier Diode D4 is turned off at zero current to avoid diode reverse recovery problems; switch S5 and switch S6 continue to conduct, and the excitation current and resonance current iLr are equal and remain unchanged;
  • Switching mode 6 [t 5 , t 6 ]: As shown in Figure 25, at t 5 , the switch S6 is turned off while the switch S5 continues to conduct; the resonance current iLr is equal to the excitation current iLm, and the secondary rectifier diode is still in reverse off-state; iLr resonant current to the output capacitor C oss1 charge switch S1, the output capacitance C oss3 discharging switch S3, S6 to switch an output charging capacitor C oss6; C oss3 when the voltage across the capacitor drops At 0 o'clock, the body diode of the switching tube S3 is turned on, which provides conditions for the switching tube S3 to realize zero voltage turn-on; at t6, the switching tube S3 realizes ZVS, and the circuit enters the second half cycle.
  • the selection of high, medium and low voltage section is divided according to the standard of gain range.
  • the input voltage range is 40 ⁇ 320VDC
  • the FBLLC frequency conversion modal gain range is 1 ⁇ 2
  • the FBLLC variable duty cycle modal gain range is 0.5 ⁇ 1
  • the HBLLC variable duty cycle modal gain range is 0.25 ⁇ 0.5
  • 180-320V belongs to the high-voltage section.
  • the wider the gain range the lower the efficiency of the circuit. Therefore, when dividing the voltage range, the span of the gain range needs to be considered, and the continuity of the gain range corresponding to the high, medium, and low voltages should be ensured after being superimposed.
  • each switching device of the converter can realize zero voltage turn-on, and the rectifier device on the secondary side can also realize zero current turn-off. There is no diode reverse recovery problem. All switches The devices are all in the working state of soft switching.
  • the invention adopts the switching control mode of variable frequency PFM and fixed frequency PWM, and is combined with full bridge and half bridge topologies, so as to cooperate with the circuit to achieve a wider voltage gain range and higher efficiency, so that the converter can be applied to a wider gain Where the scope is required.
  • the control method of the present invention has a small frequency conversion range, low requirements for magnetic components such as transformers, inductors, etc., does not have a leading bridge arm and a lagging bridge arm, and it is easy to realize the zero voltage turn-on of the primary side switch tube and the zero current turn-off of the secondary side rectifier tube Off.

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Abstract

Disclosed is a wide gain control method for a variable topology LLC resonant converter, the method being applied to a variable topology LLC resonant converter composed of an inverter circuit, an LLC resonant cavity, a transformer and a secondary rectifying and filtering output circuit. In the present invention, the range of an input voltage is divided into a low voltage section, a medium voltage section and a high voltage section respectively corresponding to three different modals; the variable topology LLC resonant converter uses variable-frequency PFM control of an FBLLC structure in the low voltage section of the input voltage, and an output voltage gain is changed by means of changing a switch frequency; the variable topology LLC resonant converter uses fixed-frequency PWM control of the FBLLC structure in the medium voltage section of the input voltage, and the output voltage gain is changed by means of changing a duty ratio of a switch tube (S1) of the inverter circuit; and the variable topology LLC resonant converter uses fixed-frequency PWM control of an HBLLC structure in the high voltage section of the input voltage, and the output voltage gain is changed by means of changing the duty ratio of the switch tube (S1) of the inverter circuit.

Description

一种变拓扑LLC谐振变换器的宽增益控制方法A wide-gain control method for LLC resonant converter with variable topology 技术领域Technical field
本发明涉及开关变换器技术领域，具体地说涉及一种变拓扑LLC谐振变换器的宽增益控制方法。The invention relates to the technical field of switching converters, in particular to a wide gain control method of a variable topology LLC resonant converter.
背景技术Background technique
随着电力电子技术的迅猛发展，开关变换器得到了广泛的应用。人们对开关变换器提出更多要求：高功率密度、高可靠性和小体积。LLC谐振变换器作为一种谐振变换器，具有低噪声、低应力、开关损耗低等诸多优势。LLC谐振变换器一般采用变频和定频控制两种方式，然而当输入电压和负载变化范围很宽时，单独采用变频控制的LLC谐振变换器的工作频率变化范围很宽，使得电路中的磁性元件设计困难，而且当电压增益较宽时，传统变频控制LLC谐振变换器的效率明显下降；而单独采用定频移相控制的LLC，由于工作频率固定因而磁性元件便于设计，但是为了使得输入电压和负载在宽范围下保证输出电压不变，需要电路在较大的移相角下工作，又会导致移相电路中的滞后桥臂难以实现软开关，所以在这种控制方式下，为了满足在最大移相角下滞后桥臂可以实现软开关的需求，移相角就会受到限制，进而导致传统定频移相控制的LLC变换器增益范围受限。总之，当LLC谐振变换器应用于超宽输入场合下时，电路不能兼顾高效率和高增益的特点。With the rapid development of power electronics technology, switching converters have been widely used. People put forward more requirements for switching converters: high power density, high reliability and small size. As a kind of resonant converter, LLC resonant converter has many advantages such as low noise, low stress and low switching loss. LLC resonant converter generally adopts frequency conversion and fixed frequency control. However, when the input voltage and load change range is wide, the operating frequency of the LLC resonant converter that uses frequency conversion control alone has a wide range of changes, which makes the magnetic components in the circuit The design is difficult, and when the voltage gain is wide, the efficiency of the traditional frequency conversion control LLC resonant converter is significantly reduced; while the LLC that uses the fixed frequency and phase shift control alone, because the operating frequency is fixed, the magnetic components are easy to design, but in order to make the input voltage and The load ensures the output voltage remains unchanged in a wide range, and the circuit needs to work at a large phase shift angle, which will also make it difficult for the lagging bridge arm in the phase shift circuit to achieve soft switching. Therefore, in this control mode, in order to meet the Under the maximum phase shift angle, the lagging bridge arm can realize the requirement of soft switching, and the phase shift angle will be limited, which in turn leads to the limitation of the gain range of the traditional fixed frequency phase shift control LLC converter. In short, when LLC resonant converter is used in ultra-wide input occasions, the circuit cannot take into account the characteristics of high efficiency and high gain.
目前，有相关研究通过变拓扑的形式拓宽LLC的增益，实现宽电压范围。由廖政伟、张雪、尤伟等人2013年发表于浙江大学学报(工学报)的《应用于超宽输入范围的变拓扑LLC电路》中，在全桥LLC(FBLLC)拓扑中找到半桥LLC(HBLLC)结构，当输入电压为低压段时采用FBLLC拓扑，当输入电压为高压段时采用HBLLC拓扑，在两种拓扑下均采用变频控制，等效电路如图1所示，图2为两种拓扑下的增益比较，由附图可以看出，通过在全桥与半桥结构间切换，可以将电路增益提高一倍，并且电路效率也得到有利提高。但是，该电路仅能提高一倍的电压增益，不能满足更宽输入电压的要求，且对于全桥/半桥结构均采用变频控制，磁性元件的设计还是较为复杂，效率受开关频率的影响较大。At present, there are related studies to broaden the gain of LLC by changing the topology to achieve a wide voltage range. In 2013 by Liao Zhengwei, Zhang Xue, You Wei and others, published in the Journal of Zhejiang University (Journal of Engineering), "Variable Topology LLC Circuit Applied to Ultra-Wide Input Range", half-bridge LLC was found in the full-bridge LLC (FBLLC) topology (HBLLC) structure, when the input voltage is in the low-voltage section, the FBLLC topology is used, when the input voltage is in the high-voltage section, the HBLLC topology is used, and frequency conversion control is used in both topologies. The equivalent circuit is shown in Figure 1, and Figure 2 is two The gain comparison under this topology can be seen from the figure, by switching between the full-bridge and half-bridge structure, the circuit gain can be doubled, and the circuit efficiency is also beneficially improved. However, this circuit can only double the voltage gain, which cannot meet the requirements of a wider input voltage, and adopts variable frequency control for the full-bridge/half-bridge structure. The design of magnetic components is still more complicated, and the efficiency is more affected by the switching frequency. Big.
发明内容Summary of the invention
本发明所要解决的技术问题是，提供一种变拓扑LLC谐振变换器的宽增益控制方法，以满足更宽输入电压的要求。The technical problem to be solved by the present invention is to provide a wide gain control method of a variable topology LLC resonant converter to meet the requirement of a wider input voltage.
本发明控制方法采用技术方案如下：一种变拓扑LLC谐振变换器的宽增益控制方法，应用于由逆变电路、LLC谐振腔、变压器和副边整流滤波输出电路组成的变拓扑LLC谐振变换器，所述逆变电路包括开关管S1、S2、S3、S4，能够实现全桥到半桥结构的切换；所述LLC谐振腔包括开关管S5、S6、谐振电感Lr、变压器励磁电感Lm和谐振电容Cr；开关管S1的漏极连于开关管S2的漏极和输入电源Vin的正端，开关管S1的源极连于开关管S3的漏极和谐振电容Cr的一端，谐振电容Cr的另一端连于谐振电感Lr的一端和开关管S5的漏极，谐振电感Lr的另一端连于励磁电感Lm的一端和变压器T原边绕组Np的1端，变压器T原边绕组Np的2端连于励磁电感Lm的另一端、开关管S2的源极、开关管S4的漏极、开关管S6的漏极，开关管S4的源极连于开关管S3的源极和输入电源Vin的负极，开关管S6的源极连于开关管S5的源极；其特征在于：分别将输入电压范围分为低、中、高三个电压段，分别对应三种不同的模态，The control method of the present invention adopts the technical scheme as follows: a wide gain control method of a variable topology LLC resonant converter, which is applied to a variable topology LLC resonant converter composed of an inverter circuit, an LLC resonant cavity, a transformer and a secondary rectifier filter output circuit , The inverter circuit includes switch tubes S1, S2, S3, S4, which can switch from full bridge to half bridge structure; the LLC resonant cavity includes switch tubes S5, S6, resonance inductance Lr, transformer magnetizing inductance Lm, and resonance Capacitor Cr; the drain of the switching tube S1 is connected to the drain of the switching tube S2 and the positive terminal of the input power supply Vin, the source of the switching tube S1 is connected to the drain of the switching tube S3 and one end of the resonant capacitor Cr, the resonant capacitor Cr The other end is connected to one end of the resonant inductor Lr and the drain of the switch S5. The other end of the resonant inductor Lr is connected to one end of the magnetizing inductance Lm and the first end of the transformer T primary winding Np, and the second end of the transformer T primary winding Np Connected to the other end of the magnetizing inductance Lm, the source of the switching tube S2, the drain of the switching tube S4, and the drain of the switching tube S6. The source of the switching tube S4 is connected to the source of the switching tube S3 and the negative electrode of the input power Vin , The source of the switching tube S6 is connected to the source of the switching tube S5; it is characterized in that the input voltage range is divided into three voltage segments, low, medium, and high, respectively, corresponding to three different modes,
所述变拓扑LLC谐振变换器在输入电压低压段采用FBLLC结构的变频PFM控制，通过改变开关频率来改变输出电压增益；The variable-topology LLC resonant converter adopts variable-frequency PFM control of the FBLLC structure in the low-voltage section of the input voltage, and changes the output voltage gain by changing the switching frequency;
所述变拓扑LLC谐振变换器在输入电压中压段采用FBLLC结构的定频PWM控制，通过改变逆变电路开关管(S1)的占空比来改变输出电压增益；The variable-topology LLC resonant converter adopts fixed-frequency PWM control of the FBLLC structure in the medium voltage section of the input voltage, and changes the output voltage gain by changing the duty cycle of the switch tube (S1) of the inverter circuit;
所述变拓扑LLC谐振变换器在输入电压高压段采用HBLLC结构的定频PWM控制，通过改变逆变电路开关管(S1)的占空比来改变输出电压增益。The variable-topology LLC resonant converter adopts the fixed frequency PWM control of the HBLLC structure in the high input voltage section, and changes the output voltage gain by changing the duty cycle of the switch tube (S1) of the inverter circuit.
具体：当输入电压位于低压段，逆变电路工作在FBLLC结构的变频PFM控制模态(FBLLC变频模态)，开关管S5～S6持续关断，开关管S1～S4保持占空比为0.5且固定，开关管S1和开关管S2互补导通，开关管S1和开关管S4同时导通、同时关断，开关管S2和开关管S3同时导通、同时关断，通过调节开关管S1～S4的开关频率大小实现输出电压V 0的控制，开关频率越小，输出电压增益越大； Specifically: When the input voltage is in the low voltage section, the inverter circuit works in the variable frequency PFM control mode of the FBLLC structure (FBLLC variable frequency mode), the switching tubes S5 to S6 are continuously turned off, and the switching tubes S1 to S4 maintain a duty cycle of 0.5 and Fixed, the switching tube S1 and the switching tube S2 are turned on complementarily, the switching tube S1 and the switching tube S4 are turned on and turned off at the same time, the switching tube S2 and the switching tube S3 are turned on and turned off at the same time, by adjusting the switching tubes S1～S4 The output voltage V 0 can be controlled by the switching frequency of, the smaller the switching frequency, the greater the output voltage gain;
当输入电压位于中压段，逆变电路工作在FBLLC结构的定频PWM控制模态(FBLLC变占空比模态)，开关管S1～S6的开关频率相等且固定，开关管S1 和开关管S5互补导通，开关管S2和开关管S6互补导通，开关管S1和开关管S4同时导通、同时关断，开关管S2和开关管S3同时导通、同时关断，开关管S1的占空比与开关管S2的占空比相等，均不大于0.5且两者相位差180°，开关管S5的占空比与开关管S6的占空比相等，均不小于0.5且两者相位差180°，通过调节开关管S1的占空比大小实现输出电压V 0的控制(开关管S1的占空比变化，双向开关的导通时间同步变化)，开关管S1的占空比越大，输出电压增益越大； When the input voltage is in the medium voltage section, the inverter circuit works in the fixed frequency PWM control mode of the FBLLC structure (FBLLC variable duty cycle mode), the switching frequencies of the switching tubes S1～S6 are equal and fixed, and the switching tube S1 and the switching tube S5 is complementarily turned on, the switching tube S2 and the switching tube S6 are complementarily turned on, the switching tube S1 and the switching tube S4 are turned on and turned off at the same time, and the switching tube S2 and the switching tube S3 are turned on and turned off at the same time. The duty cycle is equal to the duty cycle of the switching tube S2, neither is greater than 0.5 and the phase difference between the two is 180°. The duty cycle of the switching tube S5 is equal to the duty cycle of the switching tube S6, neither is less than 0.5 and both phases The difference is 180°. The output voltage V 0 is controlled by adjusting the duty cycle of the switching tube S1 (the duty cycle of the switching tube S1 changes, and the conduction time of the bidirectional switch changes synchronously). The larger the duty cycle of the switching tube S1 is , The greater the output voltage gain;
当输入电压位于高压段，逆变电路工作在HBLLC结构的定频PWM控制模态(HBLLC变占空比模态)，开关管S1～S6的开关频率相等且固定，开关管S1和开关管S5互补导通，开关管S3和开关管S6互补导通，开关管S4持续导通，开关管S2持续关断，开关管S1的占空比与开关管S3的占空比相等，均不大于0.5且两者相位差180°，开关管S5的占空比与开关管S6的占空比相等，均不小于0.5且两者相位差180°，通过调节开关管S1的占空比大小实现输出电压V 0的控制，开关管S1的占空比越大，输出电压增益越大。 When the input voltage is in the high voltage section, the inverter circuit works in the fixed frequency PWM control mode of the HBLLC structure (HBLLC variable duty cycle mode), the switching frequencies of the switching tubes S1～S6 are equal and fixed, and the switching tube S1 and the switching tube S5 Complementary conduction, the switching tube S3 and the switching tube S6 are complementarily turned on, the switching tube S4 is continuously turned on, and the switching tube S2 is continuously turned off. The duty cycle of the switching tube S1 is equal to that of the switching tube S3, and both are not greater than 0.5 And the phase difference between the two is 180°, the duty cycle of the switching tube S5 is equal to the duty cycle of the switching tube S6, both are not less than 0.5 and the phase difference between the two is 180°, the output voltage is realized by adjusting the duty cycle of the switching tube S1 For the control of V 0 , the greater the duty cycle of the switch S1, the greater the output voltage gain.
有益效果：Beneficial effects:
本发明采用变频PFM控制与定频PWM控制相结合，并通过全桥、半桥拓扑结构的切换，从而配合电路实现更宽的电压增益范围和更高的效率，使变换器可适用于有更宽增益范围要求的场合。本发明控制方法频率变换范围小，对变压器、电感等磁性元件的要求低，不存在超前桥臂和滞后桥臂，即使不结合变拓扑，该控制方法相比于单一的变频PFM控制，其电压增益范围宽、效率高。The invention adopts the combination of variable frequency PFM control and fixed frequency PWM control, and through the switching of full-bridge and half-bridge topological structures, thereby cooperating with the circuit to achieve a wider voltage gain range and higher efficiency, so that the converter can be applied to more Where a wide gain range is required. The control method of the present invention has a small frequency conversion range, low requirements on magnetic components such as transformers, inductors, and the like. There are no leading bridge arms and lagging bridge arms. Even if the variable topology is not combined, the control method is compared with single frequency conversion PFM control. Wide gain range and high efficiency.
附图说明Description of the drawings
图1为采用HBLLC拓扑时，谐振变换器的等效电路图；Figure 1 shows the equivalent circuit diagram of the resonant converter when the HBLLC topology is used;
图2为HBLLC和FBLLC增益曲线比较；Figure 2 shows the comparison of HBLLC and FBLLC gain curves;
图3为本发明较佳实施例的变拓扑LLC谐振变换器的电路原理图；Fig. 3 is a circuit schematic diagram of a variable topology LLC resonant converter according to a preferred embodiment of the present invention;
图4为本发明较佳实施例的变拓扑LLC谐振变换器的增益曲线；Fig. 4 is a gain curve of a variable topology LLC resonant converter according to a preferred embodiment of the present invention;
图5为本发明较佳实施例的变拓扑LLC谐振变换器工作在FBLLC变频模态时的主要工作波形；Fig. 5 is the main working waveform of the variable topology LLC resonant converter in the FBLLC frequency conversion mode of the preferred embodiment of the present invention;
图6～11为本发明较佳实施例的变拓扑LLC谐振变换器工作在FBLLC变频模态时各开关模态的等效电路图；6-11 are equivalent circuit diagrams of each switch mode when the variable topology LLC resonant converter of the preferred embodiment of the present invention works in the FBLLC variable frequency mode;
图12为本发明较佳实施例的变拓扑LLC谐振变换器工作在FBLLC变占空比模态时的主要工作波形；FIG. 12 is the main working waveform of the variable topology LLC resonant converter in the FBLLC variable duty cycle mode of the preferred embodiment of the present invention;
图13～18为本发明较佳实施例的变拓扑LLC谐振变换器工作在FBLLC变占空比模态时各开关模态的等效电路图；13-18 are equivalent circuit diagrams of various switching modes when the variable topology LLC resonant converter of the preferred embodiment of the present invention works in the FBLLC variable duty cycle mode;
图19为本发明较佳实施例的变拓扑LLC谐振变换器工作在HBLLC变占空比模态时的主要工作波形；FIG. 19 is the main working waveform of the variable topology LLC resonant converter in the HBLLC variable duty cycle mode of the preferred embodiment of the present invention;
图20～25为本发明较佳实施例的变拓扑LLC谐振变换器工作在HBLLC变占空比模态时各开关模态的等效电路图。20-25 are equivalent circuit diagrams of various switching modes when the variable topology LLC resonant converter of the preferred embodiment of the present invention works in the HBLLC variable duty cycle mode.
具体实施方式Detailed ways
如图3所示，本实施例的变拓扑LLC谐振变换器，包括从输入到输出依次连接的逆变电路10、LLC谐振腔20、变压器T和整流网络30。图中Vin为变换器的输入电源，Ro为变换器的输出负载R 0。 As shown in FIG. 3, the variable topology LLC resonant converter of this embodiment includes an inverter circuit 10, an LLC resonant cavity 20, a transformer T, and a rectifier network 30 that are sequentially connected from input to output. In the figure, Vin is the input power of the converter, and Ro is the output load R 0 of the converter.
逆变电路10为全桥/半桥相结合的变拓扑电路，由开关管S1、开关管S2、开关管S3、开关管S4组成。LLC谐振腔20包括谐振电感Lr、励磁电感Lm和谐振电容Cr，还增设有由开关管S5、开关管S6构成的双向开关。整流网络30由4个二极管D1-D4构成的全桥整流电路并联输出滤波电容C 0构成。 The inverter circuit 10 is a full-bridge/half-bridge combined variable topology circuit, which is composed of a switching tube S1, a switching tube S2, a switching tube S3, and a switching tube S4. The LLC resonant cavity 20 includes a resonant inductance Lr, an excitation inductance Lm and a resonant capacitor Cr, and is additionally provided with a bidirectional switch composed of a switch tube S5 and a switch tube S6. The rectifier network 30 is composed of a full-bridge rectifier circuit composed of four diodes D1-D4 in parallel with an output filter capacitor C 0 .
开关管S1的漏极连于开关管S2的漏极和输入电源Vin的正端，开关管S1的源极连于开关管S3的漏极和谐振电容Cr的一端，谐振电容Cr的另一端连于谐振电感Lr的一端和开关管S5的漏极，谐振电感Lr的另一端连于励磁电感Lm的一端和变压器T原边绕组Np的1端，变压器T原边绕组Np的2端连于励磁电感Lm的另一端、开关管S2的源极、开关管S4的漏极、开关管S6的漏极，开关管S4的源极连于开关管S3的源极和输入电源Vin的负极，开关管S6的源极连于开关管S5的源极；变压器T的副边绕组Ns的1端连于副边整流二极管D1的阳极和副边整流二极管D3的阴极，副边整流二极管D1的阴极连于副边整流二极管D2的阴极、副边输出滤波电容Co的一端和输出负载Ro的一端，输出负载Ro的另一端连于副边输出滤波电容Co的另一端、副边整流二级管D3的阳极和副边整流二级管D4的阳极，副边整流二级管D4的阴极连于副边整流二级管D2的阳极和变压器T的副边绕组Ns的2端。The drain of the switch S1 is connected to the drain of the switch S2 and the positive terminal of the input power Vin, the source of the switch S1 is connected to the drain of the switch S3 and one end of the resonant capacitor Cr, and the other end of the resonant capacitor Cr is connected One end of the resonant inductor Lr and the drain of the switch S5, the other end of the resonant inductor Lr is connected to one end of the excitation inductance Lm and one end of the transformer T primary winding Np, and the second end of the transformer T primary winding Np is connected to the excitation The other end of the inductor Lm, the source of the switching tube S2, the drain of the switching tube S4, and the drain of the switching tube S6. The source of the switching tube S4 is connected to the source of the switching tube S3 and the negative electrode of the input power supply Vin. The source of S6 is connected to the source of the switch S5; the 1 end of the secondary winding Ns of the transformer T is connected to the anode of the secondary rectifier diode D1 and the cathode of the secondary rectifier diode D3, and the cathode of the secondary rectifier diode D1 is connected to The cathode of the secondary side rectifier diode D2, one end of the secondary side output filter capacitor Co and one end of the output load Ro, the other end of the output load Ro is connected to the other end of the secondary side output filter capacitor Co and the anode of the secondary side rectifier diode D3 And the anode of the secondary rectifier diode D4, the cathode of the secondary rectifier diode D4 is connected to the anode of the secondary rectifier diode D2 and the 2 ends of the secondary winding Ns of the transformer T.
变压器原边绕组与副边绕组的1端互为同名端，变压器原边绕组与副边绕组的2端互为同名端。 Terminal 1 of the primary winding and secondary winding of the transformer are mutually homonymous terminals, and terminals 2 of the primary winding and secondary winding of the transformer are mutually homonymous terminals.
上述超宽增益范围的变拓扑LLC谐振变换器可以实现8:1的增益范围，半桥的增益范围是全桥的一半；在传统LLC谐振电路的谐振腔增设双向开关，在电路拓扑结构不变的前提下，通过控制双向开关S5、S6的导通时间，采用定频PWM控制可使电路的输出电压增益至少减小一半。如果全桥变频PFM控制的增益范围是2-1，全桥定频PWM控制的增益范围则为1-0.5，半桥定频PWM控制的增益范围就是0.5-0.25，那么由全桥拓扑变频切换为全桥拓扑PWM再切换为半桥拓扑PWM的电路增益范围就可以做到2-0.25，如附图4所示。The above-mentioned ultra-wide gain range variable topology LLC resonant converter can achieve a gain range of 8:1, and the gain range of the half bridge is half of the full bridge; a bidirectional switch is added to the resonant cavity of the traditional LLC resonant circuit, and the circuit topology is unchanged Under the premise of controlling the on-time of the bidirectional switches S5 and S6, the fixed-frequency PWM control can reduce the output voltage gain of the circuit by at least half. If the gain range of full-bridge variable-frequency PFM control is 2-1, the gain range of full-bridge fixed-frequency PWM control is 1-0.5, and the gain range of half-bridge fixed-frequency PWM control is 0.5-0.25, then the full-bridge topology is variable-frequency switching For the full-bridge topology PWM and then switch to the half-bridge topology PWM, the circuit gain range can achieve 2-0.25, as shown in Figure 4.
上述超宽增益范围的变拓扑LLC谐振变换器可以采取如下变频控制方法：The above-mentioned variable topology LLC resonant converter with ultra-wide gain range can adopt the following variable frequency control methods:
当输入电压位于低压段，逆变电路工作在FBLLC结构的变频PFM控制模态，开关管S5～S6持续关断，开关管S1～S4保持占空比为0.5且固定，开关管S1和开关管S2互补导通，开关管S1和开关管S4同时导通、同时关断，开关管S2和开关管S3同时导通、同时关断，通过调节开关管S1～S4的开关频率大小实现输出电压V 0的控制，开关频率越小，输出电压增益越大； When the input voltage is in the low-voltage section, the inverter circuit works in the variable frequency PFM control mode of the FBLLC structure, the switching tubes S5～S6 are continuously turned off, the switching tubes S1～S4 keep the duty cycle at 0.5 and fixed, the switching tube S1 and the switching tube S2 is complementarily turned on, the switching tube S1 and the switching tube S4 are turned on and off at the same time, and the switching tube S2 and the switching tube S3 are turned on and off at the same time. The output voltage V is achieved by adjusting the switching frequency of the switching tubes S1～S4. 0 control, the smaller the switching frequency, the greater the output voltage gain;
上述超宽增益范围的变拓扑LLC谐振变换器可以采取如下定频PWM控制方法：The above-mentioned variable topology LLC resonant converter with ultra-wide gain range can adopt the following fixed frequency PWM control method:
当输入电压位于中压段，逆变电路工作在FBLLC结构的定频PWM控制模态(FBLLC变占空比模态)，开关管S1～S6的开关频率相等且固定，开关管S1和开关管S5互补导通，开关管S2和开关管S6互补导通，开关管S1和开关管S4同时导通、同时关断，开关管S2和开关管S3同时导通、同时关断，开关管S1的占空比与开关管S2的占空比相等，均不大于0.5且两者相位差180°，开关管S5的占空比与开关管S6的占空比相等，均不小于0.5且两者相位差180°，通过调节开关管S1的占空比大小实现输出电压V 0的控制，开关管S1的占空比越大，输出电压增益越大； When the input voltage is in the medium voltage range, the inverter circuit works in the fixed frequency PWM control mode of the FBLLC structure (FBLLC variable duty cycle mode), the switching frequencies of the switching tubes S1～S6 are equal and fixed, and the switching tube S1 and the switching tube S5 is complementarily turned on, the switching tube S2 and the switching tube S6 are complementarily turned on, the switching tube S1 and the switching tube S4 are turned on and turned off at the same time, and the switching tube S2 and the switching tube S3 are turned on and turned off at the same time. The duty cycle is equal to the duty cycle of the switch S2, neither is greater than 0.5 and the phase difference between the two is 180°. The duty cycle of the switch S5 is equal to the duty cycle of the switch S6, neither is less than 0.5 and both phases The difference is 180°, the output voltage V 0 is controlled by adjusting the duty ratio of the switching tube S1. The larger the duty ratio of the switching tube S1, the greater the output voltage gain;
当输入电压位于高压段，逆变电路工作在HBLLC结构的定频PWM控制模态(FBLLC变占空比模态)，开关管S1～S6的开关频率相等且固定，开关管S1和开关管S5互补导通，开关管S3和开关管S6互补导通，开关管S4持续导通，开关管S2持续关断，开关管S1的占空比与开关管S3的占空比相等，均不大于 0.5且两者相位差180°，开关管S5的占空比与开关管S6的占空比相等，均不小于0.5且两者相位差180°，通过调节开关管S1的占空比大小实现输出电压V 0的控制，开关管S1的占空比越大，输出电压增益越大。 When the input voltage is in the high voltage section, the inverter circuit works in the fixed frequency PWM control mode of the HBLLC structure (FBLLC variable duty cycle mode), the switching frequencies of the switching tubes S1～S6 are equal and fixed, and the switching tube S1 and the switching tube S5 Complementary conduction, the switching tube S3 and the switching tube S6 are complementarily turned on, the switching tube S4 is continuously turned on, and the switching tube S2 is continuously turned off. The duty cycle of the switching tube S1 is equal to that of the switching tube S3, and both are not greater than 0.5 And the phase difference between the two is 180°, the duty cycle of the switching tube S5 is equal to the duty cycle of the switching tube S6, both are not less than 0.5 and the phase difference between the two is 180°, the output voltage is realized by adjusting the duty cycle of the switching tube S1 For the control of V 0 , the greater the duty cycle of the switch S1, the greater the output voltage gain.
在具体实施时，开关管S1和开关管S5的开关信号之间必须设置合理的死区时间以实现开关管S1、开关管S4、开关管S5的软开关；开关管S2和开关管S6的开关信号之间必须设置合理的死区时间以实现开关管S2、开关管S3、开关管S6的软开关。Coss1～Coss6分别表示至六只开关管S1～S6的输出电容。In specific implementation, a reasonable dead time must be set between the switching signals of the switching tube S1 and the switching tube S5 to realize the soft switching of the switching tube S1, the switching tube S4, and the switching tube S5; the switching of the switching tube S2 and the switching tube S6 A reasonable dead time must be set between the signals to realize the soft switching of the switching tube S2, the switching tube S3, and the switching tube S6. Coss1 to Coss6 respectively represent the output capacitance to the six switching tubes S1 to S6.
下面结合附图3，具体说明变拓扑LLC谐振变换器采用变频PFM控制与定频PWM控制时的工作过程。With reference to Figure 3, the working process of the variable-topology LLC resonant converter using variable-frequency PFM control and fixed-frequency PWM control will be described in detail below.
在本实施例中，参数选取如下：Lr＝1.672uH，Lm＝6.668uH，Cr＝168.3nF，输入电压范围40～320VDC，变压器原副边匝比为3:1。In this embodiment, the parameters are selected as follows: Lr=1.672uH, Lm=6.668uH, Cr=168.3nF, the input voltage range is 40-320VDC, and the primary and secondary turns ratio of the transformer is 3:1.
当变换器的输入电压在40V～80V时，变换器工作在FBLLC变频模态下，此时开关管S5、S6保持关断状态，图5为这种谐振变换器采用FBLLC变频模态时的主要工作波形图，Vgs1/4为开关管S1、S4的驱动信号，Vgs2/3为开关管S2、S3的驱动信号，Vc、iLr、iLm、i 0分别表示Cr两端电压、通过Lr的电流、通过Lm的电流和通过电阻R 0的电流。从图5可以看出，本发明输出电流I 0变化平缓，器件应力小。变换器在半个周期内共有六种开关模态，分别如图6～11所示(LLC谐振变换器后半个周期与前半个周期工作模态对称从波形图也可以看出，一般对LLC谐振变换器的描述只描述半个周期即可)。 When the input voltage of the converter is between 40V and 80V, the converter works in the FBLLC frequency conversion mode. At this time, the switches S5 and S6 remain in the off state. Figure 5 shows the main features of this resonant converter when the FBLLC frequency conversion mode is used. Working waveform diagram, Vgs1/4 is the driving signal of the switching tubes S1 and S4, Vgs2/3 is the driving signal of the switching tubes S2 and S3, Vc, iLr, iLm, and i 0 respectively represent the voltage across Cr, the current through Lr, The current through Lm and the current through resistor R 0 . It can be seen from FIG. 5 that the output current I 0 of the present invention changes smoothly, and the device stress is small. The converter has six switching modes in a half cycle, as shown in Figures 6 to 11 (the working modes of the second half cycle and the first half cycle of the LLC resonant converter are symmetrical. It can also be seen from the waveform diagram. Generally speaking, the LLC resonant converter has six switching modes. The description of the resonant converter only describes half a cycle).
开关模态1[t 0，t 1]：如附图6所示，t0时刻，开关管S1、开关管S4零电压开通；整流二极管D1、整流二极管D4导通，流经二极管的电流与谐振电流和励磁电流的差值成正比；励磁电感Lm两端电压被输出钳位至nV O(n为变压器匝比)；原边谐振电感Lr与谐振电容Cr参与谐振，谐振电流iLr为标准正弦波且为负值，励磁电感电流iLm线性增加，但小于谐振电流iLr； Switch mode 1 [t 0 , t 1 ]: As shown in Figure 6, at t0, the switching tube S1 and the switching tube S4 are turned on at zero voltage; the rectifier diode D1 and the rectifier diode D4 are turned on, and the current flowing through the diode is in resonance The current is proportional to the difference between the excitation current; the voltage across the excitation inductance Lm is output clamped to nV O (n is the transformer turns ratio); the primary resonant inductance Lr and the resonant capacitor Cr participate in resonance, and the resonant current iLr is a standard sine wave If it is a negative value, the magnetizing inductance current iLm increases linearly, but it is less than the resonant current iLr;
开关模态2[t 1，t 2]：如附图7所示，在t 1时刻，谐振电流iLr过零点；整流二极管D1、整流二极管D4继续导通；励磁电感Lm两端电压被输出钳位至nVo；原边谐振电感Lr与谐振电容Cr参与谐振，谐振电流iLr为标准正弦波且为正值，励磁电感电流iLm线性增加，仍小于谐振电流iLr； Switching mode 2 [t 1 , t 2 ]: As shown in Figure 7, at t 1 , the resonant current iLr crosses the zero point; the rectifier diode D1 and the rectifier diode D4 continue to conduct; the voltage across the excitation inductor Lm is clamped by the output Bit to nVo; the primary side resonant inductance Lr and resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and is positive, and the magnetizing inductor current iLm increases linearly and is still less than the resonant current iLr;
开关模态3[t 2，t 3]：如附图8所示，在t 2时刻开关管S1、开关管S4保持导通，励磁电流ILm过零点变为正方向；整流二极管D1、整流二极管D4继续导通；励磁电感Lm两端电压被输出钳位至nVo；原边谐振电感Lr与谐振电容Cr参与谐振，谐振电流iLr为标准正弦波且为正值，励磁电感电流iLm线性增加，仍小于谐振电流iLr； Switching mode 3 [t 2 , t 3 ]: As shown in Figure 8, at t 2 the switching tube S1 and the switching tube S4 remain conductive, and the zero-crossing point of the excitation current ILm changes to the positive direction; the rectifier diode D1, the rectifier diode D4 continues to conduct; the voltage across the magnetizing inductance Lm is output clamped to nVo; the primary resonant inductance Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and a positive value, and the magnetizing inductance current iLm increases linearly, but still Less than the resonant current iLr;
开关模态4[t 3，t 4]：如附图9所示，在t3时刻，励磁电流ILm与谐振电流ILr相等，励磁电感不再被钳位(变压器无能量传输)；流过整流二极管D1的电流自然过0，副边整流二极管D1、D4零电流关断，避免二极管反向恢复问题；原边谐振电感Lr、谐振电容Cr、励磁电感Lm一起参与谐振，负载能量完全由输出电容Co提供； Switch mode 4 [t 3 , t 4 ]: As shown in Figure 9, at t3, the excitation current ILm is equal to the resonant current ILr, and the excitation inductance is no longer clamped (the transformer has no energy transmission); it flows through the rectifier diode The current of D1 naturally crosses 0, and the secondary side rectifier diodes D1 and D4 are turned off with zero current to avoid the diode reverse recovery problem; the primary side resonant inductance Lr, resonant capacitor Cr, and magnetizing inductance Lm participate in resonance together, and the load energy is completely driven by the output capacitor Co provide;
开关模态5[t 4，t 5]：如附图10所示，该时段为死区时间，励磁电流和谐振电流iLr相等且保持不变，副边整流二极管仍处于反向截止状态；所有功率管关断；原边谐振电感Lr、谐振电容Cr、励磁电感Lm一起参与谐振；谐振电流iLr给开关管S1、开关管S4的输出电容C oss1、C oss4充电，给开关管S2、开关管S3的输出电容C oss2、C oss3放电，负载能量完全由输出电容Co提供； Switching mode 5 [t 4 , t 5 ]: As shown in Figure 10, this period is the dead time, the excitation current and the resonant current iLr are equal and remain unchanged, and the secondary side rectifier diode is still in the reverse cut-off state; all The power tube is turned off; the primary resonance inductance Lr, the resonance capacitor Cr, and the excitation inductance Lm participate in resonance together; the resonance current iLr charges the output capacitors C oss1 and C oss4 of the switching tube S1 and the switching tube S4, and charging the switching tube S2 and the switching tube The output capacitors C oss2 and C oss3 of S3 are discharged, and the load energy is completely provided by the output capacitor Co;
开关模态6[t 5，t 6]：如附图11所示，该时段仍为死区时间，开关管S2、开关管S3的输出电容C oss2、C oss3两端电压降到0时，开关管S2、开关管S3的体二极管导通，为开关管S2、开关管S3实现零电压开通提供条件；t6时刻，开关管S2、开关管S3实现ZVS，电路进入后半个周期。 Switch mode 6 [t 5 , t 6 ]: As shown in Figure 11, this period is still dead time, when the voltage across the output capacitors C oss2 and C oss3 of the switch S2 and S3 drops to 0, The body diodes of the switching tube S2 and the switching tube S3 are turned on to provide conditions for the switching tube S2 and the switching tube S3 to realize zero voltage turn-on; at t6, the switching tube S2 and the switching tube S3 realize ZVS, and the circuit enters the second half cycle.
当变换器的输入电压在80V～160V时，变换器工作在FBLLC变占空比模态下，图12为这种谐振变换器采用定频PWM控制时的主要工作波形图，Vgs1/4为开关管S1、S4的驱动信号，Vgs2/3为开关管S2、S3的驱动信号，Vgs5为开关管S5的驱动信号，Vgs6为开关管S6的驱动信号，Vc、iLr、iLm、i 0分别表示Cr两端电压、通过Lr的电流、通过Lm的电流和通过电阻R 0的电流。从图4可以看出，本发明输出电流I 0变化平缓，器件应力小。从图11可以看出，本发明输出电流I 0变化平缓，器件应力小。变换器在本半个周期内同样有六种开关模态，分别如图13～18所示。 When the input voltage of the converter is between 80V and 160V, the converter works in the FBLLC variable duty cycle mode. Figure 12 shows the main operating waveforms of this resonant converter when fixed-frequency PWM control is used. Vgs1/4 is the switch The driving signal of the tubes S1 and S4, Vgs2/3 is the driving signal of the switching tubes S2 and S3, Vgs5 is the driving signal of the switching tube S5, Vgs6 is the driving signal of the switching tube S6, Vc, iLr, iLm, and i 0 respectively represent Cr The voltage across the terminals, the current through Lr, the current through Lm, and the current through the resistor R 0 . It can be seen from Fig. 4 that the output current I 0 of the present invention changes smoothly and the device stress is small. It can be seen from FIG. 11 that the output current I 0 of the present invention changes smoothly and the device stress is small. The converter also has six switching modes in this half cycle, as shown in Figures 13-18.
开关模态1[t 0，t 1]：如附图13所示，在t 0时刻前，开关管S6已导通，开关管S5关断，其体二极管承受反向电压而反向截至；t0时刻，开关管S1、开 关管S4零电压开通；整流二极管D1、整流二极管D4导通，流经二极管的电流与谐振电流和励磁电流的差值成正比；励磁电感Lm两端电压被输出钳位至nV O(n为变压器匝比)；原边谐振电感Lr与谐振电容Cr参与谐振，谐振电流iLr为标准正弦波且为负值，励磁电感电流iLm线性增加，但小于谐振电流iLr； Switch mode 1 [t 0 , t 1 ]: As shown in Figure 13, before time t 0 , the switch S6 has been turned on, the switch S5 is turned off, and its body diode bears a reverse voltage and reversely cuts off; At t0, the switching tube S1 and the switching tube S4 are turned on at zero voltage; the rectifier diode D1 and the rectifier diode D4 are turned on, and the current flowing through the diode is proportional to the difference between the resonance current and the excitation current; the voltage across the excitation inductance Lm is output clamped Bit to nV O (n is the transformer turns ratio); the primary side resonant inductor Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and is negative, and the magnetizing inductor current iLm increases linearly, but is less than the resonant current iLr;
开关模态2[t 1，t 2]：如附图14所示，在t 1时刻，谐振电流iLr过零点；整流二极管D1、整流二极管D4继续导通；励磁电感Lm两端电压被输出钳位至nVO；原边谐振电感Lr与谐振电容Cr参与谐振，谐振电流iLr为标准正弦波且为正值，励磁电感电流iLm线性增加，但小于谐振电流iLr； Switching mode 2 [t 1 , t 2 ]: As shown in Figure 14, at t 1 , the resonant current iLr crosses the zero point; the rectifier diode D1 and the rectifier diode D4 continue to conduct; the voltage across the excitation inductor Lm is clamped by the output Bit to nVO; the primary side resonant inductor Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and is positive, and the excitation inductor current iLm increases linearly, but is less than the resonant current iLr;
开关模态3[t 2，t 3]：如附图15所示，在t 2时刻开关管S1、开关管S4关断，谐振电流iLr仍大于励磁电感电流iLm，整流二极管D1、整流二极管D4继续导通；谐振电流iLr给开关管S1、开关管S4的输出电容C oss1、C oss4充电，给开关管S2、开关管S3输出电容C oss2、C oss3放电，给开关管S5的输出电容C oss5放电；当电容C oss5两端电压降到零时，开关管S5的体二极管管导通，为开关管S5实现零电压开通提供条件； Switch mode 3 [t 2 , t 3 ]: As shown in Figure 15, at t 2 the switch S1 and the switch S4 are turned off, the resonance current iLr is still greater than the excitation inductance current iLm, the rectifier diode D1 and the rectifier diode D4 continues to conduct; resonant current iLr to switch Sl, the switch output capacitance C oss1 S4 of, C oss4 charge, to the switch S2, the switch S3 output capacitor C oss2, C oss3 discharge, a switch S5, the output capacitor C Oss5 discharges; when the voltage across the capacitor C oss5 drops to zero, the body diode of the switching tube S5 is turned on, which provides conditions for the switching tube S5 to realize zero voltage turn-on;
开关模态4[t 3，t 4]：如附图16所示，在t 3时刻开关管S5零电压开通，开关管S6和整流二极管D1、整流二极管D4继续导通；励磁电感Lm依然被输出电压钳位，励磁电流iLm继续线性增加，谐振电流iLr线性下降； Switch Mode 4 [t 3, t 4] : As shown in FIG. 16, the switch S5 ZVS at time t 3, the switch S6 and the rectifying diode D1, a rectifier diode D4 is still turned on; magnetizing inductance Lm is still The output voltage is clamped, the excitation current iLm continues to increase linearly, and the resonant current iLr decreases linearly;
开关模态5[t 4，t 5]：如附图17所示，在t 4时刻，谐振电流iLr等于励磁电流iLm，流过整流二极管D1的电流自然过0，副边整流二极管D1、整流二极管D4零电流关断，避免二极管反向恢复问题；开关管S5和开关管S6继续导通，励磁电流和谐振电流iLr相等且保持不变； Switching mode 5 [t 4 , t 5 ]: As shown in Figure 17, at t 4 , the resonant current iLr is equal to the excitation current iLm, the current flowing through the rectifier diode D1 naturally passes through 0, and the secondary side rectifier diode D1 is rectified Diode D4 is turned off at zero current to avoid diode reverse recovery problems; switch S5 and switch S6 continue to conduct, and the excitation current and resonance current iLr are equal and remain unchanged;
开关模态6[t 5，t 6]：如附图18所示，t 5时刻，开关管S6关断而开关管S5继续导通；谐振电流iLr等于励磁电流iLm，副边整流二极管仍处于反向截止状态；谐振电流iLr给开关管S1、开关管S4的输出电容C oss1、C oss4充电，给开关管S2、开关管S3的输出电容C oss2、C oss3放电，给开关管S6的输出电容C oss6充电；当电容C oss2、C oss3两端电压降到0时，开关管S2、开关管S3的体二极管管导通，为开关管S2、开关管S3实现零电压开通提供条件；t6时刻，开关管S2、开关管S3实现ZVS，电路进入后半个周期。 Switching mode 6 [t 5 , t 6 ]: As shown in Figure 18, at t 5 , the switch S6 is turned off while the switch S5 continues to conduct; the resonant current iLr is equal to the excitation current iLm, and the secondary rectifier diode is still in reverse off-state; resonant current iLr to switch Sl, switch the output capacitor C oss1 S4, C oss4 charge, to the switch S2, the switch of the output capacitor C oss2 S3, C oss3 discharge, to the output of switch S6 The capacitor C oss6 is charged; when the voltage across the capacitors C oss2 and C oss3 drops to 0, the body diodes of the switch S2 and S3 are turned on, providing conditions for the switch S2 and S3 to achieve zero voltage turn-on; t6 At the moment, the switch tube S2 and the switch tube S3 realize ZVS, and the circuit enters the second half cycle.
当变换器的输入电压在160V～320V时，变换器工作在HBLLC变占空比模态下，即开关管S2恒定关断，开关管S4恒定导通，图19为这种谐振变换器采用定频PWM控制时的主要工作波形图，Vgs1～Vgs6分别为开关管S1～S6的驱动信号，Vc、iLr、iLm、i 0分别表示Cr两端电压、通过Lr的电流、通过Lm的电流和通过电阻R 0的电流。从图19可以看出，本发明输出电流I 0变化平缓，器件应力小。变化器在本半个周期内同样有六种开关模态，分别如图20～25所示。 When the input voltage of the converter is between 160V and 320V, the converter works in the HBLLC variable duty cycle mode, that is, the switching tube S2 is constantly turned off, and the switching tube S4 is constantly turned on. Figure 19 shows that this kind of resonant converter adopts constant The main working waveforms during PWM control. Vgs1～Vgs6 are the driving signals of the switching tubes S1～S6, and Vc, iLr, iLm, and i 0 respectively represent the voltage across Cr, the current through Lr, the current through Lm, and the current through Lm. The current of the resistor R 0. It can be seen from Fig. 19 that the output current I 0 of the present invention changes smoothly and the device stress is small. The variator also has six switching modes in this half cycle, as shown in Figures 20-25.
开关模态1[t 0，t 1]：如附图20所示，在t 0时刻前，开关管S6已导通，开关管S5关断，其体二极管承受反向电压而反向截至；t 0时刻，开关管S1零电压开通；整流二极管D1、整流二极管D4导通，流经二极管的电流与谐振电流和励磁电流的差值成正比；励磁电感Lm两端电压被输出钳位至nV O；原边谐振电感Lr与谐振电容Cr参与谐振，谐振电流iLr为标准正弦波且为负值，励磁电感电流iLm线性增加，但小于谐振电流iLr； Switch mode 1 [t 0 , t 1 ]: As shown in Figure 20, before time t 0 , the switching tube S6 is turned on, the switching tube S5 is turned off, and its body diode bears the reverse voltage and reversely stops; At t 0 , the switching tube S1 is turned on with zero voltage; the rectifier diode D1 and the rectifier diode D4 are turned on, and the current flowing through the diode is proportional to the difference between the resonant current and the excitation current; the voltage across the excitation inductance Lm is output clamped to nV O ; The primary side resonant inductance Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and is negative, and the excitation inductance current iLm increases linearly, but is less than the resonant current iLr;
开关模态2[t 1，t 2]：如附图21所示，在t 1时刻，谐振电流iLr过零点；整流二极管D1继续导通；励磁电感Lm两端电压被输出钳位至nV O；原边谐振电感Lr与谐振电容Cr参与谐振，谐振电流iLr为标准正弦波且为正值，励磁电感电流iLm线性增加，但小于谐振电流iLr； Switching mode 2 [t 1 , t 2 ]: As shown in Figure 21, at time t 1 , the resonant current iLr crosses the zero point; the rectifier diode D1 continues to conduct; the voltage across the magnetizing inductance Lm is output clamped to nV O ; The primary side resonant inductance Lr and the resonant capacitor Cr participate in resonance, the resonant current iLr is a standard sine wave and a positive value, and the excitation inductance current iLm increases linearly, but is less than the resonant current iLr;
开关模态3[t 2，t 3]：如附图22所示，在t 2时刻开关管S1、开关管S4关断，谐振电流iLr仍大于励磁电感电流iLm，整流二极管D1、整流二极管D4继续导通；谐振电流iLr给开关管S1的输出电容C oss1充电，给开关管S3输出电容C oss3放电，给开关管S5的输出电容C oss5放电；当电容C oss5两端电压降到零时，开关管S5的体二极管管导通，为开关管S5实现零电压开通提供条件。 Switching mode 3 [t 2 , t 3 ]: As shown in Figure 22, at t 2 the switch S1 and S4 are turned off, the resonance current iLr is still greater than the magnetizing inductance current iLm, the rectifier diode D1, and the rectifier diode D4 continues to conduct; iLr resonant current to the output capacitor C oss1 charge switch S1, the switch S3 to the output capacitor C oss3 discharge the output capacitance C oss5 discharge switch S5; when the voltage across the capacitor drops to zero C oss5 , The body diode of the switching tube S5 is turned on, which provides conditions for the switching tube S5 to realize zero voltage turn-on.
开关模态4[t 3，t 4]：如附图23所示，在t 3时刻开关管S5零电压开通，开关管S6和整流二极管D1、整流二极管D4继续导通；励磁电感Lm依然被输出电压钳位，励磁电流iLm继续线性增加，谐振电流iLr线性下降； Switching mode 4 [t 3 , t 4 ]: As shown in Figure 23, at t 3, the switch S5 is turned on at zero voltage, and the switch S6, the rectifier diode D1, and the rectifier diode D4 continue to conduct; the magnetizing inductance Lm is still The output voltage is clamped, the excitation current iLm continues to increase linearly, and the resonant current iLr decreases linearly;
开关模态5[t 4，t 5]：如附图24所示，在t 4时刻，谐振电流iLr等于励磁电流iLm，流过整流二极管D1的电流自然过0，副边整流二极管D1、整流二极管D4零电流关断，避免二极管反向恢复问题；开关管S5和开关管S6继续导通，励磁电流和谐振电流iLr相等且保持不变； Switching mode 5 [t 4 , t 5 ]: As shown in Figure 24, at time t 4 , the resonant current iLr is equal to the excitation current iLm, and the current flowing through the rectifier diode D1 naturally crosses 0, and the secondary side rectifier diode D1, rectifier Diode D4 is turned off at zero current to avoid diode reverse recovery problems; switch S5 and switch S6 continue to conduct, and the excitation current and resonance current iLr are equal and remain unchanged;
开关模态6[t 5，t 6]：如附图25所示，t 5时刻，开关管S6关断而开关管S5继续导通；谐振电流iLr等于励磁电流iLm，副边整流二极管仍处于反向截止状态；谐振电流iLr给开关管S1的输出电容C oss1充电，给开关管S3的输出电容C oss3放电，给开关管S6的输出电容C oss6充电；当电容C oss3两端电压降到0时，开关管S3的体二极管管导通，为开关管S3实现零电压开通提供条件；t6时刻，开关管S3实现ZVS，电路进入后半个周期。 Switching mode 6 [t 5 , t 6 ]: As shown in Figure 25, at t 5 , the switch S6 is turned off while the switch S5 continues to conduct; the resonance current iLr is equal to the excitation current iLm, and the secondary rectifier diode is still in reverse off-state; iLr resonant current to the output capacitor C oss1 charge switch S1, the output capacitance C oss3 discharging switch S3, S6 to switch an output charging capacitor C oss6; C oss3 when the voltage across the capacitor drops At 0 o'clock, the body diode of the switching tube S3 is turned on, which provides conditions for the switching tube S3 to realize zero voltage turn-on; at t6, the switching tube S3 realizes ZVS, and the circuit enters the second half cycle.
高、中、低电压段的选择以增益范围为标准来划分。例如，输入电压范围为40～320VDC，若FBLLC变频模态增益范围是1～2，则FBLLC变占空比模态增益范围为0.5～1，HBLLC变占空比模态增益范围即为0.25～0.5，按40V(最低输入电压)时的增益取2，那么工作在FBLLC定频模态的电压段(低压段)的范围就是40～80V(80＝40*2/1)，工作在FBLLC变频模态的电压段(中压段)的范围就是80～160V(160＝80*1/0.5)，180-320V就属于高压段。通常增益范围越宽电路效率越低，因此对电压段范围划分时，需考虑增益范围的跨度，并保证高、中、低电压对应的增益范围叠加后具有连续性。The selection of high, medium and low voltage section is divided according to the standard of gain range. For example, the input voltage range is 40～320VDC, if the FBLLC frequency conversion modal gain range is 1～2, the FBLLC variable duty cycle modal gain range is 0.5～1, and the HBLLC variable duty cycle modal gain range is 0.25～ 0.5, take 2 as the gain at 40V (lowest input voltage), then the range of the voltage section (low voltage section) working in FBLLC fixed frequency mode is 40～80V (80=40*2/1), working in FBLLC frequency conversion The range of the modal voltage section (medium voltage section) is 80-160V (160=80*1/0.5), and 180-320V belongs to the high-voltage section. Generally, the wider the gain range, the lower the efficiency of the circuit. Therefore, when dividing the voltage range, the span of the gain range needs to be considered, and the continuity of the gain range corresponding to the high, medium, and low voltages should be ensured after being superimposed.
根据上述变换器的工作过程的描述可知，该变换器各开关器件均可以实现零电压开通，副边的整流器件也都可以实现零电流关断，不存在二极管反向恢复的问题，所有的开关器件都是软开关的工作状态。According to the description of the working process of the above converter, each switching device of the converter can realize zero voltage turn-on, and the rectifier device on the secondary side can also realize zero current turn-off. There is no diode reverse recovery problem. All switches The devices are all in the working state of soft switching.
本发明采用变频PFM与定频PWM切换控制的方式，并与全桥和半桥拓扑结合，从而配合电路实现更宽的电压增益范围和更高的效率，使变换器可适用于有更宽增益范围要求的场合。本发明控制方法的频率变换范围小，对变压器、电感等磁性元件的要求低，不存在超前桥臂和滞后桥臂，易实现原边开关管的零电压开通与副边整流管的零电流关断。The invention adopts the switching control mode of variable frequency PFM and fixed frequency PWM, and is combined with full bridge and half bridge topologies, so as to cooperate with the circuit to achieve a wider voltage gain range and higher efficiency, so that the converter can be applied to a wider gain Where the scope is required. The control method of the present invention has a small frequency conversion range, low requirements for magnetic components such as transformers, inductors, etc., does not have a leading bridge arm and a lagging bridge arm, and it is easy to realize the zero voltage turn-on of the primary side switch tube and the zero current turn-off of the secondary side rectifier tube Off.
以上实施例的说明只是用于帮助理解本申请的发明构思，并不用以限制本发明，对于本技术领域的普通技术人员来说，凡在不脱离本发明原理的前提下，所作的任何修改、等同替换、改进等，均应包含在本发明的保护范围之内。The description of the above embodiments is only used to help understand the inventive concept of the present application, and is not used to limit the present invention. For those of ordinary skill in the art, any modifications and changes made without departing from the principle of the present invention Equivalent replacements, improvements, etc., should all be included in the protection scope of the present invention.

Claims (4)

  1. 一种变拓扑LLC谐振变换器的宽增益控制方法，应用于由逆变电路、LLC谐振腔、变压器和副边整流滤波输出电路组成的变拓扑LLC谐振变换器，所述逆变电路包括开关管S1、S2、S3、S4，所述LLC谐振腔包括开关管S5、S6、谐振电感Lr、变压器励磁电感Lm和谐振电容Cr；开关管S1的漏极连于开关管S2的漏极和输入电源Vin的正端，开关管S1的源极连于开关管S3的漏极和谐振电容Cr的一端，谐振电容Cr的另一端连于谐振电感Lr的一端和开关管S5的漏极，谐振电感Lr的另一端连于励磁电感Lm的一端和变压器T原边绕组Np的第一端，变压器T原边绕组Np的第二端连于励磁电感Lm的另一端、开关管S2的源极、开关管S4的漏极、开关管S6的漏极，开关管S4的源极连于开关管S3的源极和输入电源Vin的负极，开关管S6的源极连于开关管S5的源极；其特征在于：分别将输入电压范围分为低、中、高三个电压段，分别对应三种不同的模态，A wide-gain control method for a variable-topology LLC resonant converter, which is applied to a variable-topology LLC resonant converter composed of an inverter circuit, an LLC resonant cavity, a transformer, and a secondary-side rectification filter output circuit. The inverter circuit includes a switch tube S1, S2, S3, S4, the LLC resonant cavity includes switching tubes S5, S6, resonance inductance Lr, transformer magnetizing inductance Lm, and resonance capacitor Cr; the drain of switching tube S1 is connected to the drain of switching tube S2 and the input power supply The positive terminal of Vin, the source of the switch S1 is connected to the drain of the switch S3 and one end of the resonant capacitor Cr, the other end of the resonant capacitor Cr is connected to one end of the resonant inductor Lr and the drain of the switch S5, the resonant inductor Lr The other end is connected to one end of the excitation inductance Lm and the first end of the transformer T primary winding Np, the second end of the transformer T primary winding Np is connected to the other end of the excitation inductance Lm, the source of the switching tube S2, the switching tube The drain of S4, the drain of the switching tube S6, the source of the switching tube S4 is connected to the source of the switching tube S3 and the negative electrode of the input power supply Vin, and the source of the switching tube S6 is connected to the source of the switching tube S5; its characteristics The point is: The input voltage range is divided into three voltage segments, low, medium, and high, respectively, corresponding to three different modes,
    所述变拓扑LLC谐振变换器在输入电压低压段采用FBLLC结构的变频PFM控制，通过改变开关频率来改变输出电压增益；The variable-topology LLC resonant converter adopts variable-frequency PFM control of the FBLLC structure in the low-voltage section of the input voltage, and changes the output voltage gain by changing the switching frequency;
    所述变拓扑LLC谐振变换器在输入电压中压段采用FBLLC结构的定频PWM控制，通过改变逆变电路开关管(S1)的占空比来改变输出电压增益；The variable-topology LLC resonant converter adopts fixed-frequency PWM control of the FBLLC structure in the medium voltage section of the input voltage, and changes the output voltage gain by changing the duty cycle of the switch tube (S1) of the inverter circuit;
    所述变拓扑LLC谐振变换器在输入电压高压段采用HBLLC结构的定频PWM控制，通过改变逆变电路开关管(S1)的占空比来改变输出电压增益。The variable-topology LLC resonant converter adopts the fixed frequency PWM control of the HBLLC structure in the high input voltage section, and changes the output voltage gain by changing the duty cycle of the switch tube (S1) of the inverter circuit.
  2. 根据权利要求1所述的变拓扑LLC谐振变换器的宽增益控制方法，其特征在于：The wide gain control method of a variable topology LLC resonant converter according to claim 1, characterized in that:
    当输入电压位于低压段，逆变电路工作在FBLLC结构的变频PFM控制模态(FBLLC变频模态)，开关管S5～S6持续关断，开关管S1～S4保持占空比为0.5且固定，开关管S1和开关管S2互补导通，开关管S1和开关管S4同时导通、同时关断，开关管S2和开关管S3同时导通、同时关断，通过调节开关管S1～S4的开关频率大小实现输出电压V 0的控制，开关频率越小，输出电压增益越大。 When the input voltage is in the low voltage section, the inverter circuit works in the frequency conversion PFM control mode of the FBLLC structure (FBLLC frequency conversion mode), the switching tubes S5 ~ S6 are continuously turned off, and the switching tubes S1 ~ S4 maintain a fixed duty cycle of 0.5. The switching tube S1 and the switching tube S2 conduct complementary conduction, the switching tube S1 and the switching tube S4 are turned on and turned off at the same time, and the switching tube S2 and the switching tube S3 are turned on and turned off at the same time, by adjusting the switching of the switching tubes S1～S4 The frequency realizes the control of the output voltage V 0. The smaller the switching frequency, the greater the output voltage gain.
  3. 根据权利要求1所述的变拓扑LLC谐振变换器的宽增益控制方法，其特征在于：The wide gain control method of a variable topology LLC resonant converter according to claim 1, characterized in that:
    当输入电压位于中压段，逆变电路工作在FBLLC结构的定频PWM控制模态(FBLLC变占空比模态)，开关管S1～S6的开关频率相等且固定，开关管S1和开关管S5互补导通，开关管S2和开关管S6互补导通，开关管S1和开关管S4同时导通、同时关断，开关管S2和开关管S3同时导通、同时关断，开关管S1的占空比与开关管S2的占空比相等，均不大于0.5且两者相位差180°，开关管S5的占空比与开关管S6的占空比相等，均不小于0.5且两者相位差180°，通过调节开关管S1的占空比大小实现输出电压V 0的控制(开关管S1的占空比变化，双向开关的导通时间同步变化)，开关管S1的占空比越大，输出电压增益越大。 When the input voltage is in the medium voltage section, the inverter circuit works in the fixed frequency PWM control mode of the FBLLC structure (FBLLC variable duty cycle mode), the switching frequencies of the switching tubes S1 to S6 are equal and fixed, and the switching tube S1 and the switching tube S5 is complementarily turned on, the switching tube S2 and the switching tube S6 are complementarily turned on, the switching tube S1 and the switching tube S4 are turned on and turned off at the same time, and the switching tube S2 and the switching tube S3 are turned on and turned off at the same time. The duty cycle is equal to the duty cycle of the switch S2, neither is greater than 0.5 and the phase difference between the two is 180°. The duty cycle of the switch S5 is equal to the duty cycle of the switch S6, neither is less than 0.5 and both phases The difference is 180°. The output voltage V 0 is controlled by adjusting the duty cycle of the switching tube S1 (the duty cycle of the switching tube S1 changes, and the conduction time of the bidirectional switch changes synchronously). The larger the duty cycle of the switching tube S1 is , The greater the output voltage gain.
  4. 根据权利要求1所述的变拓扑LLC谐振变换器的宽增益控制方法，其特征在于：The wide gain control method of a variable topology LLC resonant converter according to claim 1, characterized in that:
    当输入电压位于高压段，逆变电路工作在HBLLC结构的定频PWM控制模态(HBLLC变占空比模态)，开关管S1～S6的开关频率相等且固定，开关管S1和开关管S5互补导通，开关管S3和开关管S6互补导通，开关管S4持续导通，开关管S2持续关断，开关管S1的占空比与开关管S3的占空比相等，均不大于0.5且两者相位差180°，开关管S5的占空比与开关管S6的占空比相等，均不小于0.5且两者相位差180°，通过调节开关管S1的占空比大小实现输出电压V 0的控制，开关管S1的占空比越大，输出电压增益越大。 When the input voltage is in the high voltage section, the inverter circuit works in the fixed frequency PWM control mode of the HBLLC structure (HBLLC variable duty cycle mode), the switching frequencies of the switching tubes S1～S6 are equal and fixed, and the switching tube S1 and the switching tube S5 Complementary conduction, the switching tube S3 and the switching tube S6 are complementarily turned on, the switching tube S4 is continuously turned on, and the switching tube S2 is continuously turned off. The duty cycle of the switching tube S1 is equal to that of the switching tube S3, and both are not greater than 0.5 And the phase difference between the two is 180°, the duty cycle of the switching tube S5 is equal to the duty cycle of the switching tube S6, both are not less than 0.5 and the phase difference between the two is 180°, the output voltage is realized by adjusting the duty cycle of the switching tube S1 For the control of V 0 , the greater the duty cycle of the switch S1, the greater the output voltage gain.
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