KR20180004675A - Bidirectional Converter with Auxiliary LC Resonant Circuit and Operating Method thereof - Google Patents

Abstract

The present invention relates to a bidirectional converter having an auxiliary LC resonant circuit for realizing a constant current (CC) mode charge and a constant voltage (CV) mode charge for a high-efficiency battery charging application in a narrow switching frequency range, and to a driving method thereof. The bidirectional converter having an auxiliary LC resonant circuit of the present invention comprises: a transformation unit including primary and secondary windings to perform voltage conversion; an auxiliary LC resonator unit connected to the transformation unit through a tertiary winding; a first conversion unit connected to the primary winding, and transferring input power transformed by the transformation unit to an output capacitor in accordance with a switching operation of a first full-bridge circuit including a first switch (S_1) to a fourth switch (S_4); and a second conversion unit connected to a battery for supplying the input power, and transferring the input power to the secondary winding in accordance with the switching operation of a second full-bridge circuit including a fifth switch (Q_1) to an eighth switch (Q_4).

Description

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a bidirectional converter having an auxiliary LC resonance circuit,

The present invention relates to a bidirectional converter having a secondary LC resonant circuit and a driving method thereof, and more particularly to a constant current (CC) mode charging and a constant voltage (CV) mode charging for a high efficiency battery charging application in a narrow switching frequency range To a bidirectional converter having one auxiliary LC resonant circuit and a driving method thereof.

Demand for electric vehicles (EVs) is rapidly increasing globally due to significant advantages such as environmentally friendly characteristics, higher efficiency and quietness. One of the important functions of the EVs may be V2G operation, which transfers power from the battery in the EV to the power network upon demand demand. To use the V2G feature, electric vehicles require a bi-directional charger consisting of a bidirectional AC-DC converter with a bidirectional DC-DC converter. DC-DC converters can handle bi-directional power flow from the power grid to the battery during the charging mode and from the battery to the power grid in the discharge mode. Many bidirectional DC-DC converter topologies have been proposed. Their main goals are to achieve soft switching of power switches for a wide range of load variations and to operate at high frequencies so that the size and cost of the magnetic and filter components can be reduced.

Korean Patent Laid-Open No. 10-2013-0013092 (Feb. 23, 2013) has the same circuit structure symmetrical in both directions regardless of the flow of electric power at high pressure or low pressure, and since a CLLC resonance structure is obtained in any direction, There is disclosed a symmetrical bidirectional resonance type converter in which a gain can be freely obtained by a frequency. In a symmetrical dichroic resonance type converter for converting an input DC power supply into a step-up or step-down DC conversion, the first bridge circuit and the second bridge circuit Wherein the first resonant circuit comprises a first inductor and a first capacitor is connected between both ends of the output of the first bridge circuit and a first resonant circuit connected to the first resonant circuit, An inductor, a primary side coil of the transformer, and the first capacitor are connected in series, and between the input terminals of the second bridge circuit, And a second capacitor connected in series with the second inductor, the secondary coil of the transformer, and the second capacitor connected in series, wherein the voltage direction of the first resonant circuit is constantly A second resonant circuit for generating a resonant voltage rectified through the second bridge circuit or a second resonant circuit for varying a voltage direction at a constant cycle through the second bridge circuit, And generates a regenerated voltage rectified through the one bridge circuit. According to the disclosed technology, it is possible to achieve high efficiency of the circuit through a certain soft switching from the light load to the full load, and the same circuit structure regardless of the power transmission direction is easy to design, and the controller structure is the same regardless of the power direction. Design and implementation can be simplified because the values of the resonant elements are the same when the input and output of the isolation transformer are equalized.

Korean Patent No. 10-1000561 (Dec. 2010) discloses that by adding a separate capacitor having a smaller value of the capacitor than the resonance capacitor of the LC resonance circuit portion on the secondary side of the transformer, The primary side resonant current can be rapidly increased while flowing through the capacitor, and a current waveform having a roughly trapezoidal shape can be obtained. As a result, when the current is operated at the same frequency as that of the conventional sine wave type current, A switching unit including a plurality of switches for alternately switching a direct-current voltage to an alternating-current voltage; An LC resonance circuit unit for converting a frequency characteristic of an AC voltage transmitted from the switching unit using a resonance inductor connected in series to the switching unit and resonance of the resonance capacitor; A transformer including a primary side winding connected to the LC resonance circuit and a secondary side winding provided at the primary side winding and the predetermined winding ratio; A secondary side capacitor connected in parallel to the transformer on the secondary side of the transformer; A bridge rectifying circuit part including a plurality of rectifying diodes for rectifying an AC voltage induced on a secondary side of the transformer to a DC voltage; And a gate driving circuit unit for sensing a conduction state of the anti-parallel diode connected to each of the switches and outputting a turn-on gate signal for turning on the switch at the time of conduction of the anti-parallel diode, A first resistor connected to an input terminal to which a signal is input; A capacitor connected to the input terminal in parallel with the first resistor and charged by a turn-on pulse voltage applied through an input terminal; A semiconductor switch having a source, a gate, and a drain connected to the input end and the output end of the capacitor, and the gate end of the output end connected to the switch of the switching unit, respectively; A third resistor connected between the drain of the semiconductor switch and the gate connection of the output terminal; And a fourth resistor connected to the input terminal through a first resistor and a capacitor to form a current conduction path.

Conventional phase shift full-bridge (PSFB) converters as described above are the most widely used topologies in high power applications. This converter can improve power conversion efficiency by the zero-voltage transition feature. However, this converter suffers from some problems with a wide load range such as EMI noise, high stress voltage for rectifier diodes. Also, to achieve ZVS (zero voltage switching) for the full load, a large leakage inductance must be specified. This results in increased duty cycle losses, resulting in reduced converter efficiency. To improve the performance of the converter, many full-bridge zero-current switching DC-DC converters have been developed by using lossless, snubbers or active clamp circuits to suppress the voltage stress of the switches. However, additional snubber circuits increase in size, cost, and complexity, making the topology unsuitable for high-power voltage applications.

Resonant converters with excellent performance: full primary power switches, very high frequency operation, low EMI, simple control, soft switching for high efficiency and low part count, are attractive candidates for interactive applications. Most resonant converters for wide voltage range applications are LLC resonant full-bridge converters. This converter can achieve a full zero voltage switching range for all primary switches regardless of load variations. A small current turn-off can be achieved with a large magnetizing inductance design. However, LLC resonant converters have their own disadvantages. In discharge mode, the magnetizing inductance is in parallel with the bridge voltage, converting the topology into a series resonant converter (SRC). As the operating frequency moves away from the series resonant frequency to operate in the capacitive range, the efficiency of the SRC is significantly reduced.

The bidirectional full-bridge CLLC resonant converter developed at present is introduced for the V2G system by using the LC circuit on the secondary side to achieve zero voltage switching in even discharge mode. However, this topology faces many challenges when applied to battery charger applications that require a wide range of output voltage applications. The magnetizing inductance needs to be small enough to produce the desired output voltage range in constant current (CC) mode charging. This causes a high turn-off current, resulting in a high switching loss of the primary side switches. The switching frequency also varies greatly with wide load variations. This causes high circulating currents in the primary switches, reduces power conversion efficiency, and makes the system difficult to optimize.

To reduce the turn-off current, a method of using an additional LC circuit in the LLC resonant converter has been proposed. An LC circuit operating as a variable inductor across the frequency is connected in parallel with the magnetizing inductance. Thus, a higher effective inductance is generated. As a result, the primary-side switches can achieve almost ZRO (Zero Current Switching) conditions with small turn-off currents, while the output voltage gain remains wide for charging during wide load fluctuations. However, this topology is also a unidirectional topology and is therefore not suitable for application in V2G application systems.

Korean Patent Publication No. 10-2013-0013092 Korean Patent No. 10-1000561

One aspect of the present invention is to add a secondary LC resonant circuit to the tertiary winding to increase effective magnetizing inductance to provide constant current (CC) mode charge and constant voltage (CV) mode charge for high efficiency battery charging applications in a narrow switching frequency range A bidirectional converter having a built-in auxiliary LC resonant circuit and a driving method thereof are provided.

The technical problem of the present invention is not limited to the technical problems mentioned above, and other technical problems which are not mentioned can be understood by those skilled in the art from the following description.

) 내지 제4스위치(

)를 포함하는 제1풀-브리지회로의 스위칭 동작에 따라 상기 변압부에서 변압된 입력전원을 출력 커패시터로 전달하는 제1변환부; 및 입력 전원을 공급하는 배터리와 연결되며, 제5스위치(

) 내지 제8스위치(

)를 포함하는 제2풀-브리지회로의 스위칭 동작에 따라 입력 전원을 상기 2차측권선으로 전달하는 제2변환부를 포함하며, 상기 제1변환부 또는 상기 제2변환부는, 배터리를 충전시키는 배터리충전모드 또는 배터리를 방전시키는 배터리방전모드로 동작한다.A bidirectional converter having an auxiliary LC resonance circuit according to an embodiment of the present invention includes: a transformer including a primary winding and a secondary winding to perform voltage conversion; A secondary LC resonator connected to the transformer through a tertiary winding; A first switch connected to the primary winding,

) To the fourth switch (

A first conversion unit for transferring the input power transformed by the transforming unit to the output capacitor according to a switching operation of the first full-bridge circuit including the first full-bridge circuit; And a battery supplying input power, and the fifth switch (

) To eighth switch

And a second converter for transferring the input power to the secondary winding according to a switching operation of the second full-bridge circuit, wherein the first converter or the second converter includes a battery charging Mode or a battery discharge mode for discharging the battery.

에서 상기 제2스위치(

) 및 상기 제3스위치(

)가 제로 전류 스위칭(Zero Current Switching) 조건 하에서 턴-오프(Turned-Off) 되는 제1배터리충전모드(

)를 포함할 수 있다.In one embodiment, the battery charging mode comprises:

The second switch

And the third switch

) Is turned off under a zero current switching condition in a first battery charging mode (

).

에서 상기 제2스위치(

) 및 상기 제3스위치(

)가 제로 전류 스위칭 조건 하에서 턴-온(Turned-On) 되는 제2배터리충전모드(

)를 더 포함할 수 있다.In one embodiment, the battery charging mode comprises:

The second switch

And the third switch

) Is turned on under zero current switching conditions.

).

에서 상기 제5스위치(

) 및 상기 제8스위치(

)가 제로 전류 스위칭 조건 하에서 턴-오프 되고, 상기 제6스위치(

) 및 상기 제7스위치(

)가 제로 전류 스위칭 조건 하에서 턴-온 되는 제3배터리충전모드(

)를 더 포함할 수 있다.In one embodiment, the battery charging mode comprises:

The fifth switch

And the eighth switch

) Is turned off under zero current switching conditions, and the sixth switch

And the seventh switch

) Is turned on under zero current switching conditions.

).

에서 상기 제1스위치(

) 및 상기 제4스위치(

)가 제로 전류 스위칭 조건 하에서 턴-오프 되며, 상기 제2스위치(

) 및 상기 제3스위치(

)가 제로 전류 스위칭 조건 하에서 턴-온 되는 제4배터리충전모드(

)를 더 포함할 수 있다.In one embodiment, the battery charging mode comprises:

The first switch

) And the fourth switch (

) Is turned off under zero current switching conditions, and the second switch

And the third switch

) Is turned on under the zero-current switching condition.

).

에서 상기 제2스위치(

) 및 상기 제3스위치(

)가 제4배터리충전모드에서 생성된 제로 전류 스위칭으로 턴-온 되는 제5배터리충전모드(

)를 더 포함할 수 있다.In one embodiment, the battery charging mode comprises:

The second switch

And the third switch

Is turned on with the zero current switching generated in the fourth battery charging mode (step < RTI ID = 0.0 >

).

에서 상기 제6스위치(

) 및 상기 제7스위치(

)가 제로 전류 스위칭 조건으로 턴-오프 되고, 상기 제5스위치(

) 및 상기 제8스위치(

)가 제로 전압 스위칭 조건으로 턴-온 되는 제6배터리충전모드(

)를 더 포함할 수 있다.In one embodiment, the battery charging mode comprises:

The sixth switch

And the seventh switch

) Is turned off under the zero current switching condition, and the fifth switch

And the eighth switch

) Is turned on with a zero voltage switching condition.

).

에서 상기 제6스위치(

) 및 상기 제7스위치(

)가 턴-오프 되는 제1배터리방전모드(

)를 포함할 수 있다.In one embodiment, the battery discharge mode comprises:

The sixth switch

And the seventh switch

) Is turned off in the first battery discharge mode (

).

에서 상기 제5스위치(

) 및 상기 제8스위치(

)가 턴-온 되고, 상기 제1스위치(

) 및 상기 제4스위치(

)가 제로 전류 스위칭 조건으로 턴-오프 되는 제2배터리방전모드(

)를 더 포함할 수 있다.In one embodiment, the battery discharge mode comprises:

The fifth switch

And the eighth switch

) Is turned on, and the first switch

) And the fourth switch (

) Is turned off under the zero-current switching condition in the second battery discharge mode

).

에서 공진이 정지되어 상기 제2변환부로의 전력 전달이 중단되는 제3배터리방전모드(

)를 더 포함할 수 있다.In one embodiment, the battery discharge mode comprises:

And a third battery discharge mode in which the resonance is stopped in the second conversion unit and the power transmission to the second conversion unit is stopped

).

에서 상기 제5스위치(

) 및 상기 제8스위치(

)가 턴-오프 되며, 상기 제6스위치(

) 및 상기 제7스위치(

)가 제로 전류 스위칭 조건 하에서 턴-온 되는 제4배터리방전모드(

)를 더 포함할 수 있다.In one embodiment, the battery discharge mode comprises:

The fifth switch

And the eighth switch

Is turned off, and the sixth switch

And the seventh switch

) Is turned on under the zero-current switching condition in the fourth battery discharge mode (

).

에서 상기 제6스위치(

) 및 상기 제7스위치(

)가 턴-온 되는 제5배터리방전모드(

)를 더 포함할 수 있다.In one embodiment, the battery discharge mode comprises:

The sixth switch

And the seventh switch

) Is turned on in the fifth battery discharge mode

).

에서 상기 제6스위치(

) 및 상기 제7스위치(

)가 제로 전류 스위칭 조건 하에서 턴-오프 되는 제6배터리방전모드(

)를 더 포함할 수 있다.In one embodiment, the battery discharge mode comprises:

The sixth switch

And the seventh switch

) Is turned off under the zero current switching condition

).

A method of driving a bidirectional converter having a secondary LC resonant circuit according to another embodiment of the present invention includes a primary winding and a secondary winding, And a first transformer connected to the primary coil, the transformer transforming the input power to the output capacitor according to a switching operation of the first full-bridge circuit, And a second conversion section for transferring an input power source to the secondary winding in accordance with a switching operation of the second full-bridge circuit, the method comprising the steps of: The unit is driven in a battery charging mode for charging the battery or a battery discharging mode for discharging the battery.

In one embodiment, the battery charging mode may include a first battery charging mode to a sixth battery charging mode in response to a turn-on or a turn-off of switches constituting the first full-bridge circuit or the second full- And can be driven in the charge mode.

In one embodiment, the battery discharge mode may include a first battery discharge mode to a sixth battery discharge mode corresponding to turn-on or turn-off of the switches constituting the first full-bridge circuit or the second full- And can be driven in the discharge mode.

In accordance with one aspect of the present invention described above, it is possible to provide superior power efficiency performance over conventional bidirectional converters, to ensure soft switching in all switches, to minimize zero current switching and circulation losses on the primary side, And the required voltage gain is still met by employing a small magnetizing inductance, allowing it to operate in a narrow frequency range.

)를 나타낸 회로도이다.
도 4는 본 발명의 실시예에 따른 양방향 컨버터의 유효 자화 인덕턴스(

)와 CLLC 컨버터의 자화 인덕턴스(

)의 임피던스를 나타낸 그래프이다.
도 5는 본 발명의 입력 임피던스의 위상을 나타낸 그래프이다.
도 6은 본 발명의 CLLC 공진 컨버터의 입력 임피던스의 위상을 나타낸 그래프이다.
도 7은 본 발명의 전압 이득 및 출력 전류를 나타낸 그래프이다.
도 8은 본 발명의 정전류(CC) 모드 충전의 공진 주파수(

)에서의 동작을 나타낸 회로도이다.
도 9는 본 발명의 정전압(CV) 모드 충전의 공진 주파수(

)에서의 동작을 나타낸 회로도이다.
도 10은 본 발명의 방전 모드를 위한 등가 회로를 나타낸 회로도이다.
도 11은 본 발명의 방전 모드에서의 입력 임피던스의 전압 이득 및 위상을 나타낸 그래프이다.
도 12는 본 발명의 낮은 공진 주파수(

)에서 2차 측 스위치의 전류를 나타낸 그래프이다.
도 13은 본 발명의 보조 LC 회로를 사용하는 양방향 CLLC 공진 컨버터의 동작 모드를 나타낸 회로도이다.
도 14는 본 발명의 정전압(CV) 및 정전류(CC) 모드 충전에서 제안된 공진 컨버터의 주요파형을 나타낸 그래프이다.
도 15는 본 발명의 정전압(CV) 모드 충전에서 보조 LC 회로를 사용하는 양방향 CLLC 공진 컨버터의 동작 모드를 나타낸 회로도이다.
도 16은 본 발명의 방전 모드에서 제안된 공진 컨버터의 주요 파형을 나타낸 그래프이다.
도 17은 본 발명의 보조 LC 회로를 사용하는 양방향 CLLC 공진 컨버터의 동작 모드를 나타낸 회로도이다.
도 18은 본 발명의 충전 도중에 EV 배터리의 정전류(CC)/정전압(CV) 충전 프로파일 및 배터리의 등가 임피던스를 나타낸 그래프이다.
도 19는 본 발명의 컨버터를 위한 전압 및 전류 파형을 나타낸 그래프이다.
도 20은 본 발명의 컨버터의 DC 출력 전압 및 전류 파형을 나타낸 그래프이다.
도 21은 본 발명의 충전 모드에서 동작하는 컨버터에 대한 전력 스테이지의 측정된 효율을 나타낸 그래프이다.
도 22는 본 발명의 PM에서 동작하는 컨버터의 손실 분석한 그래프이다.
도 23은 본 발명의 방전 모드에서 동작하는 컨버터에 대한 전압 및 전류 파형을 나타낸 그래프이다.
도 24는 본 발명의 PM 내에서 컨버터의 DC 출력 전압 및 전류 파형을 나타낸 그래프이다.
도 25는 본 발명의 상이한 배터리 전압에 대한 전력 스테이지의 측정된 효율을 나타낸 그래프이다.
도 26은 본 발명의 방전 모드에서 동작하는 컨버터에 대한 손실 분석한 그래프이다.1 is a circuit diagram showing a bidirectional converter having an auxiliary LC resonant circuit according to an embodiment of the present invention.
2 is an AC equivalent circuit diagram of a bidirectional converter with the auxiliary LC resonant circuit of Fig.
FIG. 3 is a graph showing the effective magnetization impedances (

Fig.
FIG. 4 is a graph illustrating the effective magnetizing inductance of a bidirectional converter according to an embodiment of the present invention

) And the magnetizing inductance of the CLLC converter (

). ≪ / RTI >
5 is a graph showing the phase of the input impedance of the present invention.
6 is a graph showing the phase of input impedance of the CLLC resonance converter of the present invention.
7 is a graph showing voltage gain and output current according to the present invention.
8 is a graph showing the relationship between the resonant frequency of the constant current (CC)

Fig.
9 is a graph showing the relationship between the resonance frequency of the constant voltage (CV)

Fig.
10 is a circuit diagram showing an equivalent circuit for the discharge mode of the present invention.
11 is a graph showing the voltage gain and phase of the input impedance in the discharge mode of the present invention.
12 is a graph showing the relationship between the resonance frequency

) Of the secondary side switch.
13 is a circuit diagram showing an operation mode of a bi-directional CLLC resonant converter using the auxiliary LC circuit of the present invention.
14 is a graph showing the main waveforms of the resonance converter proposed in the constant voltage (CV) and constant current (CC) mode charging of the present invention.
15 is a circuit diagram showing an operation mode of a bidirectional CLLC resonant converter using a secondary LC circuit in constant voltage (CV) mode charging of the present invention.
16 is a graph showing a main waveform of the resonance converter proposed in the discharge mode of the present invention.
17 is a circuit diagram showing an operation mode of a bi-directional CLLC resonant converter using the auxiliary LC circuit of the present invention.
18 is a graph showing the constant current (CC) / constant voltage (CV) charging profile and the equivalent impedance of the battery during charging of the present invention.
19 is a graph showing voltage and current waveforms for the converter of the present invention.
20 is a graph showing the DC output voltage and current waveform of the converter of the present invention.
21 is a graph illustrating the measured efficiency of the power stage for a converter operating in the charge mode of the present invention.
22 is a graph showing a loss analysis of a converter operating in the PM of the present invention.
23 is a graph showing voltage and current waveforms for a converter operating in the discharge mode of the present invention.
24 is a graph showing the DC output voltage and current waveform of the converter in the PM of the present invention.
25 is a graph showing the measured efficiency of the power stage for different battery voltages of the present invention.
26 is a graph illustrating loss analysis for a converter operating in the discharge mode of the present invention.

The following detailed description of the invention refers to the accompanying drawings, which illustrate, by way of illustration, specific embodiments in which the invention may be practiced. These embodiments are described in sufficient detail to enable those skilled in the art to practice the invention. It should be understood that the various embodiments of the present invention are different, but need not be mutually exclusive. For example, certain features, structures, and characteristics described herein may be implemented in other embodiments without departing from the spirit and scope of the invention in connection with an embodiment. It is also to be understood that the position or arrangement of the individual components within each disclosed embodiment may be varied without departing from the spirit and scope of the invention. The following detailed description is, therefore, not to be taken in a limiting sense, and the scope of the present invention is to be limited only by the appended claims, along with the full scope of equivalents to which such claims are entitled, if properly explained. In the drawings, like reference numerals refer to the same or similar functions throughout the several views.

Hereinafter, preferred embodiments of the present invention will be described in more detail with reference to the drawings.

1 is a circuit diagram showing a bidirectional converter having an auxiliary LC resonant circuit according to an embodiment of the present invention.

The bidirectional converter having the auxiliary LC resonance circuit according to an embodiment of the present invention includes a transformer 100, a subsidiary LC resonator 200, a first transformer 300 and a second transformer 400, .

)(111)의 일단이 연결되고, 제1공진커패시터(

)(111)의 타단에 제1외부인덕터(

)(112)의 일단이 연결되며, 2차측권선의 일단에 제2공진커패시터(

)(121)일단이 연결되고, 제2공진커패시터(

)(121)의 타단에 제2외부인덕터(

)(122)의 일단이 연결된다.The transformer 100 includes a primary side winding and a secondary side winding having a winding ratio of n: 1, performs voltage conversion, and has a first resonance capacitor

) 111 is connected, and one end of the first resonant capacitor

) 111 is connected to the first external inductor

) 112 is connected to one end of the secondary side winding, and a second resonance capacitor

) 121 is connected, and a second resonant capacitor (

) 121 is connected to a second external inductor

) 122 are connected to each other.

)(201)의 일단이 연결되고, 보조커패시터(

)(201)의 타단에 보조인덕터(

)(202)의 일단이 연결된다.The auxiliary LC resonance section 200 is connected to the transforming section 100 through a tertiary winding providing a winding ratio m and is connected to an auxiliary capacitor

) 201 is connected, and one end of the auxiliary capacitor (

) 201 is connected to the other end of the auxiliary inductor

) 202 are connected to each other.

The first converter 300 is connected to the primary winding and transfers the transformed input power from the transformer 100 to the output capacitor according to the switching operation of the first full-bridge circuit 310.

)(314)를 포함할 수 있다.In one embodiment, the first full-bridge circuit 310 includes a first switch ) 311 to the fourth switch

) ≪ / RTI >

)(311)는, 일단이 제2스위치(

)의 일단 및 출력 커패시터에 연결되며, 타단이 제1외부인덕터(

)(112)의 타단 및 제3스위치(

)(313)의 인달에 연결된다.The first switch (

) 311 has one end connected to the second switch

) And the output capacitor, and the other end is connected to the first external inductor

) 112 and the other end of the third switch

) ≪ / RTI >

)(312)는, 일단이 제1스위치(

)(311)의 일단 및 출력 커패시터에 연결되고, 타단이 1차측권선의 타단 및 제4스위치(

)(314)에 연결된다.The second switch

) 312 has one end connected to the first switch

) 311 and the output capacitor, and the other end is connected to the other end of the primary winding and the fourth switch

) ≪ / RTI >

)(313)는, 일단이 제1외부인덕터(

)(112)의 타단 및 제1스위치(

)(311)의 타단에 연결되고, 타단이 제4스위치(

)(314)의 타단 및 출력 커패시터에 연결된다.The third switch (

) 313 is connected at one end to the first external inductor

) 112 and the other end of the first switch

) 311, and the other end is connected to the fourth switch

) 314 and the output capacitor.

)(314)는, 일단이 1차측권선의 타단 및 제2스위치(

)(312)의 타단에 연결되며, 타단이 제3스위치(

)(313)의 타단 및 출력 커패시터에 연결된다.The fourth switch (

) 314 has one end connected to the other end of the primary winding and the other end of the second switch

) 312, and the other end is connected to the third switch

) 313 and the output capacitor.

The second converter 400 is connected to a battery for supplying input power and transmits the input power to the secondary winding according to the switching operation of the second full-bridge circuit 410.

)(411) 내지 제8스위치(

)(414)를 포함할 수 있다.In one embodiment, the second full-bridge circuit 410 includes a fifth switch

) 411 to an eighth switch

) ≪ / RTI >

)(411)는, 일단이 제6스위치(

)(412)의 일단 및 입력 커패시어터에 연결되고, 타단이 제2외부인덕터(

)(122)의 타단 및 제7스위치(

)(413)의 일단에 연결된다.The fifth switch (

) 411 has one end connected to the sixth switch

) 412 and the input capacitor, and the other end is connected to the second external inductor

) 122 and the seventh switch (

) 413, respectively.

)(412)는, 일단이 제5스위치(

)(411)의 일단 및 입력 커패시터에 연결되며, 타단이 2차측권선의 타단 및 제7스위치(

)(413)의 일단에 연결된다.The sixth switch (

) 412 has one end connected to the fifth switch

) 411 and the input capacitor, and the other end is connected to the other end of the secondary winding and the seventh switch

) 413, respectively.

)(413)는, 일단이 제2외부인덕터(

)(122)의 타단 및 제5스위치(

)(411)의 타단에 연결되고, 타단이 제8스위치(

)(414)의 타단 및 입력 커패시터에 연결된다.Seventh switch (

) 413 has one end connected to the second external inductor

) 122 and the other end of the fifth switch

) 411, and the other end is connected to the eighth switch

) 414 and the input capacitor.

)(414)는, 일단이 2차측권선의 타단 및 제6스위치(

)(412)의 타단에 연결되며, 타단이 제7스위치(

)(413)의 타단 및 입력 커패시터에 연결된다.Eighth switch (

) 414 has one end connected to the other end of the secondary winding and the sixth switch

) 412, and the other end is connected to the seventh switch

) 413 and the input capacitor.

The bidirectional converter having the coarse LC resonance circuit having the above-described configuration is configured such that the auxiliary LC resonance unit 200 is connected to the transforming unit 100 through an additional winding, . 1, the switches of the first conversion unit 300 form a high-frequency full-bridge inverter that converts the input DC voltage into a quasi-square wave AC for transmission through a transformer, The switches of the rectifier 400 act as rectifiers that convert high frequency AC into AC to DC output voltage. As a result, an AC equivalent circuit diagram of the present invention converter is shown in Fig.

), 제1외부인덕터(

)(112), 제2외부인덕터(

)(122), 제1공진커패시터(

)(111), 제2공진커패시터(

)(121), 보조인덕터(

)(202) 및 보조커패시터(

)(201)는 공진 탱크를 구성를 구성한다. In the present invention, the transformer 100, which is a transformer Tr having a winding ratio of n: 1, provides galvanic insulation between the primary and secondary sides, and the magnetization inductance of the transformer 100

), A first outer inductor

) 112, a second outer inductor (

) 122, a first resonant capacitor (

) 111, a second resonant capacitor (

) 121, an auxiliary inductor (

) 202 and an auxiliary capacitor (

) 201 constitute a resonance tank.

)(111) 및 제2공진커패시터(

)(121)는 자동 자속 평형(flux balancing)을 행하는데 참여하고, 각각 제1외부인덕터(

)(112)와 제2외부인덕터(

)(122)와 공진한다. 보조LC공진부(200)는 스위치의 제로 전류 스위칭을 달성하기 위해 등가 자화 인턱턴스(

)를 더 크게 만들기 위해 다른 권선비 m을 제공하는 3차 권선을 통해 삽입된다. 따라서, 스위치의 턴-오프 전류는 상당히 감소되며, 등가 자화 인덕턴스(

)는 수학식 1과 같이 표현된다.The first resonant capacitor (

) 111 and the second resonant capacitor (

) 121 are involved in performing automatic flux balancing, each of which is connected to a first external inductor

) 112 and a second outer inductor (

(122). The auxiliary LC resonator 200 may be configured to provide an equivalent magnetizing in- putance (< RTI ID = 0.0 >

Lt; RTI ID = 0.0 > m < / RTI > to make it larger. Therefore, the turn-off current of the switch is considerably reduced and the equivalent magnetizing inductance (

) Is expressed by Equation (1).

는 스위칭 주파수이고,

, 및

이다.In Equation (1)

Is the switching frequency,

, And

to be.

)를 설명하면 다음과 같다.The effective magnetization impedance in the case of the embodiment of the present invention (

) Is described as follows.

), 유효 자화 임피던스(

) 및 입력 입피던스(

) 공식은 다음의 수학식 2 내지 수학식 4와 같이 유도된다.The magnetization impedance of the AC equivalent circuit shown in Fig. 2 (

), Effective magnetization impedance (

) And input-in-ground (

) Is derived according to the following equations (2) to (4).

)를 나타낸 도면이다.FIG. 3 is a graph showing the effective magnetization impedances (

Fig.

)와 CLLC 컨버터의 자화 인덕턴스(

)의 임피던스를 나타낸 도면이다.Fig. 4 is a graph showing the effective magnetizing inductance of a bidirectional converter having an auxiliary LC resonant circuit according to an embodiment of the present invention

) And the magnetizing inductance of the CLLC converter (

). ≪ / RTI >

)가 공진 주파수(

)에서

및

와 공진하도록 설계되면, 유효 자화 임피던스(

)는 무한대가 됨을 알 수 있다.4, by using an auxiliary circuit in parallel with the magnetizing inductance, the effective magnetizing impedance is much larger than the magnetizing inductance. From Equation (1), the auxiliary capacitor

) Is the resonant frequency (

)in

And

, The effective magnetization impedance (< RTI ID = 0.0 >

) Is infinite.

)의 이점을 이해하기 위하여, 주파수에 따른 입력 임피던스의 위상 그래프가 사용된다. 큰 유효 자화 인덕턴스에 의해, 본 발명은, 이후 공진 주파수(

)에서 제로 위상 각도 조건(ZPA)을 달성할 수 있는 반면, 공진 CLLC 컨버터는 도 5 및 도 6에 도시된 바와 같이 큰 위상을 갖는 유도 영역에서 동작한다.Effective magnetization impedance (

), A phase graph of the input impedance along the frequency is used. Due to the large effective magnetizing inductance,

), While the resonant CLLC converter operates in an inductive region having a large phase as shown in Figs. 5 and 6.

In the CLLC resonant converter, the main loss component is the hard switching turn-off loss and the circulating current in the primary switches. However, the magnetizing inductance also needs to be small enough to obtain zero voltage switching conditions and a wide range of voltage gains in the constant current (CC) and discharge modes as well. This causes a high turn-off current and a large cycling current in the primary side switch. As a result, there is a trade-off between high efficiency and high voltage gain in conventional CLLC resonant converters.

)을 채택함으로써 고전압 이득을 제공하지만, 큰 유효 자화 인덕턴스(

)의 상당한 진보와 함께 작은 턴-오프 전류 및 순환 전류를 달성할 수 있다.On the contrary, the present invention is also applicable to small magnetizing inductance (

) To provide high voltage gain, but large effective magnetizing inductance (

) And a small turn-off current and a circulating current can be achieved.

The voltage gain characteristic according to the embodiment of the present invention will be described as follows.

Using the first-order harmonic approximation, the voltage gain M of the present invention in the battery charging mode is obtained as shown in Equation (5).

)에서 컨버터의 전압 이득은 부하 변동에 따라 변한다. 그러나 출력 전류는 일정하게 유지되고, 따라서 이러한 주파수에서 정전류(CC) 모드 충전을 동작시키는 것이 예상된다. 이득은

과

의 범위에서 부하 변동에 따라 변한다. 이 영역에서, 컨버터는 방전 모드를 위해 이를 사용한다.The voltage gain and output current versus operating switching frequency curve for the converter are shown in FIG. Since the gain of the converter is 1 at the resonance frequency (? 0) independent of the load, it is desirable to design for constant voltage (CV) mode charging in battery charger applications. Resonance frequency

), The voltage gain of the converter varies with the load variation. However, the output current is kept constant, and therefore it is expected to operate the constant current (CC) mode charge at this frequency. The gain is

and

Lt; RTI ID = 0.0 > of < / RTI > In this area, the converter uses it for the discharge mode.

The implementation of the constant current (CC) charging mode according to an embodiment of the present invention will now be described.

The simplified equivalent circuit in the constant current (CC) mode, as shown in Figure 8, is analyzed based on Kirchhoff's theory. The Kirchhoff equation for the circuit is derived as shown in equations (6) to (11).

)가 부하에 관계없이 거의 일정하며, 전류 크기가 입력 전압과 자화 인덕턴스 값에 의존한다는 것을 알 수 있다.

과

가 공진 주파수(

)에서 0을 달성하므로, 출력 전류(

)는 오로지 유효 자화 임피던스(

)와 입력 전압(

)에 의존한다. From equation (11), the charging current

) Is almost constant regardless of the load, and the current magnitude depends on the input voltage and magnetization inductance value.

and

Is the resonance frequency (

), So that the output current (

) Is solely dependent on the effective magnetization impedance (

) And input voltage (

).

)에서 출력 전류는 등가 부하 저항(

)의 변화에 따라 일정하게 유지된다. 따라서 이 주파수에서 정전류(CC) 모드를 구현하는 것이 적합하다.FIG. 7 shows the average output current value through the frequency in accordance with the change of the load. Resonance frequency

), The output current is the equivalent load resistance

). ≪ / RTI > It is therefore appropriate to implement a constant current (CC) mode at this frequency.

)는 공진 주파수(

)에서 실수이고, 따라서 제로 위상 각도 조건은 수학식 12와 같이 얻어질 수 있다.The input impedance of the system (

) Is the resonance frequency (

), So that a zero phase angle condition can be obtained as shown in equation (12).

An implementation of the constant voltage (CV) charging mode according to an embodiment of the present invention will be described below.

The simplified equivalent circuit in the constant voltage (CV) mode shown in FIG. 9 is analyzed based on Kirchhoff's theory. Thus, the Kirchhoff equation for the circuit is derived as: " (13) "

)은 부하에 관계없이 거의 일정하다. 공진 주파수에서

이면 충족될 수 있다. 그러면 충전 전압(

)는 반사된 출력 전압(

)와 동일하고, 이는 정전압(CV) 모드 충전의 구현을 가능케 한다. 이 경우 제안된 시스템의 입력 임피던스는 이 경우 수학식 15와 같이 계산된다.From the diagrams of (14) and (9), the charging voltage

) Is almost constant regardless of load. At the resonant frequency

Can be satisfied. Then charge voltage (

) Is the reflected output voltage (

), Which enables the implementation of constant voltage (CV) mode charging. In this case, the input impedance of the proposed system is calculated as shown in Equation 15 in this case.

)가 수학식 15에서와 같이 부하(

)의 함수이기 때문에,

에 의해 결정된

의 위상은,

의 값이 충분히 크면 거의 0이 된다. 따라서 본 설계의 가장 중요한 양상은 유효 자화 임피던스(

)를 최대화하는 것이다. 주파수(

)에서

및

를 갖는 커패시터(

)의 공진 주파수를 선택함으로써,

의 최대값을 만들고 ZPA 조건을 달성하는 것이 가능하다.Input Impedance (

) Is calculated as the load (

) ≪ / RTI >

Determined by

Phase,

Is sufficiently large, it becomes almost zero. Therefore, the most important aspect of this design is the effective magnetization impedance (

) Is maximized. frequency(

)in

And

A capacitor having

),

And to achieve the ZPA condition.

An implementation of the discharge mode according to the embodiment of the present invention will now be described.

및

사이에서 동작한다. 종래의 CLLC 컨버터와 동일하게, 원하는 전압 이득 및 작은 주파수 변동을 충족시키기 위하여 작은 자화 인덕턴스(

)가 요구된다. 도 11에 도시된 바와 같이, 입력 임피던스(

)의 위상은 유도성이어서, 컨버터는 2차 측 스위치를 위한 제로 전압 스위칭 조건을 달성할 수 있다.Figure 11 shows the voltage gain of the proposed converter in the discharge mode, in which the converter has a resonant frequency

And

Lt; / RTI > Similar to a conventional CLLC converter, a small magnetizing inductance (< RTI ID = 0.0 >

) Is required. As shown in Fig. 11, the input impedance (

) Is inductive, so that the converter can achieve zero voltage switching conditions for the secondary side switch.

)의 피크 값과 동일하다. 그러나 자화 인덕턴스는 넓은 전압 이득을 얻기 위하여 충분히 작아야 하는 것이 요구된다. 결과적으로, 자화 전류(

)의 피크 값은 상당히 커지고, 이에 의해 도 12의 (b)에 도시된 종래의 컨버터에서 큰 순환 전류를 야기한다.12 shows waveforms of a secondary side current of the present invention and a conventional CLLC resonance converter, in a CLLC converter. The circulating current is the magnetizing inductance current (

). ≪ / RTI > However, the magnetizing inductance is required to be small enough to obtain a wide voltage gain. As a result, the magnetizing current (

) Is considerably large, thereby causing a large circulating current in the conventional converter shown in Fig. 12 (b).

)의 피크 값과 동일하다. 그러나 전류(

)는 자화 인덕턴스 전류(

)와 반대 값을 갖는다. 그러므로, 순환 전류는 도 12의 (a)에 도시된 바와 같이, 작고 항상

보다 낮다.For the present invention, the circulating current is the total current

). ≪ / RTI > However,

) Is the magnetizing inductance current (

). ≪ / RTI > Therefore, as shown in Fig. 12 (a), the circulating current is small and always

.

), 제1외부인덕터(

)(112), 제2외부인덕터(

)(122), 제1공진커패시터(

)(111), 제2공진커패시터(

)(121), 보조인덕터(

)(202) 및 보조커패시터(

)(201)는 공진 탱크를 구성할 수 있다.In the bi-directional converter having the auxiliary LC resonant circuit having the above-described configuration, the magnetizing inductance (

), A first outer inductor

) 112, a second outer inductor (

) 122, a first resonant capacitor (

) 111, a second resonant capacitor (

) 121, an auxiliary inductor (

) 202 and an auxiliary capacitor (

) 201 can constitute a resonance tank.

In the bidirectional converter having the auxiliary LC resonant circuit according to another embodiment of the present invention, the first conversion unit 300 or the second conversion unit 400 may be a battery charging mode charging the battery or a battery discharging mode discharging the battery Can be driven.

In one embodiment, the battery charging mode may be configured to switch from the first battery charge mode to the second full-bridge mode in response to the turn-on or turn-off of the switches constituting the first full-bridge circuit 310 or the second full- And may be driven in the sixth battery charging mode.

In one embodiment, the battery discharge mode is a mode in which either the first full-bridge circuit 310 or the second full-bridge circuit 410 is turned on or off in response to a turn-on or a turn- And can be driven in the sixth battery discharge mode.

)(311) 내지 제4스위치(

)(314)가 제어되고, 제5스위치(

)(411) 내지 제8스위치(

)(414)는 정류 다이오드로서 작용한다. 방전 모드에서, 제1스위치(

)(311) 내지 제4스위치(

)(314)의 보디 다이오드가 브리지 정류기 회로서 작용하는 동안, 제5스위치(

)(411) 내지 제8스위치(

)(414)은 조절된다.In the bidirectional converter having the above-described configuration and having the coarse LC resonant circuit, in the battery charging mode, the first switch

) 311 to the fourth switch

) 314 are controlled, and the fifth switch

) 411 to an eighth switch

) 414 serves as a rectifying diode. In the discharge mode, the first switch

) 311 to the fourth switch

) 314 serves as a bridge rectifier circuit, the fifth switch

) 411 to an eighth switch

) 414 are adjusted.

)이 없는 이상적인 구성요소로 간주되며, 2) 모든 커패시터와 인덕터의 기생 저항은 무시한다는 가정 하에 설명하기로 한다.In the present invention, in order to avoid the complexity of the analysis, 1) all semiconductor switches are connected to body diodes and output capacitors (

), And 2) the parasitic resistance of all capacitors and inductors is ignored.

)에서 동작한다.The constant current operation of the method of driving the auxiliary LC resonant circuit according to another embodiment of the present invention has six modes during the switching period. The main waveforms and current paths for the constant current (CC) mode are shown in FIG. In this mode, as shown in Equation (16), the resonance frequency

).

)는, 제2스위치(

)(312) 및 제3스위치(

)(313)가

에서의 제로 전류 스위칭(Zero Current Switching) 조건 하에서 턴-오프(Turned-Off) 되며, 제2변환부(400)로 전력이 전송되지 아니하며, 1차측전류(

)가 제1스위치(

)(311) 및 제4스위치(

)(314)의 출력커패시터를 방전시킨 후 제로 전압 스위칭 조건 하에서 제1스위치(

)(311) 및 제4스위치(

)(314)의 보디 다이오드(body diode)를 통과할 수 있다.The first battery charging mode (FIG. 13

Is connected to the second switch

) 312 and the third switch (

) 313

Off state under the Zero Current Switching condition at the second converter 400 and no power is transmitted to the second converter 400 and the primary current

Is connected to the first switch

) 311 and a fourth switch

After discharging the output capacitor of the first switch 314, under the zero voltage switching condition,

) 311 and a fourth switch

(Not shown) through a body diode.

)는, 제2스위치(

)(312) 및 제3스위치(

)(313)가 자화 전류(

)및 보조 전류(

)의 합인 작은 전류로 턴-오프 된다. 이들 2가지 전류는 제1스위치(

)(311) 및 제4스위치(

)(314)의 출력 커패시터를 방전시킨다. 방전 과정 이후, 1차 전류(

)는 보디 다이오드를 통해 전도되어, 스위치가 제로 전압 스위칭 조건 하에서 동작하게 한다. 이 모드에서는 2차 측 정류 회로 어떠한 전력도 전달되지 않는다.The first battery charging mode shown in Fig. 15 (a)

Is connected to the second switch

) 312 and the third switch (

) 313 is the magnetizing current (

) And auxiliary current (

Lt; RTI ID = 0.0 > a < / RTI > These two currents flow through the first switch

) 311 and a fourth switch

) ≪ / RTI > After the discharge process, the primary current (

) Is conducted through the body diode, causing the switch to operate under zero voltage switching conditions. In this mode, no power is delivered to the secondary rectifier circuit.

)는,

에서 제2스위치(

)(312) 및 제3스위치(

)(313)가 제로 전류 스위칭으로 턴-온(Turned-On) 되며, 1차측전류(

) 및 전압 소스(

)의 방향이 마이너스에서 플러스로 변경되며, 전력이 변압부(100), 제5스위치(

)(411) 및 제8스위치(

)(414)를 통해 배터리로 전송될 수 있다.The second battery charging mode (FIG. 13B)

),

To the second switch (

) 312 and the third switch (

) 313 is turned on by zero current switching, and the primary side current (

) And a voltage source (

) Is changed from minus to plus, and power is supplied to the transformer 100, the fifth switch

) 411 and an eighth switch

Lt; RTI ID = 0.0 > 414 < / RTI >

)는 수학식 17과 같이 얻을 수 있다.Assuming a negligible dead time period, the primary current (< RTI ID = 0.0 >

Can be obtained as shown in Equation (17).

이며, 각 성분들은 수학식 18과 같이 계산될 수 있다.In Equation 17,

, And each component can be calculated as shown in Equation (18).

은 변압기 권선 양단의 전압이고,

은

의 양단의 전압이며,

는 스위칭 주기이다. 또한 자화 전류(

)및 보조 전류(

)는 수학식 19 내지 수학식 21과 같이 유도될 수 있다.In the above equation,

Is the voltage across the transformer winding,

silver

Lt; RTI ID = 0.0 >

Is the switching period. Also,

) And auxiliary current (

) Can be derived as shown in Equations (19) to (21).

In Equation (20)

)는, 1차 전류(

)가 자화 전류(

)및 보조 전류(

,(

))의 총 전류에 도달할 때 종료될 것이고, 이는 1차 측으로부터 2차 측으로 전력을 전달하는 공진 동작의 종료를 의미한다. 동작하는 컨버터가

에서 동작할 때, Δ는 0과 같다.The above-described second battery charging mode (

), The primary current (

) Is the magnetizing current

) And auxiliary current (

, (

)), Which means the end of the resonant operation which transfers power from the primary side to the secondary side. An active converter

, Δ is equal to zero.

)는,

에서, 제1스위치(

)(311) 및 제4스위치(

)(314)는, 제로 전압 스위칭 조건으로 턴-온 된다. 전력은 변압부(100) 및 2차 정류기 제5스위치(

)(411) 및 제8스위치(

)(414)를 통해 배터리로 전달된다. 1차전류(

)와 전압원(

)은 방향을 음에서 양으로 변화시킨다.The second battery charging mode (FIG. 15 (b)

),

, The first switch (

) 311 and a fourth switch

) 314 are turned on with a zero voltage switching condition. Power is supplied to the transformer 100 and the second rectifier fifth switch

) 411 and an eighth switch

) ≪ / RTI > Primary current (

) And a voltage source

) Changes the direction from negative to positive.

)는, 제2공진커패시터(

)(121)와 제2외부인덕터(

)(122) 와 자화 인덕턴스(

)의 총합 사이의 공진은

에서 종료되며, 제5스위치(

)(411) 및 제8스위치(

)(414)가 제로 전류 스위칭 조건으로 턴-오프 되는 반면, 제6스위치(

)(412) 및 제7스위치(

)(413)가 제로 전압 스위칭 조건으로 턴-온 되며, 1차 전류는 수학식 22와 같이 계산될 수 있다.The third battery charging mode (FIG. 13 (c)

Is connected to the second resonant capacitor (

) 121 and a second outer inductor (

) 122 and magnetizing inductance (

) Is the resonance between the sum of

, And the fifth switch (

) 411 and an eighth switch

) 414 is turned off under the zero current switching condition, while the sixth switch

) 412 and a seventh switch

) 413 is turned on under the zero voltage switching condition, and the primary current can be calculated as shown in Equation (22).

이다.In Equation 22,

to be.

)는,

에서, 제1스위치(

)(311) 및 제4스위치(

)(314)는 거의 제로 전류 스위칭 조건으로 턴-오프 된다. 이 모드 동안의 동작은 제1배터리충전모드와 반대로 스위치 쌍이 변경되는 것과 유사하다. 1차 전류(

)는 제2스위치(

)(312) 및 제3스위치(

)(313)의 출력 커패시터를 방전시키고, 이후 보디 다이오드를 통과한다. 이것은 이들 스위치가 제로 전압 스위칭 조건 하에서 턴-온 되게 한다.The third battery charging mode (FIG. 15 (c)

),

, The first switch (

) 311 and a fourth switch

) 314 are turned off with almost zero current switching conditions. The operation during this mode is analogous to the change of the switch pair as opposed to the first battery charge mode. Primary current (

Is connected to the second switch

) 312 and the third switch (

) 313, and then passes through the body diode. This allows these switches to be turned on under zero voltage switching conditions.

)는,

에서 제1스위치(

)(311) 및 제4스위치(

)(314)가 제로 전류 스위칭 조건 하에서 턴-오프 되며, 1차측전류(

)가 제2스위치(

)(312) 및 제3스위치(

)(313)의 출력 커패시터를 방전시킨 후 제2스위치(

)(312) 및 제3스위치(

)(313)의 보디 다이오드를 통과하여 제2스위치(

)(312) 및 제3스위치(

)(313)를 제로 전압 스위칭 조건 하에서 턴-온시킬 수 있다.The fourth battery charging mode shown in Fig. 13 (d)

),

The first switch (

) 311 and a fourth switch

) 314 is turned off under zero current switching conditions, and the primary side current (

Is connected to the second switch

) 312 and the third switch (

After discharging the output capacitor of the second switch 313,

) 312 and the third switch (

) 313 to be connected to the second switch (

) 312 and the third switch (

) 313 may be turned on under zero voltage switching conditions.

)는,

에서 제2스위치(

)(312) 및 제3스위치(

)(313)는 제로 전압 스위칭 조건으로 턴-온 된다. 컨버터는 변압부(100)와 2차 정류기 제6스위치(

)(412) 및 제7스위치(

)(413)를 통해 1차 측으로부터 배터리로 전력을 전달하기 시작한다. 1차전류(

)와 전압원(

)은 방향을 양에서 음으로 변경한다.The fourth battery charging mode shown in Fig. 15 (d)

),

To the second switch (

) 312 and the third switch (

) 313 are turned on with a zero voltage switching condition. The converter includes a transformer 100 and a second rectifier sixth switch

) 412 and a seventh switch

) ≪ / RTI > 413 from the primary side to the battery. Primary current (

) And a voltage source

) Changes the direction from positive to negative.

)는,

에서 제2스위치(

)(312) 및 제3스위치(

)(313)가 제4배터리충전모드에서 생성된 제로 전압 스위칭으로 턴-온 되며, 1차측전류(

)와 전압 소스(

)의 방향이 플러스에서 마이너스로 변경되고, 전력이 변압부(100), 제6스위치(

)(412) 및 제7스위치(

)(413)를 통해 배터리로 전달될 수 있다. 이때, 1차 전류(

)의 방향은 제2배터리충전모드의 방향과 반대이지만, 수학식 23에서과 같이 상이한 스위치 쌍을 갖는 동일한 동작 공식이다.The fifth battery charging mode shown in Fig. 13 (e)

),

To the second switch (

) 312 and the third switch (

) 313 is turned on with the zero voltage switching generated in the fourth battery charging mode, and the primary side current (

) And the voltage source (

) Is changed from positive to negative, and the electric power is supplied to the transformer 100, the sixth switch

) 412 and a seventh switch

) ≪ / RTI > At this time, the primary current

Is opposite to the direction of the second battery charging mode, but is the same operating formula with a different pair of switches as in Equation 23. < RTI ID = 0.0 >

)는,

에서 제2공진커패시터(

)(121)와 제2외부인덕터(

)(122)와 공진 자화 인덕턴스(

)의 총합 사이의 공진이 시작하며, 제6스위치(

)(412) 및 제7스위치(

)(413)가 제로 전류 스위칭 조건으로 턴-오프 되는 반면, 제5스위치(

) (411)및 제8스위치(

)(414)가 제로 전압 스위칭 조건으로 턴-온 될 수 있으며, 제3배터리충전모드의 방식과 같이, 1차 전류(

)는 수학식 24와 같이 유도된다.The sixth battery charging mode (FIG. 13 (f)

),

The second resonant capacitor (

) 121 and a second outer inductor (

) 122 and the resonance magnetization inductance (

) Starts, and the resonance between the sum of the sixth switch

) 412 and a seventh switch

) 413 is turned off under the zero-current switching condition, while the fifth switch

) 411 and an eighth switch

) 414 may be turned on with a zero voltage switching condition and the primary current (e.g.,

Is derived as shown in equation (24).

이다.In Equation 24,

to be.

)에서 동작한다.There are four operating modes in one switching period in constant voltage operation. The main waveforms and current paths for the constant current (CC) mode are shown in FIGS. 14 and 15. FIG. The converter has a resonant frequency (

).

에서

에 이르며, 제1배터리방전모드 내지 제6배터리방전모드의 6개의 모드로 구동될 수 있다.In the bidirectional converter having the auxiliary LC resonant circuit having the above-described configuration, in the case of the battery discharge mode in which the battery is discharged,

in

And can be driven in six modes of the first battery discharge mode to the sixth battery discharge mode.

)는,

에서 제6스위치(

)(412) 및 제7스위치(

)(413)가 턴-오프 되며, 제2변환부(400)로 전력을 전달하지 아니하며, 2차측전류(

)가 제5스위치(

)(411) 및 제8스위치(

)(414)의 출력 커패시터를 방전시킨 후 제1스위치(

)(311) 및 제4스위치(

)(314)의 보디 다이오드를 통과할 수 있다.The first battery discharge mode shown in Fig. 17 (a)

),

To the sixth switch (

) 412 and a seventh switch

) 413 is turned off, the power is not transmitted to the second conversion unit 400, and the secondary current (

Is connected to the fifth switch

) 411 and an eighth switch

After discharging the output capacitor of the first switch 414,

) 311 and a fourth switch

) ≪ / RTI >

)는,

에서 제5스위치(

)(411) 및 제8스위치(

)(414)가 턴-온 되며, 2차측전류(

)가 제5스위치(

)(411)를 통해 흐르며, 전력이 변압부(100)를 통해 출력단으로 전달하며, 2차측전류(

)의 전류 방향이 제5스위치(

)(411) 및 제8스위치(

)(414)에 대응하여 플러스로 변경되며, 제1외부인덕터(

)(112)가 제1공진커패시터(

)(111)와 공진하며, 제2외부인덕터(

)(122)가 제2공진커패시터(

)(121)와 공진하며, 2차측전류(

)가

및

의 전류 총 합보다 항상 크며, 2차측전류(

)가

와

의 합과 같은 경우 모드가 종료되며, 제1스위치(

)(311) 및 제4스위치(

)(314)가 제로 전류 스위칭 조건으로 턴-오프 될 수 있다.The second battery discharge mode shown in Fig. 17 (b)

),

To the fifth switch (

) 411 and an eighth switch

) 414 is turned on, and the secondary current (

Is connected to the fifth switch

) 411, the electric power is transmitted through the transformer 100 to the output terminal, and the secondary current (

The current direction of the fifth switch

) 411 and an eighth switch

) 414, and the first external inductor (

) 112 is connected to the first resonant capacitor

) 111, and the second external inductor (

) 122 is connected to the second resonant capacitor

) 121, and the secondary side current (

)end

And

And the secondary current (< RTI ID = 0.0 >

)end

Wow

The mode is ended, and when the first switch (

) 311 and a fourth switch

) 314 may be turned off with a zero current switching condition.

)는,

에서 공진이 정지되며, 전력이 더 이상 제2변환부(400)로 전달되지 아니하며, 전류(

)가 0이 되어 출력 커패시터가 1차측전류(

)에 의해 더 이상 충전되지 아니하며, 2차측전류(

)가 모드 동안

및

의 합과 동일할 수 있다.The third battery discharge mode shown in Fig. 17 (c)

),

The resonance is stopped, the electric power is no longer transmitted to the second conversion unit 400, and the current (

) Becomes 0 and the output capacitor is the primary current (

), And the secondary current (< RTI ID = 0.0 >

) Is in mode

And

May be equal to the sum of.

)는,

에서 제5스위치(

)(411) 및 제8스위치(

)(414)가 턴-오프 되며, 해당 모드 역시 휴지 지속 시간(Dead Time Duration)으로서 제1배터리방전모드와 유사하게 동작하나, 스위치 쌍의 방전 커패시턴스가 제6스위치(

)(412) 및 제7스위치(

)(413)로 변경되며, 제6스위치(

)(412) 및 제7스위치(

)(413)의 보디 다이오드를 통해 전도되는 2차측전류(

)가 제로 전압 스위칭 조건 하에서 제6스위치(

)(412) 및 제7스위치(

)(413)를 턴-온시킬 수 있다.The fourth battery discharge mode shown in Fig. 17 (d)

),

To the fifth switch (

) 411 and an eighth switch

) 414 is turned off and the corresponding mode also operates similarly to the first battery discharge mode as the dead time duration, but the discharge capacitance of the switch pair is the sixth switch

) 412 and a seventh switch

) 413, and the sixth switch

) 412 and a seventh switch

) 413 that is conducted through the body diode of the secondary side current

Lt; / RTI > under a zero voltage switching condition.

) 412 and a seventh switch

) ≪ / RTI >

)는,

에서 제6스위치(

)(412) 및 제7스위치(

)(413)가 턴-온 되며, 제2변환부(400)에서 제1변환부(300)로 전력을 전송하기 시작하며, 2차측전류(

)의 전류 방향이 제2배터리방전모드와 반대이지만, 제6스위치(

)(412) 및 제7스위치(

)(413)의 스위치 쌍과 동일한 연산식을 가질 수 있다.The fifth battery discharge mode shown in Fig. 17 (e)

),

To the sixth switch (

) 412 and a seventh switch

) 413 is turned on and starts to transmit power from the second conversion unit 400 to the first conversion unit 300 and the secondary side current

) Is opposite to the second battery discharge mode, but the sixth switch

) 412 and a seventh switch

) ≪ / RTI > 413 of FIG.

)는,

에서 2차측전류(

)가

와

의 총 전류에 도달된 후 공진 및 전력 전달이 중단되며, 변압부(100)를 통해 전력이 변환되지 않기 때문에 1차측전류(

)가 0이 되며, 제6스위치(

)(412) 및 제7스위치(

)(413)가 제로 전류 스위칭 조건으로 턴-오프 될 수 있다.The sixth battery discharge mode shown in Fig. 17 (f)

),

The secondary current (

)end

Wow

Resonance and power transmission are stopped after the total current of the transformer 100 is reached, and the power is not converted through the transformer 100,

Becomes 0, and the sixth switch (

) 412 and a seventh switch

) 413 may be turned off with zero current switching conditions.

)(111) 및 제1외부인덕터()(112)와 2차 측의 제2공진커패시터(

)(121) 및 제2외부인덕터(

)(122) 및 자화 인덕턴스(

)의 공진 탱크는 전압 이득의 범위에 대한 여유를 고려함으로써 설계된다. 반면, 보조 회로의 보조커패시터(

)(201) 및 보조인덕터(

)(202)는, 제로 전압 스위칭 영역 및 작은 제로 전류 스위칭 스위칭 손실을 달성하기 위하여, 유효 자화 인덕턴스(

)를 최대화하도록 설계된다. 또한, 공진 주파수(

)는 정전류(CC) 모드 충전에서 일정한 출력 전류를 설계하기 위하여 선택된다.As with the above analysis, the principles of operation of the proposed converter and conventional CLLC converter in the battery discharge mode are the same over the entire input voltage range generated from the EV battery. Therefore, the design procedure of the converter proposed in the present invention is similar to the conventional CLLC resonance converter. And the first resonance capacitor of the primary side

) 111 and the first outer inductor ( ) 112 and the second resonance capacitor (

) 121 and a second outer inductor (

) 122 and magnetizing inductance (

) Are designed by taking into account the margin for the range of voltage gain. On the other hand, the auxiliary capacitor (

) 201 and an auxiliary inductor (

) 202 may be configured to achieve a zero voltage switching region and a small zero current switching switching loss,

). Further, the resonance frequency (

) Is selected to design a constant output current in constant current (CC) mode charging.

First, the effective magnetization inductance will be described as follows.

)(202) 및 보조커패시터(

)(201)는 등가 자화 인덕턴스(

)를 최대화하기 위하여 수학식 26과 같이 설계된다.The large effective magnetizing inductance will result in a small turn-off current, which reduces the switching loss of the primary side switch. Therefore, the auxiliary inductor

) 202 and an auxiliary capacitor (

) 201 is the equivalent magnetizing inductance (

) ≪ / RTI >

)는 수학식 27을 충족시키도록 설계된다.However, the primary side current must be large enough to discharge the output capacitor of the primary side switch during the dormant period, and the magnetization of this current depends on the duration of the equivalent magnetizing inductance and downtime. Therefore, the zero-voltage switching on the primary side is dependent on the equivalent magnetizing inductance, the switch output capacitor, the operating switching frequency and the dwell time. Therefore, equivalent magnetizing inductance (

) Is designed to satisfy equation (27).

는 1차 스위치의 휴지 시간이고,

는 스위치의 출력 커패시터이며,

는 최대 동작 주파수이다.In the above equation,

Is the dwell time of the primary switch,

Is the output capacitor of the switch,

Is the maximum operating frequency.

)(201)의 전압 응력을 설명하면 다음과 같다.The auxiliary capacitor (

) 201 will be described as follows.

)(201)의 전압 응력은 본 발명의 또 다른 제약 조건이다. 커패시터의 전압 응력은 수학식 28과 같이 계산된다.Auxiliary capacitor (

) 201 is another constraint of the present invention. The voltage stress of the capacitor is calculated as shown in equation (28).

)(201)는, 제1외부인덕터(

)(112) 및 제1공진커패시터(

)(111)와 동일한 주파수에서 보조인덕터(

)(202)및

과 공진한다. 그러나 제1외부인덕터(

)(112)의 값은 보조인덕터(

)(202)및

보다 상당히 작다. 보조커패시터(

)(201)가 매우 작은 값으로 설계되면, 수학식 28과 같이 큰 전압 응력이 보조커패시터(

)(201)에서 생성된다. 그러므로 보조인덕터(

)(202)는 보조커패시터(

)(201)에서의 전압 응력을 줄이기 위하여 더 큰 보조커패시터(

)(201)를 만들기에 충분히 작게 설계되어야 한다. 본 발명에서, 보조인덕터(

)(202)는 자화 인덕터(

)의 절반이 되도록 선택된다. 이는 본 발명의 성능을 저하시킬 수 있는 보조인덕터(

)(202)의 상당한 코어 손실을 회피할 뿐만 아니라 보조커패시터(

)(201)의 전압 응력을 작은 값으로 유지할 수 있다.Auxiliary capacitor (

) 201 includes a first outer inductor

) 112 and the first resonant capacitor (

) 111 at the same frequency as the auxiliary inductor

) 202 and

. However, the first outer inductor

) ≪ / RTI > (112)

) 202 and

Lt; / RTI > Auxiliary capacitor (

) 201 is designed to be a very small value, a large voltage stress is applied to the auxiliary capacitor (

) ≪ / RTI > Therefore, the auxiliary inductor

) 202 is connected to the auxiliary capacitor (

) 201 in order to reduce the voltage stress in the auxiliary capacitor < RTI ID = 0.0 >

Lt; RTI ID = 0.0 > 201 < / RTI > In the present invention, the auxiliary inductor

) 202 is a magnetization inductor

). ≪ / RTI > This is because the auxiliary inductor (which can deteriorate the performance of the present invention

0.0 > 202) < / RTI > as well as the auxiliary capacitor < RTI ID =

) 201 can be kept at a small value.

Next, the soft switching condition in the constant current (CC) mode will be described as follows.

)에서 동작한다. ZV제로 전류 스위칭를 달성하기 위하여, 시스템의 입력 임피던스는 0이 되도록 요구된다. 이는 1차전압(

)이 1차 전류(

)와 동위상인 것을 의미한다. 그러므로 정전류(CC) 모드에서의 공진 주파수는,

에서 입력 임피던스(

)의 위상을 0으로 만들기 위하여, 수학식 29와 같이 설계된다.In constant current (CC) mode charging, the converter converts the resonant frequency

). In order to achieve ZV zero current switching, the input impedance of the system is required to be zero. This is because the primary voltage (

) Is the primary current

Quot;) < / RTI > Therefore, the resonant frequency in the constant current (CC)

Input Impedance at (

Is designed as shown in Equation (29) in order to make the phase of the input signal " 0 "

=380~400V,

=250~420V로 구현되었다. 따라서 컨버터의 요구되는 전압 이득은 정전압(CV) 모드 충전에서 약 1.05이다. RM에서, 요구 전압 이득은 약 0.95~1.6이다. 직렬 공진 주파수는 따라서 60kHz로 설계된다. 토폴로지의 규격은 표 1에 상세하게 도시된다.A 3.3 kW bidirectional resonant prototype was implemented in the laboratory to validate the proposed converter. The experimental converter is designed to meet the following specifications

= 380 to 400 V,

= 250 ~ 420V, respectively. Therefore, the required voltage gain of the converter is about 1.05 in constant voltage (CV) mode charging. At RM, the required voltage gain is about 0.95 to 1.6. The series resonant frequency is thus designed to be 60 kHz. The specifications of the topology are shown in detail in Table 1.

=570m²)을 갖는 변압기 PQ 72/54가 변압기(

)를 설계하기 위하여 사용되는 반면, 변압기 PQ 5050은 두 개의 직렬 인덕터를 위해 채택된다. 코일의 공진 보상을 위해, B32653 및 B32654의 필름 커패시터가 고려된다. 제안된 컨버터의 설계 파라미터는 표 2에 상세하게 나열되었다.The DC-DC converter includes three parts of a second full-bridge circuit coupled to the inverter circuit, the resonant network, and the output filter capacitor. In the full-bridge circuit, the switching device used is FCH76N60. As shown in Table 2, the total effective area (

= 570 m ² ) transformer PQ 72/54 is connected to the transformer

), While the transformer PQ 5050 is employed for two series inductors. For resonance compensation of the coil, film capacitors of B32653 and B32654 are considered. The design parameters of the proposed converter are listed in detail in Table 2.

First, the operation of the converter in the power supply mode will be described as follows.

)(311) 및 제4스위치(

)(314)는 고주파 인버터로서 작용하는 반면, 제5스위치(

)(411) 내지 제8스위치(

)(414)의 보디 다이오드는 정류기 다이오드로서 작용한다. 전력 공급 모드에서 배터리의 충전 프로파일이 도 18에 도시되었다. 정전류(CC) 모드 충전 도중에, 출력 전류는 낮은 배터리 전압에 대해 7.85A를 유지한다. 다음 스테이지인 정전압(CV) 모드에서, 출력 전압이 420V의 최대 전압으로 유지되는 동안 충전 전류는 감소된다.In the power supply mode, the first switch

) 311 and a fourth switch

) 314 serves as a high-frequency inverter, while a fifth switch

) 411 to an eighth switch

) 414 serves as a rectifier diode. The charge profile of the battery in the power supply mode is shown in Fig. During constant current (CC) mode charging, the output current maintains 7.85A for low battery voltage. In the next stage, constant voltage (CV) mode, the charge current is reduced while the output voltage is held at the maximum voltage of 420V.

)와 전압 파형(

)이 도 19에 도시되었다.The measured current of the primary MOSFET for a converter operating in battery charging mode for two-mode charging under full load conditions

) And the voltage waveform (

Is shown in Fig.

)(311)및 2차 측 스위치 5스위치(

)(411)의 파형이다. 적색 선은 1차 스위치에 양단의 전압 응력을 나타내고, 녹색 선은 순방향 전류에 대응한다. 컨버터는 공진 주파수(f_{정전류(CC)}=38kHz)에서 동작한다. 모든 1차 스위치가 턴-온 제로 전압 스위칭 및 턴-오프 제로 전류 스위칭를 달성한 다는 것을 알 수 있다. 2차 측의 스위치는 전체 부하 범위에 대해 완전한 ZVZCS를 달성한다. 정전압(CV) 모드 충전에 대한 동작 파형은 도 19의 (b)에 도시되었다. 컨버터는 59kHz의 공진 주파수(

) 하에서 동작하여, 최대 부하 조건의 70%에서 필요한 전압 이득에 도달한다. 1차 MOSFET는 제로 전압 스위칭 조건을 보장하는 음의 값에서 턴-온 된다. 턴-오프 전류는 이 경우 약 4A 정도로 매우 작다.Figure 19 (a) shows a first switch first switch (

) 311 and the secondary side switch 5 switch (

) ≪ / RTI > The red line represents the voltage stress at both ends of the primary switch, and the green line corresponds to the forward current. The converter operates at the resonant frequency (f _{constant current (CC)} = 38 kHz). It can be seen that all primary switches achieve turn-on zero voltage switching and turn-off zero current switching. The secondary switch achieves full ZVZCS for the entire load range. The operation waveform for charging in the constant voltage (CV) mode is shown in Fig. 19 (b). The converter has a resonance frequency of 59 kHz (

) To reach the required voltage gain at 70% of the full load condition. The primary MOSFET is turned on at a negative value to ensure zero voltage switching conditions. The turn-off current is very small, about 4A in this case.

=3.3kW의 PM에서 컨버터의 DC 출력 전압과 전류 파형을 도시한 것이며, 도 21은 YOKOGAWA WT3000에 의해 측정된 상이한 모드 충전에서의 PM 하의 컨버터에 대한 변환 효율을 도시한다. 효율은 컨버터가 공진 주파수에서 작동할 때 입력 전압 400V에 대해 최대이다. 컨버터 효율은 공칭 입력 전압에서 500W의 낮은 부하 (부하의 15 %)에서도 96 %보다 높다. 피크 전력 스테이지 효율은

=2.3kW에서 98.1%이다.20

= 3.3 kW, and Fig. 21 shows the conversion efficiency for the converter under PM at different mode charge measured by YOKOGAWA WT3000. Efficiency is maximum for an input voltage of 400V when the converter is operating at the resonant frequency. Converter efficiency is higher than 96% even at 500W low load (15% of load) at nominal input voltage. Peak power stage efficiency

= 2.3 kW to 98.1%.

The pie graph of Figure 22 shows the lossy yield of the converter operating at 2.3 kW output power PM. The losses in the power stage can be divided into loss of the primary switch, loss of the drain-source diode switch, and loss of the magnetic component. The drain-source diode accounts for a major loss of 26% of the converter, followed by a transformer (T ₁ ). The difference between the calculated total loss and the measured value still remains, and thus the different parts are marked with the "others" part.

Next, the operation of the converter in the discharge mode will be described as follows.

)(411) 내지 제8스위치(

)(414)은 스위치로서 작용하고, 제1스위치(

)(311) 및 제4스위치(

)(314)의 역병렬 다이오드는 전력을 DC 버스에 전달하는 정류기로서 작용한다. 방전 모드를 위한 동작 파형은 또한 도 23의 (a)-(d)에 제공된다. 요구되는 전압 이득을 달성하기 위해, 주파수는

와

사이에서 감소된다. 도 23의 (a)는

=420V에서 1차 MOSFET의 측정된 파형을 도시한다. 이는 파형의 형상이 정전압(CV) 모드 충전 내에서 거의 유사한 파형임을 나타내고, 이는 넓은 범위의 부하에 대해 제로 전압 스위칭 및 거의 제로 전류 스위칭를 얻는다. 그러나 입력 전압이

=350V,

=250V일 때, 주파수는 공진 주파수(

)로부터 이동하고, 따라서 낮은 부하 조건에서 1차 MOSFET 파형의 형상이 변경된다. MOSFET은 여전히 ????제로 전압 스위칭을 달성하지만, 제로 전류 스위칭 조건을 얻지는 못한다. 턴-오프 전류는 공진 주파수보다 높다. 그러나 상당한 자화 인덕턴스 때문에, 전류 턴-오프는 종래의 LLC 회로와 같이 자화 전류의 피크보다 작다. 도 23의 (c)-(d)에 도시된 바와 같은 완전한 부하 조건 하에서, 입력 임피던스 위상이 거의 제로이기 때문에, 1차 스위치는 완전한 ZVZCS를 달성할 수 있다. 도 24는

=3.3kW인 RM에서 컨버터의 DC 출력 전압과 전류 파형을 도시한다.In the discharge mode, the fifth switch (

) 411 to an eighth switch

) 414 acts as a switch, and the first switch

) 311 and a fourth switch

) 314 acts as a rectifier that transfers power to the DC bus. The operating waveforms for the discharge mode are also provided in Figures 23 (a) - (d). To achieve the required voltage gain, the frequency is

Wow

. 23 (a)

≪ / RTI > = 420V. This indicates that the shape of the waveform is almost a similar waveform in the constant voltage (CV) mode charge, which results in zero voltage switching and almost zero current switching for a wide range of loads. However,

= 350 V,

= 250 V, the frequency is the resonant frequency (

), Thus changing the shape of the primary MOSFET waveform at low load conditions. The MOSFET still achieves zero voltage switching, but does not achieve zero current switching conditions. The turn-off current is higher than the resonant frequency. However, due to the considerable magnetization inductance, the current turn-off is smaller than the peak of the magnetizing current like a conventional LLC circuit. Under full load conditions as shown in Figures 23 (c) - (d), the primary switch can achieve a complete ZVZCS because the input impedance phase is nearly zero. Fig. 24

= 3.3 kW, the DC output voltage and current waveform of the converter are shown.

=2.4kW,

=420V에서 97.9%를 초과한다. 컨버터가 낮은 입력 전압 하에서 작동하면, 효율은 큰 순환 및 턴-오프 손실로 인해 감소하는 경향이 있다. 2.3kW 출력 전력의 방전 모드에서 동작하는 컨버터의 손실 항복은 도 26의 원형 그래프에 도시되었다.The highest efficiency

= 2.4 kW,

= ≪ / RTI > 420V. If the converter is operating at a low input voltage, efficiency tends to decrease due to large cycling and turn-off losses. The lossy yield of the converter operating in the discharge mode of 2.3 kW output power is shown in the circle graph in Fig.

In conclusion, the present invention proposes a new bidirectional LLC converter using an additional LC auxiliary circuit for the vehicle for a power grid system. This converter provides superior performance over conventional bidirectional converters. Power conversion efficiency is improved by ensuring soft switching in all switches and minimizing zero current switching and circulation losses on the primary side. The required voltage gain is still met by employing a small magnetizing inductance, allowing the proposed converter to operate in a narrow frequency range. In the present invention, detailed theory analysis and operation principle are presented. An experimental prototype converter based on the proposed method is implemented to verify the proposed technique. Laboratory test prototype measurements show high efficiency for both PM and RM over medium and high power levels. The peak efficiency was 98.1% under the PM of 2.3 kW and 97.9% under the RM of 2.4 kW.

While the present invention has been particularly shown and described with reference to exemplary embodiments thereof, it will be understood by those skilled in the art that various changes and modifications may be made therein without departing from the spirit and scope of the invention as defined in the appended claims. It will be possible.

)
112: 제1외부인덕터(

)
121: 제2공진커패시터(

)
122: 제2외부인덕터(

)
200: 보조LC공진부
201: 보조커패시터(

)
202: 보조인덕터(

)
300: 제1변환부
310: 제1풀-브리지회로
311: 제1스위치(

)
312: 제2스위치(

)
313: 제3스위치(

)
314: 제4스위치(

)
400: 제2변환
410: 제2풀-브리지회로
411: 제5스위치(

)
412: 제6스위치(

)
413: 제7스위치(

)
414: 제8스위치(

)100:
111: first resonant capacitor (

)
112: first outer inductor (

)
121: a second resonant capacitor (

)
122: second outer inductor (

)
200: auxiliary LC resonance part
201: auxiliary capacitor (

)
202: Auxiliary inductor (

)
300:
310: first full-bridge circuit
311: first switch (

)
312: second switch (

)
313: Third switch (

)
314: fourth switch (

)
400: second conversion
410: second full-bridge circuit
411: fifth switch (

)
412: the sixth switch (

)
413: Seventh switch (

)
414: Eighth switch (

)

Claims (16)

A transformer for performing voltage conversion including a primary winding and a secondary winding;
A secondary LC resonator connected to the transformer through a tertiary winding;
A first switch connected to the primary winding,

) To the fourth switch (

A first conversion unit for transferring the input power transformed by the transforming unit to the output capacitor according to a switching operation of the first full-bridge circuit including the first full-bridge circuit; And
Connected to a battery supplying input power, and a fifth switch

) To eighth switch

And a second converter for transferring the input power to the secondary winding according to the switching operation of the second full-bridge circuit,
Wherein the first conversion unit or the second conversion unit comprises:
A bidirectional converter having a secondary LC resonant circuit operating in a battery charging mode charging the battery or a battery discharging mode discharging the battery.
The battery charging method according to claim 1,

The second switch

And the third switch

) Is turned off under Zero Current Switching (ZCS) conditions.

) ≪ / RTI > of claim 1 or 2, characterized in that the auxiliary LC resonant circuit comprises an auxiliary LC resonant circuit.
The battery charging method according to claim 2,

The second switch

And the third switch

Is turned on under the Zero Voltage Switching (ZVS) condition,

Further comprising a second LC resonant circuit.
4. The method of claim 3,

The fifth switch

And the eighth switch

) Is turned off under zero current switching conditions, and the sixth switch

And the seventh switch

) Is turned on under the zero voltage switching condition

Further comprising a second LC resonant circuit.
5. The method of claim 4,

The first switch

) And the fourth switch (

) Is turned off under zero current switching conditions, and the second switch

And the third switch

) Is turned on under the zero voltage switching condition.

Further comprising a second LC resonant circuit.
6. The method of claim 5,

The second switch

And the third switch

Is turned on with the zero voltage switching generated in the fourth battery charging mode (step < RTI ID = 0.0 >

Further comprising a second LC resonant circuit.
7. The method of claim 6,

The sixth switch

And the seventh switch

) Is turned off under the zero current switching condition, and the fifth switch

And the eighth switch

) Is turned on with a zero voltage switching condition.

Further comprising a second LC resonant circuit.
The method of claim 1, wherein the battery discharge mode comprises:

The sixth switch

And the seventh switch

) Is turned off in the first battery discharge mode (

) ≪ / RTI > of claim 1 or 2, characterized in that the auxiliary LC resonant circuit comprises an auxiliary LC resonant circuit.
The method of claim 8, wherein the battery discharge mode comprises:

The fifth switch

And the eighth switch

) Is turned on, and the first switch

) And the fourth switch (

) Is turned off under the zero-current switching condition in the second battery discharge mode

Further comprising a second LC resonant circuit.

The method of claim 9, wherein the battery discharge mode comprises:

And a third battery discharge mode in which the resonance is stopped in the second conversion unit and the power transmission to the second conversion unit is stopped

Further comprising a second LC resonant circuit.
11. The method of claim 10, wherein the battery discharge mode comprises:

The fifth switch

And the eighth switch

Is turned off, and the sixth switch

And the seventh switch

) Is turned on under the zero voltage switching condition

Further comprising a second LC resonant circuit.
12. The method of claim 11, wherein the battery discharge mode comprises:

The sixth switch

And the seventh switch

) Is turned on in the fifth battery discharge mode

Further comprising a second LC resonant circuit.
13. The method of claim 12, wherein the battery discharge mode comprises:

The sixth switch

And the seventh switch

) Is turned off under the zero current switching condition

Further comprising a second LC resonant circuit.
Wherein a transformer section for performing voltage conversion includes a primary winding and a secondary winding, a secondary LC resonant section is connected to the transforming section through a tertiary winding, and a first converting section connected to the primary winding, And a second conversion unit connected to a battery for supplying an input power supplies the input power according to the switching operation of the second full-bridge circuit, A method of driving a bi-directional converter having an auxiliary LC resonant circuit for transferring a current to a secondary winding,
Wherein the first conversion unit or the second conversion unit has a supplemental LC resonance circuit driven in a battery charging mode for charging the battery or a battery discharge mode for discharging the battery.
15. The method of claim 14,
Wherein the first full-bridge circuit or the second full-bridge circuit is driven in a first battery charging mode to a sixth battery charging mode in response to a turn-on or a turn-off of the switches constituting the first full-bridge circuit or the second full- A method of driving a bidirectional converter having a resonant circuit.
15. The method of claim 14, wherein the battery discharge mode comprises:
Wherein the first full-bridge circuit or the second full-bridge circuit is driven in a first battery discharge mode to a sixth battery discharge mode corresponding to turn-on or turn-off of the switches constituting the first full-bridge circuit or the second full- A method of driving a bidirectional converter having a resonant circuit.

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