KR102009351B1 - High Efficiency LLC Resonant Converter with Balanced Secondary Currents using the Two Transformer Structure - Google Patents

Abstract

The present invention relates to a high efficiency LLC resonant converter having balanced secondary currents using two transformer structures. According to the high efficiency LLC resonant converter of the present invention, a half-bridge power topology with two transformers connected in series / parallel is provided. This solves the problem of unbalance of secondary currents. In the LLC resonant converter of the present invention, the transformer alternately functions as a real transformer including a magnetizing inductance for energy transfer and a resonant inductance for resonance according to an operation mode. The LLC resonant converter of the present invention can effectively vary and adjust the output voltage and operates with ZVS over a wide range of loads. The LLC resonant converter of the present invention has excellent performance as a high efficiency LLC half-bridge resonant converter with balanced secondary side currents.

Description

High Efficiency LLC Resonant Converter with Balanced Secondary Currents using the Two Transformer Structure}

The present invention relates to an inductance-inductance-capacitance (LLC) resonant converter, and more particularly, to a high efficiency LLC resonant converter having a balanced secondary current using two transformer structures.

Many efforts have been made to achieve high power density and high efficiency direct current-direct current (DC-DC) converters and have developed resonant converters capable of operating at high switching frequencies. Among the various types of topologies for resonant converters, the simplest and most widely used is an inductance-capacitance half-bridge series resonant converter (HB-SRC). In the case of HB-SRC, the rectifier load network is arranged in series with the LC resonant network. The load of the resonant network and the HB-SRC acts as a voltage divider where the DC input voltage is distributed between the resonant network impedance and the primary-reflected load to the primary side of the transformer. By varying the switching frequency of the half-bridge switch of the HB-SRC, the impedance of the resonant network changes. Thus, the DC voltage gain of the HB-SRC is always less than one. At light loads, the load impedance is very large compared to the resonant network impedance. Therefore, since most of the DC input voltage is applied to the load reflected to the primary side, it is difficult to adjust the output voltage at light load.

At light loads, LLC resonant converters are used to overcome the limitations of HB-SRC. The LLC resonant converter is a modified LC resonant converter that uses shunt-connected magnetizing inductance in the primary winding of the transformer for resonance. LLC resonant converters are widely used due to the following characteristics. 1) Since both step-up and step-down functions are possible, output voltage regulation is achieved over a wide load with only a small change in operating frequency. 2) Zero-voltage switching (ZVS) of the primary-side MOSFET (Metal Oxide Semiconductor Field Effect Transistor) switch and zero-current switching (ZCS) of the secondary-side diode rectifier are full load. In the range is achieved without an output filter inductor. 3) The magnetic components of a multi-resonant network can be easily integrated into a single magnetic core.

LLC converter topologies with integrated transformers have been proposed and analyzed using a fundamental harmonic approach (FHA). Theoretical analysis suggests soft switching conditions and optimal design methodology. In one paper, the resonant behavior of LLC-SRC in buck mode is studied. The boost characteristics of the LLC resonant converter and the requirements for regulating the output voltage over a narrow frequency range have been proposed in one paper. There are also papers discussing the multiple topologies, optimal performance and basic analysis of LLC converters.

In general, LLC converters incorporate a leakage inductance into the transformer to use a transformer that includes a resonant inductance. Resonant inductance or leakage inductance is realized with a suitable air gap inserted into the transformer core. Thus additional inductors can be eliminated and implementation costs are reduced. However, it is difficult to accurately control the amount of leakage inductance in the manufacturing process. High leakage inductance also allows the transformer to operate at low efficiency and high temperatures.

Secondary center-tapped transformers are widely used to implement commercial LLC resonant converters. Secondary center-tapped transformers have the advantage of lower conduction losses and a smaller number of secondary rectifiers required than in full-bridge construction. However, in spite of these advantages, the unbalance of the physical position between the secondary conductors in the transformer leads to an unbalance of the leakage inductance and an imbalance in the resonant circuit. As a result, an imbalance occurs in the rectifier current and the rectifier stress, and thus the operating efficiency is also reduced.

Accordingly, the present invention has been made to solve the above-described problem, an object of the present invention is to use a half-bridge power topology with two transformers connected in series / parallel, which can solve the secondary current unbalance problem, To provide a high efficiency LLC resonant converter.

First, to summarize the features of the present invention, the LLC resonant converter according to an aspect of the present invention for achieving the above object, the first switch, the resonant capacitor, the first connection in series between the positive terminal and the negative terminal of the input voltage A primary coil of the transformer, a primary coil of the second transformer and a primary total leakage inductor; A first magnetizing inductor connected in parallel to the primary coil of the first transformer; A second magnetization inductor connected in parallel to the primary coil of the second transformer; A second switch connected between the contact point of the first switch and the resonance capacitor and the negative terminal of the input voltage; And a first diode and a second diode in opposite directions, connected in series to form a closed circuit, a first leakage inductor of a secondary side, a secondary coil of the first transformer, a secondary coil of the second transformer, and a secondary side agent. And a second leakage inductor, for outputting a DC voltage between the contacts of the first diode and the second diode and the contacts of the secondary coil of the first transformer and the secondary coil of the second transformer. .

The first switch and the second switch alternately switches on and off, but there is a dead time in which both the first switch and the second switch are turned off between alternating switching, and the ZVS condition of each switch is provided. do.

During one switching period in which the first switch and the second switch are alternated, the first transformer and the second transformer have an original role of transferring energy of the transformer primary side to the secondary side, and resonating with the resonant capacitor. Alternately switch the resonant inductance role.

Preferably, the turns ratio of the first transformer and the second transformer is the same. Preferably, the inductance of the first magnetized inductor and the second magnetized inductor is also the same.

The inductance L _{m1 of} the first magnetized inductor and the inductance L _m2 of the second magnetized inductor are respectively the first equivalent leakage inductance L _kp + n ₂ ² L _ks2 and the second equivalent leakage inductance L _kp + n ₁ ² L _ks1 ) 10 times or more, where n ₁ is the winding ratio of the first transformer, n ₂ is the winding ratio of the second transformer, L _kp is the inductance of the primary side total leakage inductor, and L _ks1 is the secondary side The inductance of the first leakage inductor, L _ks2, is the inductance of the secondary leakage second inductor.

Unbalance of the secondary current due to the difference between the secondary leakage inductance of the first transformer and the second transformer can be eliminated.

The number of primary windings of the first transformer and the second transformer is at least N _{p, min} ,

Here, f _{s, min} is the minimum switching frequency of the first switch and the second switch, ΔB is the maximum magnetic flux density swing of the core of the first transformer and the second transformer, A _e is the first transformer and the The effective cross-sectional area, V _in , of the second transformer core is the input voltage.

The maximum primary current I _{p, max} flowing through the resonant capacitor satisfies the following equation,

Here, C _S is a maximum value of parasitic capacitance of the first switch and the second switch, t _dead is a dead time, and V _in is an input voltage.

The inductance L _m of the first magnetized inductor and the second magnetized inductor satisfies the following equation,

Here, C _S is the maximum value of the parasitic capacitance of the first switch and the second switch, f _{s, max} is the maximum switching frequency of the first switch and the second switch, t _dead is the dead time.

According to the high-efficiency LLC resonant converter according to the present invention, the half-bridge power topology is used together with two transformers connected in series / parallel to solve the secondary current unbalance problem. In the LLC resonant converter of the present invention, the transformer alternately functions as a real transformer including a magnetizing inductance for energy transfer and a resonant inductance for resonance according to an operation mode. The LLC resonant converter of the present invention can effectively vary and adjust the output voltage and operates with ZVS over a wide range of loads. The LLC resonant converter of the present invention has excellent performance as a high efficiency LLC half-bridge resonant converter with balanced secondary side currents.

BRIEF DESCRIPTION OF THE DRAWINGS The accompanying drawings, which are included as part of the detailed description in order to provide a thorough understanding of the present invention, provide examples of the present invention and together with the description, describe the technical idea of the present invention.
1 is a circuit diagram of a typical LLC half-bridge resonant converter.
2 is a circuit diagram of a high efficiency LLC half bridge resonant converter using two transformers connected in series / parallel in accordance with one embodiment of the present invention.
3 is an equivalent circuit of six operating modes of the high efficiency LLC half bridge resonant converter according to an embodiment of the present invention.
4 is a theoretical operating waveform of key portions of a high efficiency LLC half bridge resonant converter in accordance with an embodiment of the present invention.
5 shows an FHA circuit model of the converter of the present invention.
6 is an experimental waveform of a gate-source voltage and a drain-source voltage of each primary side switch of the converter of the present invention.
7 is an experimental waveform of anode-cathode voltage and anode current of secondary side diodes of the converter of the present invention.
8 is an experimental waveform of each output voltage, the primary side current and the secondary side diode current according to the output voltage adjustment at the maximum load condition of the converter of the present invention.
9 is an experimental waveform of primary switch voltage, primary current, output voltage and secondary current at each load condition at maximum output voltage.
10 is a graph of the efficiency of the converter of the present invention.

Hereinafter, with reference to the accompanying drawings will be described in detail with respect to the present invention. In this case, the same components in each drawing are represented by the same reference numerals as much as possible. In addition, detailed descriptions of already known functions and / or configurations are omitted. The following description focuses on parts necessary for understanding the operation according to various embodiments, and descriptions of elements that may obscure the gist of the description are omitted. In addition, some components of the drawings may be exaggerated, omitted, or schematically illustrated. The size of each component does not entirely reflect the actual size, and thus the contents described herein are not limited by the relative size or spacing of the components drawn in the respective drawings.

In describing the embodiments of the present invention, when it is determined that the detailed description of the known technology related to the present invention may unnecessarily obscure the gist of the present invention, the detailed description thereof will be omitted. In addition, terms to be described below are terms defined in consideration of functions in the present invention, which may vary according to the intention or custom of a user or an operator. Therefore, the definition should be made based on the contents throughout the specification. The terminology used in the description is for the purpose of describing particular embodiments only and should not be limiting. Unless expressly used otherwise, the singular forms “a,” “an,” and “the” include plural forms of meaning. In this description, expressions such as "comprises" or "equipment" are intended to indicate certain features, numbers, steps, actions, elements, portions or combinations thereof, and one or more than those described. It should not be construed to exclude the presence or possibility of other features, numbers, steps, actions, elements, portions or combinations thereof.

In addition, terms such as first and second may be used to describe various components, but the components are not limited by the terms, and the terms are used to distinguish one component from another component. Used only as

1 is a circuit diagram of a typical LLC half-bridge resonant converter.

Referring to Figure 1, a typical LLC half-bridge resonant converter using a secondary side center-tapped transformer uses the magnetizing inductance (L _m ) of the transformer for a different resonance than the HB-SRC. Therefore, the voltage gain characteristic of the LLC resonant converter is different from the HB-SRC since it is modified compared to the voltage gain characteristic of the HB-SRC. The HB-SRC can only operate in buck mode, while the LLC resonant converter can operate in both buck and boost modes. If the switching frequency is higher than the resonant frequency, the voltage gain of the LLC resonant converter is always less than 1 and behaves like HB-SRC with ZVS achieved. However, if the switching frequency is lower than the resonant frequency, the voltage gain can be greater than 1 and both ZVS and ZCS are achieved. The converter voltage gain is maximized at the boundary of ZVS and ZCS operation.

Since the impedance of the resonant tank at the resonant frequency becomes zero, the input and output sides are virtually connected to each other. In general, the input voltage of the DC-DC stage is generated by a power factor correction (PFC) circuit and adjusted to a reference voltage. Under these conditions, the converter can always operate at the resonant frequency by selecting the appropriate transformer turns ratio. Therefore, conduction loss and switching loss can be minimized. In addition, during holdup, energy is transferred from the bulky output holdup capacitor to the load. At this time, when the DC input voltage is lowered, the converter lowers the switching frequency to operate in boost mode, so the converter adjusts the output voltage.

2 is a circuit diagram of a high efficiency LLC half bridge resonant converter using two transformers connected in series / parallel in accordance with one embodiment of the present invention.

2, a high efficiency LLC half bridge resonant converter according to an embodiment of the present invention may include a first transformer T ₁ , a second transformer T ₂ , a first MOSFET switch Q ₁ , and a second MOSFET switch ( Q ₂ ), the capacitance (C _r ) of the resonant capacitor, the inductance (L _kp ) of the primary total leakage inductor of the transformer (T ₁ , T ₂ ), the inductance (L _m1 ) of the first magnetizing inductor, inductance (L _m2), the transformer (T _1, T ₂₎ second inductance (L _ks1) of the primary side first leakage inductor of the transformer inductance (L _ks2) of the secondary-side second leakage inductor of (T _1, T _2), The first diode D _o1 , the second diode D _o2 , the capacitance C _o of the output capacitor, and the output load resistor _Ro are included.

As shown in FIG. 2, the primary coil of the first transformer T ₁ and the primary coil of the second transformer T ₂ are connected in series, and the first magnetizing inductor (T) is connected to the primary coil of the first transformer T ₁ . L _m1 ) is connected in parallel, and the second magnetization inductor L _m2 is connected in parallel to the primary coil of the second transformer T ₂ .

Between the positive and negative terminals of the input voltage (V _in ), the first switch (Q ₁ ), the resonant capacitor (C _r ), the primary coil of the first transformer (T ₁ ), 1 of the second transformer (T ₂ ) claim the primary coil, between the primary total leakage inductor (L _kp) is a serial connection configuring the closed loop, and the first switch (Q ₁₎ and the resonant capacitor (C _r) of the negative terminal of the contact and the input voltage (V _in) 2 switch (Q ₂ ) is connected. The first switch Q ₁ and the second switch Q ₂ are described as an N-type MOSFET structure by way of example. However, it is obvious that the same function can be performed by changing the first switch Q ₁ and the second switch Q ₂ into a P-type structure and partially changing the operating conditions.

In addition, the first diode (D _o1 ) and the second diode (D _o2 ) in the opposite directions, the secondary leakage first inductor (L _ks1 ), the secondary coil of the first transformer (T ₁ ), the second transformer (T) ₂ ) the secondary coil of the secondary side, the secondary leakage inductor (L _ks2 ) of the secondary side is connected in series to form a closed circuit, the contact of the first diode (D _o1 ) and the second diode (D _o2 ) and the first transformer (T) ₁₎ providing a secondary coil and a second transformer (T _2, the DC voltage between the secondary side coil and contacts, i.e. the load, parallel-connected output capacitor (C _o) and the output load resistance (R _o in)).

Each of the transformers T ₁ and T ₂ alternates one switching of the first switch Q ₁ and the second switch Q ₂ together with the magnetizing inductances L _m1 and L _m2 connected in parallel to the primary side. During the cycle, it alternates between the original role of the transformer (T ₁ , T ₂ ), which transfers the energy of the transformer primary to the secondary side, and the other role of the resonant inductance for resonance and the capacitance (C _r ) of the resonant capacitor. . Switching control of the first switch Q ₁ and the second switch Q ₂ may be performed by a predetermined control device. The controller may provide switching control or the like that is suitable for ZVS conditions for dead time and the like as described below.

The coil turns ratio of each transformer T ₁ , T ₂ is n ₁ (= N _p1 / N _s1 ) and n ₂ (= N _p2 / N _s2 ). N _p1 , N _p2 and N _s1 , N _s2 are the primary and secondary windings of each transformer T ₁ , T ₂ . L _m1 , L _m2 and L _ks1 , L _ks2 are magnetization inductances and secondary leakage inductances of the transformers T ₁ and T ₂ , respectively, and L _kp is the total primary equivalent leakage inductance. C _r is the resonant capacitance, C ₁ , C ₂ and D ₁ , D ₂ are the parasitic capacitance and the body diode of the _first MOSFET switch Q ₁ and the second MOSFET switch Q ₂ , respectively, which are power semiconductor switches. The two switches Q ₁ and Q ₂ operate with variable switching frequency and 50% duty ratio.

3 is an equivalent circuit of six operating modes of the high efficiency LLC half bridge resonant converter according to an embodiment of the present invention.

4 is a theoretical operating waveform of key portions of a high efficiency LLC half bridge resonant converter in accordance with an embodiment of the present invention.

First, referring to FIGS. 3A and 4, the time t ₀ to t ₁ section of the operation of the first mode.

MOSFET switch Q ₁ turns on to ZVS at time t = t ₀ before the direction of primary current i _p of transformer T ₁ , T ₂ is changed. Secondary diode D _{o1 is} also turned on and current i _Do1 flows. The primary side energy is transferred to the secondary side via transformer T ₁ . At the end of this mode, the secondary diode current i _Do1 is zero. The total primary side equivalent series resonant inductance L _r1 (= L _m2 + L _kp + n ₁ ² L _ks1 ) in the first mode resonates with the resonant capacitance C _r . At this time, the magnetizing inductance L _m2 of the transformer T ₂ operates as a resonant inductance together with another series resonant inductance (= L _kp + n ₁ ² L _ks1 ). The magnetization current i _Lm1 increases linearly, and the primary side current i _p (= i _Lm2 ) flows in a sinusoidal shape. The primary current i _p , the magnetization current i _Lm1, and the voltage v _Cr of the resonant capacitor are expressed as follows.

[Equation 1]

[Equation 2]

[Equation 3]

,

은 각각 제1모드에서의 공진 각 주파수 및 특성 임피던스이다. here,

,

Are the resonant angular frequencies and characteristic impedances in the first mode, respectively.

Next, referring to FIG. 3B and the time interval t ₁ to t _{2 in} FIG. 4 for the operation of the second mode.

At time t = t _1, the primary current i 1 _p i and the magnetizing current is _Lm1 is equal. This mode is a freewheeling mode. _Secondary diode current i _Do1 goes to zero and diode D _o1 is turned off by ZCS to minimize reverse recovery loss of diode D _o1 . Since the secondary side output diodes D _o1 and D _o2 are turned off, the output is separated from the transformers (T ₁ , T ₂ ), and the total primary side equivalent series resonant inductance L _rf (= L _m1 + L _m2 + L _kp ) in this mode Resonance with the resonance capacitance C _r . Therefore, the primary current i _p is calculated as follows.

[Equation 4]

,

는 각각 각각 제2모드에서의 공진 각 주파수 및 특성 임피던스이다. here,

,

Are respectively the resonant angular frequency and characteristic impedance in the second mode.

Next, referring to (c) of FIG. 3 and the time t ₂ to t ₃ section in FIG. 4 for the operation of the third mode.

At time t = t ₂ , MOSFET switch Q ₁ is turned off, but primary side current i _p continues to flow in the positive direction. At this time, the primary current i _p charges and discharges the parasitic capacitances C ₁ and C ₂ of the MOSFET switches Q ₁ and Q ₂ , respectively. The body diode D ₂ of the switch Q ₂ is then turned on and the primary current i _p flows through the diode D ₂ to create a ZVS condition for the switch Q ₂ . The gate signal v _{Q2, gs} of the switch Q ₂ must be applied before the direction of the primary current i _p is changed. This mode is a dead-time mode.

Next, referring to FIG. ₃ (d) and time t ₃ to t _{4 in} FIG. _{4 for} the operation in the fourth mode.

At time t = t ₃ , MOSFET switch Q ₂ turns on to ZVS. The secondary rectifier diode _DO2 also turns on. The primary side energy is transferred to the secondary side via transformer T ₂ . The primary side equivalent series resonant inductance L _r4 (= L _m1 + L _kp + n ₂ ² L _ks2 ) in the fourth mode resonates with the resonant capacitance C _r . At this time, the magnetizing inductance L _m1 of the transformer T _{1 operates} as a resonant inductance together with another series resonant inductance (= L _kp + n ₂ ² L _ks2 ). The magnetization current i _Lm2 decreases linearly. However, due to the resonance of the primary inductance L _r4 and the resonant capacitance C _r, the primary current i _p (= i _Lm1 ) changes direction and flows in sinusoidal form during this mode. The primary side current i _p and the magnetizing current i _Lm2 are given by

[Equation 5]

[Equation 6]

,

는 각각 제4모드에서의 공진 각 주파수 및 특성 임피던스이다. here,

,

Are the resonant angular frequencies and characteristic impedances in the fourth mode, respectively.

Next, referring to FIG. 3E and the periods t ₄ to t ₅ of FIG. 4 for the operation of the fifth mode.

This mode is similar to the second mode. At time t = t ₄ , the secondary output diode current i _Do2 is zero, diode D _o2 is turned off to ZCS, and voltage stress and reverse recovery loss of diode D _o2 are also minimized. The two magnetizing currents i _Lm1 , i _Lm2 become equal. Since the secondary side output diodes D _o1 and D _o2 are both turned off, the output is separated from the transformer and the overall primary side equivalent series resonant inductance L _rf in this mode is equal to the series resonant inductance in the second mode, with the resonance capacitance C _r and Resonates. Therefore, the primary side current i _p is calculated as follows.

[Equation 7]

Here, ω _rf and Z _rf are resonance angular frequencies and characteristic impedances in the fifth mode, respectively, and are equal to the values in the second mode.

Next, referring to FIG. 3 (f) and the time t ₅ to t ₆ section of FIG. 4 for the operation of the sixth mode.

At time t = t ₅ , MOSFET switch Q ₂ is turned off. Primary current i _p is the discharge of the parasitic capacitance C ₁ of the MOSFET switches Q ₁ and charge the parasitic capacitance C ₂ of the switch Q _2. Then, the primary current i _p is caused to flow through the body diode D ₁ of the switch Q ₁ makes the ZVS condition for switch Q _1. The gate signal v _{Q1, gs} of the switch Q ₁ must be applied before the direction of the primary current i _p is changed. This mode is similar to the third mode.

When the switch Q ₁ is turned on, one operation cycle ends and the next operation cycle starts again.

Concept of High Efficiency LLC Half Bridge Resonant Converter of the Present Invention

The center tap structure of the transformer secondary side of the conventional LLC resonant converter has less conduction loss and fewer output rectifiers than the full-bridge structure, resulting in low manufacturing cost. However, because there is an imbalance in the physical position between the secondary winding conductors of the transformer, an imbalance between leakage inductances is caused, resulting in an imbalance of the resonant circuit. As a result, this difference can lead to an imbalance of rectifier current and rectifier stress, which reduces converter efficiency. Thus, to overcome these drawbacks, a new concept for LLC resonant converters is required. The proposed converter uses two transformers (T ₁ , T ₂ ) connected in series / parallel in the converter power circuit.

Each transformer T ₁ , T ₂ has a winding with the same turns ratio (n = n ₁ = n ₂ ) (n ₁ is the turns ratio of T ₁ , n ₂ is the turns ratio of T ₂ ), and transformers T ₁ , T ₂₎ each of the magnetizing inductance (L _{m1, m12} L) are substantially equal to each other (L = L _m L = _{m1 m2)} and much larger than the other equivalent leakage inductance (L _m1 »L _kp + n ₂ L ² and L _{ks2 m2} »L _kp + n ₁ ² L _ks1 ) Designed and implemented (eg, each magnetizing inductance is 10 times greater than its other equivalent leakage inductance). In addition, the converter of the present invention uses an actual transformer including a magnetizing inductance for energy transfer and two transformers alternately operating as resonance inductances for resonance according to an operation mode. Therefore, the primary side and secondary side currents i _p1 and i _p4 of the first mode and the fourth mode can be approximated as follows, respectively.

[Equation 8]

[Equation 9]

,

은 각각 근사화되고 수정된 공진 각 주파수 및 특성 임피던스이다. here,

,

Are approximated and modified resonant angular frequencies and characteristic impedances, respectively.

However, according to FIG. 4, the currents of the first diode D _o1 and the second diode Do _o2 , the secondary rectifiers, flow only in the first mode or the fourth mode. In the rectifier currents i _Do1 and i _Do2 of the first mode and the fourth mode, the difference between the approximated primary side current (Equation 8 or Equation 9) and the magnetization current of the energy transfer transformer is equal to the transformer turns ratio n. Corresponding to each converted current as follows.

[Equation 10]

[Equation 11]

Here, the magnetization currents i _Lm1 and i _Lm2 are modified from Equations 2 and 6 and given as follows.

[Equation 12]

[Equation 13]

Therefore, the converter of the present invention is hardly affected by the leakage inductance of the transformers T ₁ and T ₂ , and easily solves the problem of unbalanced secondary currents due to the difference in leakage inductance of the transformer secondary side in the conventional LLC resonant converter. do. In addition, the two transformers (T _1, T ₂₎ the transformer will pass by dividing the primary total energy of the converter of the invention (T _1, T ₂₎ to be mitigating the thermal problems converters of the present invention, low-profile type of using a Can be implemented as (low profile type).

<Steady state analysis>

The fundamental harmonic approach (FHA) makes it easy to obtain the converter voltage gain under input conditions (V _in ) and load conditions at steady state. The FHA model is used assuming that only the fundamental wave voltage component of the square wave input source for the resonant network contributes to the power delivery to the output. The voltage gain is a function of the ratio of the operating (switching) frequency to the resonant frequency in the selected resonant network. The fundamental voltage component v _inf of the primary input source and its root mean square (rms) _, V _{inf, rms,} are given by

[Equation 14]

[Equation 15]

Where ω _s = 2πf _s is the switching angular frequency.

5 shows an FHA circuit model of the converter of the present invention. The primary circuit can be replaced by a sinusoidal current source i _ac with a secondary voltage v _of at the input of the secondary rectifier. Since the average value of | i _ac | is equal to the output current I _o , the sinusoidal current source i _ac and its rms value I _{ac, rms} can be expressed as follows.

[Equation 16]

[Equation 17]

The fundamental voltage component v _of and its rms value V _{of, rms} at the secondary output can be calculated as

Equation 18

[Equation 19]

는 출력 전압의 지연 위상이다. 다음에, 변압기 권선비 n을 고려할 때 AC 등가 부하 저항 R_ac는 다음과 같이 계산된다.here,

Is the delay phase of the output voltage. Next, considering the transformer turns ratio n, the AC equivalent load resistance R _ac is calculated as follows.

[Equation 20]

The input-output transfer function H (s) of the FHA model is given by

[Equation 21]

The magnitude of the gain is calculated from Equation 21 as follows.

[Equation 22]

From Equations 14 to 21, the input-output relationship of the LLC resonant converter of the present invention is as follows.

 [Equation 23]

In addition, the magnitude of the gain M (f _n ) represented by the normalized frequency factor f _n is

[Equation 24]

Where f _n = f _s / f _r is the ratio of the switching frequency f _s to the resonant frequency f _r , Q = Z _r / R _ac quality factor, k = L _r / L _m is the resonance inductance (L is the ratio of inductance between _r ) and the magnetizing inductance (L _m ).

Assuming that the converter of the present invention operates at the resonant frequency in the 1-3 mode, Equation 1 or Equation 8 can be approximately changed as follows.

[Equation 25]

이며, I_p,rms는 1차측 전류의 rms값이다. 이외에, 1차측 전류 i_p (=i_Lm2)와 자화 전류 i_Lm1의 차이가 부하로 전달되므로, 평균 출력 전류 I_o는 다음과 같이 계산될 수 있다.here,

Where I _{p, rms} is the rms value of the primary current. In addition, since the difference between the primary side current i _p (= i _Lm2 ) and the magnetizing current i _Lm1 is transferred to the load, the average output current I _o can be calculated as follows.

[Equation 26]

From equations (25) and (26), the rms values I _{p, rms} of the primary side current are obtained as follows.

[Equation 27]

From equations (2), (26) and (27), the rms value I _{Do, rms} of the secondary diode current is derived as follows.

[Equation 28]

<Design Considerations>

Generally, LLC resonant converters have two resonant frequencies, one high and one low. However, LLC resonant converters are typically designed to operate at the higher of the two resonant frequencies for full load and maximum DC input voltage. As the load increases or the input voltage decreases, the operating frequency is reduced by the feedback control circuit to regulate the output voltage.

If the turn-on duty D of the primary MOSFET switch is chosen as the maximum possible nominal value (D = D _mominal = 0.49), the turns ratio n of each transformer is calculated by the input / output voltage relationship as follows.

[Equation 29]

The worst case of a transformer design is the minimum switching frequency condition that occurs at minimum input voltage (V _in = V _{in, min} ) and full load conditions. The minimum number of primary turns N _{p, min} can then be calculated by the relationship between inductance and magnetic flux as

[Equation 30]

Here, f _{s, min} is the minimum switching frequency of the gate control signal of the first switch (Q ₁ ) and the second switch (Q ₂ ), ΔB is the maximum magnetic flux density swing of the transformer (T ₁ , T ₂ ) core, A _e is the effective cross-sectional area of the transformer (T ₁ , T ₂ ) core.

The maximum current I _{Do, max} of the secondary output diode is roughly calculated using the following equations (16), (17) and [26].

Equation 31

In addition, the primary maximum current I _{p, max} is calculated as follows using Equation 31 and the transformer turns ratio n.

[Equation 32]

Therefore, the current ratings of the primary MOSFET switches and the secondary rectifier diodes are determined using Equations 31 and 32.

The primary current i _p is always positive at times t = t ₂ and t ₅ and causes primary MOSFET switches Q ₁ and Q ₂ to operate with ZVS. During the commutation period of the primary switches, it can be assumed that the magnetizing currents i _Lm1 and i _Lm2 are constant because of the large magnetizing inductance. To ensure the ZVS of each switch _, the parasitic nature of the primary switch is the maximum magnetizing current or maximum primary current I _{p, max} (the maximum value of the current flowing through the capacitance C _r of the resonant capacitor) It must be possible to discharge the capacitance completely.

[Equation 33]

Where C _S is the maximum value of the parasitic capacitances C ₁ , C ₂ of the primary switches (C ₁ , C ₂ may be the same and the maximum may be the same), and t _dead is the dead time. Of the gates of Q ₁ and Q ₂ , Between alternating switching times, a logic-low voltage is applied to the Q ₁ and Q ₂ gates to turn them all off and provide a ZVS condition. T _s is a switching period of the gate control signal of the first switch Q ₁ and the second switch Q ₂ .

To ensure minimal switching losses, the magnetizing inductance must be small enough to achieve ZVS and large enough to conduct smaller turn off currents from Eqs. Therefore, the magnetizing inductance must meet the following conditions so that the rectifier diode can operate as ZCS when the primary turn-off current meets the ZVS requirements and the switching frequency is lower than the high resonant frequency.

[Equation 34]

Where f _{s, max} is the maximum switching frequency.

The resonance capacitance C _r is derived as follows.

[Equation 35]

Here, f _r and Z _r are resonance frequencies and characteristic impedances, respectively.

<Experiment Result>

In order to show the effectiveness of the converter of the present invention, the circuit parameters of the prototype converter are designed with the following specifications based on the principles and design considerations of the present invention as described above. In other words, the input voltage V _in = 380 V _DC , the switching frequency f _s = 85-170 kHz, the output voltage V _o = 15-35 V _DC , and the output power P _o = 280 W.

Here, the input voltage is supplied by the PFC circuit and regulated to the reference voltage V _in = 380V _DC . The designed circuit parameters are shown in [Table 1]. The prototype converter was implemented using the circuit parameters shown in Table 1 and the experiment was performed.

Table 1. Designed Circuit Parameters of Prototype

6 is an experimental waveform of the gate-source voltage and drain-source voltage of each primary side switch of the converter of the present invention. 7 is an experimental waveform of anode-cathode voltage and anode current of secondary side diodes of the converter of the present invention.

6, the gate of the primary switch, each of the converter of the present invention to source voltage v _{Q1, gs,} v _{Q2, gs} and the drain-source voltage v _C1, v Referring to experiment waveform of _C2, the primary side switch current i _Q1 , i _Q2 flows backward through the body diodes (D ₁ , D ₂ ) for a short time before switching on. This means that the main switch is switched to ZVS.

Referring to the experimental waveforms of the anode-cathode voltages v _Do1 and v _Do2 and the anode currents i _Do1 and i _Do2 of the secondary side diode of the converter of FIG. 7, the secondary side diode is switched to ZCS and the secondary side of each transformer. There is a 10-fold difference between leakage inductances, but the current flows about the same. 6 and 7 show that the converter of the present invention operates with ZVS and ZCS to improve the overall efficiency and solve the unbalance of secondary currents.

8 is an experimental waveform of each output voltage, the primary side current and the secondary side diode current according to the output voltage adjustment at the maximum load condition of the converter of the present invention. 8 shows experimental waveforms of the output voltage V _o , the primary side current i _p and the secondary side diode current i _Do1 at each adjustment voltage according to the change of the output voltage at full load.

9 is an experimental waveform of primary switch voltage, primary current, output voltage and secondary current at each load condition at maximum output voltage. In Fig. 9, the first switch voltage v _C1 , the primary side current i _p , the output voltage V _o and the secondary side current i _Do2 at each load at the maximum output voltage V _o = 35V are shown.

8 and 9 show that the converter of the present invention can adjust the output voltage accurately under all load conditions.

10 is a graph of the efficiency of the converter of the present invention. Fig. 10A shows the efficiency according to the output voltage adjustment at maximum load, and Fig. 10B shows the efficiency according to the load variation at the maximum output voltage. Referring to the efficiency graph according to the output voltage adjustment at the maximum load condition and the load variation at the maximum output voltage of FIGS. 10A and 10B, the converter of the present invention has a high efficiency under all the output voltage and the load condition. You can see it works.

As described above, a high efficiency LLC resonant converter with balanced secondary currents using two transformer structures according to the present invention is a half-coiler with two transformers connected in series / parallel to solve the secondary current unbalance problem. Use a bridge power topology. The two transformers are used alternately as resonant inductances for resonance and real transformers, including magnetizing inductances for energy transfer, depending on each mode of operation. The converter of the present invention has ZVS operating characteristics over a wide load range as well as a variable output voltage function. It was found that the converter of the present invention has good performance as a high efficiency LLC half-bridge resonant converter with secondary side current balance.

In the present invention as described above has been described by the specific embodiments, such as specific components and limited embodiments and drawings, but this is provided to help a more general understanding of the present invention, the present invention is not limited to the above embodiments. For those skilled in the art, various modifications and variations may be made without departing from the essential features of the present invention. Accordingly, the spirit of the present invention should not be limited to the described embodiments, and all technical ideas having equivalent or equivalent modifications to the claims as well as the following claims are included in the scope of the present invention. Should be interpreted as.

Transformer (T ₁ , T ₂ )
MOSFET switches (Q ₁ , Q ₂ )
Capacitance of the resonant capacitor (C _r ),
Inductance of the primary leakage total inductor (L _kp ),
Inductance of the magnetized inductor (L _m1 , L _m2 )
Inductance of the secondary side leakage inductor (L _ks1 , L _ks2 )
Diode (D _o1 , D _o2 )
Capacitance of Output Capacitor (C _o )
Output load resistance (R _o )

Claims (10)

A first switch, a resonant capacitor, a primary coil of the first transformer, a primary coil of the second transformer, and a primary total leakage inductor connected in series between the positive terminal and the negative terminal of the input voltage; A first magnetizing inductor connected in parallel to the primary coil of the first transformer; A second magnetization inductor connected in parallel to the primary coil of the second transformer;
A second switch connected between the contact point of the first switch and the resonance capacitor and the negative terminal of the input voltage; And a first diode and a second diode in opposite directions, connected in series to form a closed circuit, a first leakage inductor of a secondary side, a secondary coil of the first transformer, a secondary coil of the second transformer, and a secondary side agent. A secondary leakage center tap for outputting a DC voltage between a contact of the first diode and the second diode and a contact of the secondary coil of the first transformer and the secondary coil of the second transformer; As a structure,
In order to eliminate the unbalance of the secondary current due to the difference between the secondary leakage inductances due to the physical imbalance of the secondary transformer of the first transformer and the second transformer,
The number of primary turns of the first transformer and the second transformer is _{equal to or} greater than the minimum value N _{p, min} , where

ego,
The maximum primary current I _{p, max} flowing through the resonant capacitor is

Satisfying,
The first and second switches are all turned off between the alternating switching of the first and second switches by setting the turn-on duty ratio of the first and second switches as the maximum nominal value, rather than the asymmetric pulse width modulation scheme. There is a dead time at which the ZVS condition of each switch is provided, thereby eliminating the unbalance of the stress of the secondary rectifier diodes by eliminating the unbalance of the secondary current caused by the difference between the secondary leakage inductances, resulting in converter efficiency. As to improve
Here, f _{s, min} is the minimum switching frequency of the first switch and the second switch, ΔB is the maximum magnetic flux density swing of the core of the first transformer and the second transformer, A _e is the first transformer and the The effective cross-sectional area of the second transformer core, V _in is the input voltage, C _S is the maximum value of the parasitic capacitance of the first switch and the second switch, and t _dead is the dead time. delete The method of claim 1,
During the first switching period in which the first switch and the second switch are alternated, the first transformer and the second transformer are:
LLC resonant converter to switch between the original role of transmitting the energy of the primary side of the transformer to the secondary side and the resonant inductance role for the resonance with the resonant capacitor. The method of claim 1,
The LLC resonant converter of which the turns ratio of the first transformer and the second transformer is the same. The method of claim 1,
And an inductance of the first magnetized inductor and the second magnetized inductor is the same. The method of claim 1,
The inductance L _{m1 of} the first magnetized inductor and the inductance L _m2 of the second magnetized inductor are respectively the first equivalent leakage inductance L _kp + n ₂ ² L _ks2 and the second equivalent leakage inductance L _kp + n ₁ ² L _ks1 ) 10 times or more, where n ₁ is the winding ratio of the first transformer, n ₂ is the winding ratio of the second transformer, L _kp is the inductance of the primary side total leakage inductor, and L _ks1 is the secondary side LLC inductance of the first leakage inductor, L _ks2 is the inductance of the second leakage inductor secondary LLC. delete delete delete The method of claim 5,
The inductance L _m of the first magnetized inductor and the second magnetized inductor satisfies the following equation,

Where, C _S is the maximum value of the parasitic capacitance of the first switch and the second switch, f _{s, max} is the maximum switching frequency of the first switch and the second switch, t _dead is the dead time LLC LLC converter.

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