CN114257097B - Wide-output direct-current converter with multi-mode switching and switching control thereof - Google Patents
Wide-output direct-current converter with multi-mode switching and switching control thereof Download PDFInfo
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- CN114257097B CN114257097B CN202111588111.5A CN202111588111A CN114257097B CN 114257097 B CN114257097 B CN 114257097B CN 202111588111 A CN202111588111 A CN 202111588111A CN 114257097 B CN114257097 B CN 114257097B
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Classifications
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/14—Arrangements for reducing ripples from dc input or output
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
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- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
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Abstract
The invention discloses a multimode switching wide-output direct-current converter and switching control thereof, comprising a coupling inductance circuit, a staggered parallel circuit, a clamping capacitor, a resonant cavity, a transformer, a rectifier and an output capacitor, wherein the switching control is realized by adjusting S 1 、S 2 、S 3 、S 4 The duty cycle controls the output voltage. The multimode switched wide output dc converter of the present invention can operate in three modes, namely a High Gain (HG) mode, a Medium Gain (MG) mode and a Low Gain (LG) mode. According to the invention, under different modes, the coupling inductance circuit not only plays a role of a filter inductance, but also plays a role of a transformer, so that the power density is improved, and meanwhile, the direct current converter can realize zero-voltage switching on of all switching tubes in a wide input working range under an HG mode and an MG mode, so that the switching loss is effectively reduced, and the conversion efficiency is improved.
Description
Technical Field
The invention relates to the technical field of switching converters, in particular to a method for realizing wide-range work of a direct current converter in the technical field of power electronics.
Background
With the rapid development of complex applications such as renewable energy power generation, hybrid electric vehicles, submarine cable observation stations and the like, the need for realizing high-efficiency electric energy conversion in a wide range is becoming more and more urgent. Wide voltage gain dc converters have become an integral part of this type of converter. High efficiency, high power density, wide voltage gain, low cost, and reliable and stable operation are key requirements for such converters. In addition, special requirements such as low current ripple and the like are even key technical indexes of the submarine cable observation station.
LLC resonant converter has the characteristics of zero voltage switching on, high power density and the like of a main power tube, and is widely applied to high-efficiency electric energy converters, but when the traditional LLC resonant converter is applied to a wide-range output field, a Frequency Modulation (FM) or Pulse Width Modulation (PWM) control method faces the over-wide frequency range and is difficult to realize ZVS in a light load state, and the efficiency of the converter is reduced by larger reactive circulation, so that the application requirements of a wider working range cannot be met.
In addition, in the traditional voltage type direct current converter, the voltage stress of a switching tube is relatively fixed, the output voltage is controlled intuitively, but a short-circuit protection circuit with quick action time is needed, and the input current ripple is large, so that the voltage type direct current converter is not suitable for occasions requiring long service life of equipment such as submarine cable observation stations.
Disclosure of Invention
Aiming at the defects of the prior art, the invention provides a direct current converter working in a wide range and a control method thereof, which realize zero voltage on in the wide range working, reduce the switching loss of the converter and realize the electric energy conversion of low current ripple at the input end.
In order to solve the technical problems, the invention adopts the following technical scheme:
the wide output direct current converter with multi-mode switching comprises a coupling inductance circuit 10, wherein the coupling inductance circuit 10 is electrically connected with a staggered parallel circuit 20 and a resonant cavity 30, and the resonant cavity 30 is sequentially electrically connected with a transformer 40, a rectifier 50 and an output capacitor; the coupled inductor circuit 10 is electrically connected with the rectifier 50;
the coupled inductor circuit 10 includes a winding of one N p1 And winding twoN p2 Winding one N p1 And winding two N p2 Is electrically connected with a first diode D after being connected in parallel 1 First diode D 1 Three N windings arranged in parallel are electrically connected p1 Sum winding four N p2 The method comprises the steps of carrying out a first treatment on the surface of the Winding three N p1 Comprising an excitation inductance L equivalent to a coupling inductance m1 And with a first diode D 1 Connected leakage inductance L s1 The method comprises the steps of carrying out a first treatment on the surface of the Four N of the winding p2 Comprises two exciting inductances L with equivalent coupling inductances m2 And with a first diode D 1 Connected leakage inductance two L s2 The method comprises the steps of carrying out a first treatment on the surface of the Winding three N p1 Sum winding four N p2 Electrically connected to the interleaved parallel circuit 20; the interleaved parallel circuit 20 includes two half-bridge structures including a first bridge arm and a second bridge arm, and the interleaved parallel circuit 20 is connected in parallel with a clamp capacitor C c The method comprises the steps of carrying out a first treatment on the surface of the The resonant cavity 30 comprises resonant inductors L connected in series r Exciting inductance L m And a resonance capacitor C r 。
Further improvement, the coupled inductor circuit 10 further includes a coupled inductor-N electrically connected to the rectifier 50 s1 And coupled inductance two N s2 Coupling inductance N s1 And coupled inductance two N s2 Are connected in series, and the inductance value and the transformation ratio of the windings are the same.
Further improvement, the coupling inductance is N s1 And coupled inductance two N s2 After the heterogeneous terminals are connected, the coupling inductor is N s1 Diode two D is connected in series in proper order 2 And output inductance L f The method comprises the steps of carrying out a first treatment on the surface of the Diode two D 2 Electrically connected diode tri-D 3 Cathode, diode tri-D 3 Is electrically connected with the anode of the coupling inductor II N s2 And an output negative terminal.
Further improvement, the first bridge arm comprises a first switching tube S1 and a third switching tube S2 which are connected in series; the second bridge arm comprises a switching tube II S2 and a switching tube IV S4 which are connected in series.
The application method of the multimode switched wide-output direct-current converter comprises the following steps:
s1, collecting input voltage V of a direct current converter i Output voltage V o ；
S2、Input voltage V to be collected i With a given output voltage command value V o * The controller fed into the DC converter carries out mode judgment and confirms the adjustment control quantity:
s21, when the given output voltage command value V o * Greater than or equal to input voltage V i Boundary value M of gain with HG mode HGB The product of (3) and the converter works in HG mode;
s22, when the given output voltage command value V o * Less than the input voltage V i With HG modal gain M HGB And is greater than or equal to the input voltage V i Boundary value M with MG mode gain MGB The product of (2), the converter works in MG mode;
s23, when the given output voltage command value V o * Less than the input voltage V i Boundary value M with MG mode gain MGB The transformer works in LG mode;
s3, outputting the collected output voltage V o Subtracting the given voltage command value Vo to obtain a voltage error, sending the voltage error to a proportional-integral controller, and sending the output Vcon of the proportional-integral controller to the controller for duty ratio adjustment to obtain PWM waves meeting corresponding D, ds and beta; wherein D represents a duty ratio, ds represents a control amount introduced in the MG mode, and β represents a phase shift angle between the first arm and the second arm;
s4, sending PWM waves to a driving circuit of the direct current converter to obtain a first S of respectively controlled switching tubes 1 Switch tube two S 2 Three S of switch tube 3 And switch tube four S 4 Is a driving signal g of (2) 1 、g 2 、g 3 、g 4 。
When the direct current converter works in the HG mode, the phase shift beta between the switch driving signals in the first bridge arm and the second bridge arm in the staggered parallel network is 180 degrees, the phase shift beta is complementary with the switch driving signals in the bridge arm, and the output voltage is regulated by regulating the duty ratio D of the switch tubes on the first bridge arm and the second bridge arm.
Further improved, when the direct current converter works in HG mode, the duty ratio D adjustment interval is 0.36-0.7.
Further improvement, when the direct current converter works in an MG mode, the phase shift beta between switch driving signals in a first bridge arm and a second bridge arm in a staggered parallel network is 180 degrees, and a control variable D is introduced s ，D s Is a switching tube two S 2 Is a driving signal g of (2) 2 S with switch tube 1 A phase difference of the driving signal g 1; three S switch tube 3 And switch tube four S 4 Is constant at 0.7 by adjusting D s The output voltage is regulated.
Further improvement, when the DC converter works in MG mode, D s The adjusting interval is 0-0.15.
Further improvement, when the direct-current converter works in the LG mode, no phase shift exists between switch driving signals in a first bridge arm and a second bridge arm in a staggered parallel network, and the resonant cavity inputs voltage V AB 0, diode two D 2 Start to conduct by adjusting the switching tube S 1 Three S of switch tube 3 The duty cycle D of (a) regulates the output voltage.
Compared with the prior art, the invention has the following beneficial effects:
1. the wide-output direct current converter provided by the invention adopts an interleaving structure, so that the input current ripple is obviously reduced, and the input current is divided into two transmission paths, so that the current stress of a switching tube in an interleaving parallel circuit is reduced.
2. The LLC resonant cavity is introduced into the structure of the wide-output direct-current converter, a wider zero-voltage opening range can be realized by utilizing resonant current, and the overall efficiency of the converter is higher.
3. The wide-output direct current converter provided by the invention realizes high power density through multiplexing of the coupling inductor and the switching tube, and has a simple and feasible circuit structure.
4. The wide-output direct-current converter switching control provided by the invention can enable the converter to work in three modes by switching the modulation mode, and the wide gain range is realized by adjusting the duty ratio to adjust the output voltage. And in the HG and MG working modes, all the switching tubes can realize zero-voltage switching, and the switching loss is small.
Drawings
FIG. 1 is an example of the topology of a wide output DC converter of the present invention;
FIG. 2a is a schematic diagram of a first embodiment of a wide output DC converter in HG mode;
FIG. 2b is a schematic diagram of a second main operation waveform of the wide-output DC converter in HG mode according to the embodiment of the present invention;
fig. 3 is a main operation waveform of the wide output dc converter in the MG mode according to the embodiment of the present invention;
FIG. 4 is a main operation waveform of the wide output DC converter in LG mode according to the embodiment of the present invention;
FIG. 5 is a schematic diagram illustrating three modal operating range divisions of a wide output DC converter according to an embodiment of the present invention;
FIG. 6 is a control block diagram of a wide output DC converter according to an embodiment of the invention;
wherein, the 10-coupling inductance circuit, the 20-staggered parallel circuit, the 30 resonant cavity, the 40 transformer, the 50 rectifier and the C c Clamping capacitor, C i An input bus capacitor, the coupled inductance circuit comprises N connected with a staggered parallel network p1 、N p2 A winding; and D 1 N in series p11 、N p21 A winding; n at the output end s1 、N s2 A winding; wherein L is s1 、L s2 Respectively N p1 、N p2 Leakage inductance of winding, L m1 、L m2 Exciting inductance equivalent to coupling inductance S 1 、S 2 、S 3 、S 4 Switch tube L r Resonant inductance, C r Resonant capacitor, L m Excitation inductance, L of transformer f Filter inductance C o Output capacitance, D 2 Is of the coupling inductance N s1 、N s2 Diode with serially connected windings D 3 Is connected in parallel with the coupling inductance N s1 、N s2 Winding and D 2 The diodes at two ends of the serial branch of (2) and R are loads;
g 1 ～g 4 respectively S 1 ～S 4 A driving signal of the switching tube; v (V) AB The midpoint output voltage of the staggered parallel circuit; i Lr 、I Lm Respectively, is to flow through the resonant inductance L r And excitation inductance L m Is set to be a current of (a); v (V) Ls1 、V Ls2 Respectively is leakage inductance L s1 、L s2 The voltage across the terminals; v (V) Lm1 、V Lm2 Excitation inductances L respectively being coupling inductances m1 、L m2 The voltage across the terminals; i 1 、I 2 Leakage inductance L respectively flowing through coupling inductance s1 、L s2 Is set to be a current of (a); i S1 、I S2 Respectively is flowing through a switch tube S 1 、S 2 Is set to be a current of (a); v (V) Lf For filtering inductance L f Voltage across the terminals, I D2 For flowing through diode D 2 Is of the output voltage V o 。
Detailed Description
The topological structure of the wide-output direct-current converter provided by the invention is shown in fig. 1, and comprises a 10-coupling inductance circuit, a 20-staggered parallel circuit, a 30 resonant cavity, a 40 transformer and a 50 rectifier. I i For the total input current, I 1 、I 2 Leakage inductance L respectively flowing through coupling inductance s1 、L s2 Is set to be a current of (a); v (V) AB The midpoint output voltage of the staggered parallel circuit; i Lr 、I Lm Respectively, is to flow through the resonant inductance L r And excitation inductance L m Is set to be a current of (a); i S1 、I S2 Switching tubes S respectively flowing through the first bridge arm 1 、S 2 Is set in the above-described range).
The wide-output direct current converter provided by the invention can respectively work in three working modes of HG, MG and LG by changing the modulation method: in HG working mode, the working process of the proposed wide-output DC converter is the working process of the traditional LLC resonant converter, and the resonant cavity is formed by an excitation inductance L m Resonant inductance L r Resonance capacitor C r The transformer has a transformation ratio n=n 1 :n 2 The method comprises the steps of carrying out a first treatment on the surface of the In MG operation mode, D is introduced s To adjust the output voltage gain; in LG mode, two half-bridge bridges are connected in parallel in a staggered mannerPhase shift angle of arm is zero, resonant cavity V AB The input voltage of (2) is zero and the output voltage is regulated by a variable duty cycle D, the circuit operates in the same way as a forward converter. Two coupled inductances are participated in the circuit as transformers in the forward converter topology, N P11 、N P21 The winding is a magnetic field reset winding of a transformer, N s1 、N s2 The winding provides energy output to the load end, inductance L f As an output filter inductance, flows through L f The current of (2) is I Lf Defining the turns ratio n of the primary-secondary side winding of the coupling inductor IC ＝n s1 :n p1 ＝n s2 :n p2 ，n IC2 ＝n p11 :n p1 ＝n p21 :n p2 。
In the three working modes, the staggered parallel circuit always works at a resonant frequency point, and the switching between modes can be realized by changing a switching pulse signal modulation method:
(1) In HG mode, a fixed switching frequency Pulse Width Modulation (PWM) control method is adopted, and the switching frequency f is under the rated condition s Equal to LLC resonant frequency f r . The switches on the same bridge arm operate complementarily. S is S 1 And S is 3 The switching pulse duty cycle of (a) is the same, but there is a 180 ° phase shift angle (β=180°), S 2 And S is 4 The same applies.
(2) In the MG mode, the control amount D is introduced s The dead time between the switching tubes on the same bridge arm is increased, at D>0.5, a lower voltage gain is achieved, at which time S 1 And S is 3 With the same duty cycle, but with a 180 ° phase shift angle (β=180°), S 2 And S is 4 As well as the same.
(3) In LG mode, switch tube S 1 And S is 3 Simultaneously on or off (β=0°), S 2 And S is 4 The same applies.
In HG and MG modes of operation, the input voltage V to the cavity AB For amplitude of V Cc (equal to V i Alternating rectangular wave of/D), duty cycle equal to D and minimum value N of 1-D s1 And N s2 The total voltage between them is equal to 0. Due to the coupled inductor L 1 、L 2 Is the same in value, thus I 1 、I 2 Is equal to and equal to I in /2。I 1 And I i The current ripple of (a) is denoted as Δi, respectively Ls And DeltaI i . Definition m=l m /L r ,ω r ＝2πf r ,Z r ＝L r /C r ,
Fig. 2a and 2b schematically show the main operating waveforms of the proposed wide output dc-converter operating in HG mode.
In the HG mode, one switching period can be divided into 10 working modes, and the front 5 working modes are mainly described as the front 5 working modes and the rear 5 working modes are symmetrical;
working modality 1[t 0 -t 1 ]：t 0 Before the moment S 3 Has been conducted, S 1 At t 0 The zero voltage is on at the moment. Exciting voltage V Lm Is output with voltage V o Clamping, I Lm And increases in the negative direction. Resonant cavity V AB Is zero, resonant current I Lr Negative decay occurs in a sinusoidal fashion. In addition, L s1 、L s2 The voltage across it is equal to V Cc -V i -V L Both are discharging. Due to S 1 And S is 3 Has been conducted, S 2 And S is 4 Quilt V Cc Clamping. Current I Lr 、I Lm And voltage V Cr Can be expressed as:
working modality 2[t 1 -t 2 ]: at t 1 Time of day, I Lr Equal to I Lm . In this mode, the transformer secondary winding is currentless and no energy is transferred to the secondary. L (L) r 、L m 、C r Forms resonance with a frequency f m . Due to L m The value is far greater than L r ，I Lr Is essentially unchanged in this modality. Here mouldIn the state, S 1 And S is 3 Always on, inductance L s1 、L s2 Releasing energy, S 2 And S is 4 Is subjected to capacitance voltage V Cc Clamping. Current I Lr 、I Lm And voltage V Cr Can be expressed as:
working modality 3[t 2 -t 3 ]: at t 2 Time of day, S 3 Turn off, S 1 Conducting. Inverter entry S 3 And S is 4 Is not limited to the dead zone region. The diodes of the rectifier bridge are turned off and no energy is transferred to the secondary side. S is S 3 And S is 4 And the junction capacitance of (c) begins to charge and discharge, respectively.
Working modality 4[t 3 -t 4 ]: at t 3 Time C oss4 The voltage at the two ends is reduced to zero, S 4 ZVS conduction is achieved. In this mode, the resonant cavity V AB Is V Cc ，V Lm Clamped by the output voltage. L (L) r And C r Takes part in resonance and mainly resonates with current I Lr And increases in a sinusoidal fashion. L (L) s1 And L s2 The voltages on are V respectively Cc –V i +V H And V i –V H . In addition, due to S 1 And S is 4 Conduction, L s1 And L s2 Are discharged and charged, respectively. S is S 2 、S 3 Is subjected to capacitance voltage V Cc Clamping. Voltage V of resonant capacitor Cr Current I Lr 、I Lm Can be expressed as:
working modality 5[t 4 -t 5 ]: at t 4 Time of day, S 4 Turn off, S 1 And continuing to conduct. The converter reenters S 3 And S is 4 Dead time between. In this mode, S 3 And S is 4 Is connected with electricityThe capacity is respectively through I Lr And I 2 Discharging and charging. When S is 3 Is reduced to zero, S 3 The body diode of (1) starts to conduct. Thus S 3 Zero voltage turn-on is achieved.
The working mode when D is less than or equal to 0.5 is symmetrical with the description.
Fig. 3 illustrates main operation waveforms of the proposed wide output dc converter operating in MG mode:
in the MG mode, one switching period can be divided into 8 working modes, and the front 4 working modes are mainly described as the front 4 modes and the rear 4 modes are symmetrical;
working modality 1[t 0 -t 1 ]: before this mode, S 3 Has been conducted, S 2 Shut off, L s1 ，L s2 The voltage at is equal to V Cc -V i -V L The electric energy is released. t is t 0 The latter mode is similar to the HG mode of operation mode 2, t 0 Time of day, I Lr And I Lm Equal, L r 、L m 、C r Forms resonance with a frequency f m . In this mode, the transformer does not transfer energy to the secondary side. Current I Lr 、I Lm And voltage V Cr Can be expressed as:
working modality 2[t 1 -t 2 ]: in this mode, S 3 And S is 4 All turn off and the converter enters dead time consistent with HG mode.
Working modality 3[t 2 -t 3 ]: at t 2 Time of day, S 4 ZVS on is achieved. The mode is similar to mode 4 of HG mode, where L s1 And L s2 The voltages on are V respectively Cc –V i +V 1 And V i –V 1 . Due to S 1 And S is 4 Conduction, L s1 And L s2 And discharging and charging respectively. Current I Lr 、I Lm And voltage V Cr Can be expressed as:
working modality 4[t 3 -t 4 ]: at t 3 Time of day, S 4 Turn off, S 1 Still conducting. This modality is the main difference between HG and LG modes. In this mode, V AB Zero, L s1 、L s2 The voltage at both ends is equal to V Cc –V i –V L Both release energy, I Lr And decreases linearly. Current I Lr 、I Lm And voltage V Cr Can be expressed as:
t 4 after that, the circuit enters the lower half of the operating state, the operating principle being similar to the first half cycle.
Fig. 4 illustrates the main operation waveforms of the proposed wide output dc converter operating in LG mode:
in LG mode, S 1 And S is 3 Simultaneously on and off (β=0°), S 2 And S is 4 Likewise, the same is true. Resonant cavity input voltage V AB At zero, the output voltage has a varying duty cycle dcontrol, at which time the circuit operates in a similar manner to a forward converter. In LG mode, one switching period can be divided into 6 working modes, and since the front and rear 3 modes are symmetrical, the front 3 working modes are mainly described:
working modality 1[t 0 -t 1 ]：t 0 Before the moment S 1 ，S 3 Has been turned on. Within this dead time, S 1 And S is 3 The junction capacitance of (1) starts to charge S 2 And S is 4 The junction capacitance of (a) starts to discharge. L (L) s1 And L m1 The voltages on are V respectively L2 、V i –V Cc –V L2 . By diode D 1 Winding N p11 、N p21 Magnetic reset circuitAccess, energy thereof to input capacitor C 1 Charging, effectively prevent winding N p1 、N p2 A switching tube of the interleaved parallel circuit is broken down by an excessively high reverse peak voltage. Winding N S1 And N S2 The sum of the voltages of (D) is negative 3 Conduction, D 2 And (5) disconnecting. L (L) f The voltage across it is-V o 。
Working modality 2[t 1 -t 2 ]: at t 1 Time of day, S 2 ，S 4 Conducting. In this mode, diode D 2 、D 3 Still conducting, secondary winding N of coupling inductance S1 、N S2 Is clamped to zero and the current flowing through them increases. L (L) f 、L m1 、L s1 The voltages at both ends are respectively-V o 0 and V i 。
Working modality 3[t 2 -t 3 ]: at the beginning of the mode, diode D 3 Shut off, flow through D 2 The current of (2) is I Lf . By coupling inductance L 1 And L 2 Power is delivered to the load. L (L) f 、L s1 The voltages at the two ends are respectively 2n IC (V i –V H2 )–V o And V H2 。
Working modality 4[t 3 -t 4 ]: this interval is dead time, S 1 And S is 3 The junction capacitance of (1) starts to discharge, S 2 And S is 4 And the junction capacitance of (c) begins to charge.
Working modality 5[t 4 -t 5 ]: at the beginning of the mode, S 1 、S 3 On, diode D 3 Continuing to conduct, winding N S1 、N S2 The voltage across it is clamped to zero and the current flowing through it decreases. L (L) f 、L m1 、L s1 The voltages at both ends are respectively-V o 0 and V i –V Cc 。
Working modality 6[t 5 -t 6 ]: at t 2 At the moment, flow through D 2 Is equal to zero, diode D 2 And closing. L (L) f 、L m1 、L s1 The voltages at both ends are respectively-V o 、V i –V Cc –V L2 And V L2 . Diode D 1 The magnetic reset circuit is connected to work again.
Fig. 5 illustrates a schematic diagram of the division of the operating region of the proposed wide output dc converter:
the wide output DC converter provided by the invention adopts a fixed switching frequency pulse width modulation technology to regulate output voltage, and the resonant circuit and the forward circuit are connected in parallel at a load end according to the analysis of the working mode. The voltage gains of the three modes are independent. In HG mode the converter may be seen as a combination of a Boost converter and a full-bridge LLC converter, the voltage gains of which do not affect each other. In the MG mode, the duty ratio D is fixed to 0.7, and the control variable is D s . Similarly, boost and LLC converters may be analyzed in MG mode, respectively. While in LG mode the circuit operating condition is similar to a forward circuit.
Since in HG, MG modes the converter operating conditions are analogous to the combination of a Boost converter and a full-bridge LLC converter, the gain expression is:
wherein M is LLC Gain, M, of LLC resonant converter BOOST To couple the inductance with the output voltage gain of the Boost circuit of the interleaved parallel circuit, V o To output voltage V i The input voltage, n is the transformer transformation ratio, and D is the duty cycle of the interleaved parallel circuit. As D increases, the converter gain will decrease.
In MG and LG modes, the gain of the inverter can be expressed as:
wherein, intermediate expressions X and Y:f s for switching frequency of staggered parallel circuit, L s1 Is N s1 Leakage inductance of winding, n IC For coupling the turns ratio of the primary winding of the inductor, L f For outputting the filter inductance value, R is the load resistance value.
In the MG mode, the duty ratio cannot be too small to ensure normal reset of the magnetic core, and meanwhile when the output power in the forward circuit is larger, the switching loss is larger and the gain is reduced, and the working area is divided into three modes according to the following two conditions by combining the efficiency and gain characteristics of the HG mode, the MG mode and the LG mode:
(1) Boundary conditions of HG and MG modes: d=0.7, D s ＝0。
(2) Boundary conditions of MG and LG modes: β=0, d=0.57, β represents the phase shift angle between the first leg and the second leg.
According to the analysis, d=0.7, q=0.4 boundary voltage gain M of HG mode HGB About 0.814, LG mode boundary voltage gain M LGB By substituting d=0.57 and the load condition into M LG The expression is obtained.
Fig. 6 illustrates a control block diagram of the proposed wide output dc-converter:
the proposed wide output DC converter uses voltage single closed loop control. Input voltage V is input through sensor i Output voltage V o Set output voltage reference value V o * Into a controller. First, input voltage V i And output voltage reference V o * The operation mode of the converter is determined by sending the operation mode to a mode selection controller in combination with the operation area division diagram in fig. 5. At the same time, the sampled output voltage V o With output voltage reference V o * Comparing, the difference is transmitted to a proportional-integral (PI) controller, and the output signal V con Sending to a duty ratio adjusting controller corresponding to the working mode to adjust D or D s . Different duty cycle calculation programs, i.e. different modulation, for different modes of operationAnd (5) mode control. When V is i And V o * Upon change, the circuit will switch smoothly between HG, MG and LG modes.
Claims (10)
1. The wide-output direct-current converter with the multi-mode switching function is characterized by comprising a coupling inductance circuit (10), wherein the coupling inductance circuit (10) is electrically connected with a staggered parallel circuit (20) and a resonant cavity (30), and the resonant cavity (30) is sequentially electrically connected with a transformer (40), a rectifier (50) and an output capacitor; the coupling inductance circuit (10) is electrically connected with the rectifier (50); the coupled inductor circuit (10) comprises a winding one (N) p1 ) And winding two (N) p2 ) Winding one (N) p1 ) And winding two (N) p2 ) Is electrically connected in parallel with a first diode (D 1 ) First diode (D 1 ) Electrically connected with windings three (N) arranged in parallel p1 ) Sum winding four (N) p2 ) The method comprises the steps of carrying out a first treatment on the surface of the Winding three (N) p1 ) Comprises an excitation inductance I (L) equivalent to a coupling inductance m1 ) And with a first diode (D 1 ) Connected leakage inductance one (L) s1 ) The method comprises the steps of carrying out a first treatment on the surface of the The winding four (N) p2 ) Including coupling inductance equivalent excitation inductance two (L) m2 ) And with a first diode (D 1 ) Connected leakage inductance II (L) s2 ) The method comprises the steps of carrying out a first treatment on the surface of the Winding three (N) p1 ) Sum winding four (N) p2 ) Is electrically connected with the staggered parallel circuit (20); the staggered parallel circuit (20) comprises a first bridge arm and a second bridge arm which are of two half-bridge structures, and the staggered parallel circuit (20) is connected with a clamping capacitor (C) in parallel c ) The method comprises the steps of carrying out a first treatment on the surface of the The resonant cavity (30) comprises resonant inductors (L) which are sequentially connected in series r ) Exciting inductance (L) m ) Resonance capacitor (C) r )。
2. The multimode switched wide output dc converter of claim 1, wherein the coupled inductor circuit (10) further comprises a coupled inductor one (N) electrically connected to a rectifier (50) s1 ) And coupled inductor two (N) s2 ) Coupling inductance one (N) s1 ) And coupled inductor two (N) s2 ) Are connected in series, and the inductance value and the transformation ratio of the windings are the same.
3. The multimode switch of claim 2Is characterized in that the coupling inductance one (N) s1 ) And coupled inductor two (N) s2 ) After the heterogeneous terminals are connected, the inductor one (N) s1 ) Diode two (D) 2 ) And output inductance (L) f ) The method comprises the steps of carrying out a first treatment on the surface of the Diode II (D) 2 ) Electrically connected diode tris (D 3 ) Cathode of diode three (D 3 ) Is electrically connected with the anode of the coupling inductor II (N) s2 ) And an output negative terminal.
4. The multi-mode switched wide output dc converter of claim 3 wherein the first leg comprises a first switching tube (S1) and a third switching tube (S2) in series; the second bridge arm comprises a switching tube II (S2) and a switching tube IV (S4) which are connected in series.
5. A method of using the multimode switched wide output dc converter of claim 4, comprising the steps of:
s1, collecting input voltage V of a direct current converter i Output voltage V o ；
S2, input voltage V to be collected i With a given output voltage command value V o * The controller fed into the DC converter carries out mode judgment and confirms the adjustment control quantity:
s21, when the given output voltage command value V o * Greater than or equal to input voltage V i Boundary value M of gain with HG mode HGB The product of (3) and the converter works in HG mode;
s22, when the given output voltage command value V o * Less than the input voltage V i With HG modal gain M HGB And is greater than or equal to the input voltage V i Boundary value M with MG mode gain MGB The product of (2), the converter works in MG mode;
s23, when the given output voltage command value V o * Less than the input voltage V i Boundary value M with MG mode gain MGB The transformer works in LG mode;
s3, outputting the collected powerPressure V o Subtracting the given voltage command value Vo to obtain a voltage error, sending the voltage error to a proportional-integral controller, and sending the output Vcon of the proportional-integral controller to the controller for duty ratio adjustment to obtain PWM waves meeting corresponding D, ds and beta; wherein D represents a duty ratio, ds represents a control amount introduced in the MG mode, and β represents a phase shift angle between the first arm and the second arm;
s4, a driving circuit for sending PWM waves to a driving circuit of the DC converter to obtain a first control switch tube (S 1 ) Switch tube II (S) 2 ) Three of switch tube (S) 3 ) And switch tube IV (S) 4 ) Is a driving signal g of (2) 1 、g 2 、g 3 、g 4 。
6. The method of claim 5, wherein when the dc converter is operated in HG mode, the phase shift β between the switching driving signals in the first leg and the second leg in the interleaved parallel network is 180 °, which is complementary to the switching driving signals in the legs, and the output voltage is adjusted by adjusting the duty ratio D of the switching tubes in the first leg and the second leg.
7. The method of claim 6, wherein the duty cycle D adjustment range is 0.36-0.7 when the dc converter is operating in HG mode.
8. The method of using a multimode switched wide output DC converter as in claim 5, wherein when the DC converter is operating in MG mode, a phase shift beta between switch drive signals in a first bridge arm and a second bridge arm in an interleaved parallel network is 180 DEG, and a control variable D is introduced s ，D s Is a switching tube II (S) 2 ) Is a driving signal g of (2) 2 With the first switch tube (S) 1 ) A phase difference of the driving signal g 1; switch tube III (S) 3 ) And switch tube IV (S) 4 ) Is constant at 0.7 by adjusting D s The output voltage is regulated.
9. The method of claim 8, wherein D is D when the dc converter is operating in MG mode s The adjusting interval is 0-0.15.
10. The method of claim 5, wherein when the DC converter is operated in LG mode, there is no phase shift between the switching drive signals in the first bridge arm and the second bridge arm in the interleaved parallel network, and the resonant cavity input voltage V AB 0, diode two (D 2 ) Starts to conduct by adjusting the first switch tube (S 1 ) Three of switch tube (S) 3 ) The duty cycle D of (a) regulates the output voltage.
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