CN115864860B

CN115864860B - LLC resonance type DC converter system with wide gain and application method

Info

Publication number: CN115864860B
Application number: CN202310197239.1A
Authority: CN
Inventors: 周德洪; 王龙广; 沈泽微; 邹见效
Original assignee: Higher Research Institute Of University Of Electronic Science And Technology Shenzhen
Current assignee: Higher Research Institute Of University Of Electronic Science And Technology Shenzhen
Priority date: 2023-03-03
Filing date: 2023-03-03
Publication date: 2023-05-26
Anticipated expiration: 2043-03-03
Also published as: CN115864860A

Abstract

The invention relates to the technical field of direct current converters, in particular to an LLC resonance type DC converter system with wide gain, which comprises the following steps: the system comprises a DC converter, a mode control module, a PI control module and a strategy modulation module, wherein the DC converter is a double half-bridge LLC resonant DC converter; the input end of the primary side inversion module of the DC converter is connected with the input end of the modal control module, the output end of the secondary side voltage doubling rectifier bridge of the DC converter is connected with the input end of the PI control module, and the output end of the primary side inversion module is connected with the input end of the secondary side voltage doubling rectifier bridge; the output end of the mode control module is connected with the input end of the PI control module, the output end of the PI control module is connected with the input end of the strategy modulation module, and the output end of the strategy modulation module is connected with the DC converter. According to the system, the gain range is widened through more circuit modes, the wide output gain range of 0-2 times is realized, and the influence of load conditions on the output gain is effectively reduced.

Description

LLC resonance type DC converter system with wide gain and application method

Technical Field

The invention relates to the technical field of direct current converters, in particular to an LLC resonance type DC converter system with wide gain and an application method.

Background

For a conventional half-bridge, full-bridge LLC resonant converter, the resonant cavity gain is related to the ratio of the resonant impedance and the excitation impedance. At a fixed resonant frequency, the resonant cavity gain changes as the operating frequency changes, and frequency modulation (Pulse Frequency Modulation, PFM) can be used to change the output gain value. In the process of using frequency modulation, the traditional LLC resonant converter can meet zero-voltage and current switching conditions only when working in an inductive working area, so that the frequency conversion range is limited, and the voltage gain below 1 time is difficult to realize by improving the working frequency. In addition, since the gain of the resonant cavity is affected by the load condition, the gain range in the inductive region changes when the load condition changes in the inductive operating region. Therefore, the conventional LLC resonant converter has the following problems: in the conventional PFM control mode, the output voltage of the conventional LLC resonant converter is typically limited by the operating frequency, but can be regulated only in a limited frequency range. In the application scenario of a wide voltage output range, the influence of the load on the output gain needs to be considered.

Disclosure of Invention

According to the LLC resonance type DC converter system with the wide gain and the application method, the technical problems that in the prior art, under a traditional PFM control mode, output voltage of a traditional LLC resonance converter is adjusted in a limited variable frequency range, if the output voltage is in a wide voltage output range, the output voltage is influenced by loads, the gain range is widened by constructing more circuit modes, the wide output gain range of 0-2 times is achieved, and the influence of load conditions on the output gain is effectively reduced are solved.

In a first aspect, an embodiment of the present invention provides an LLC resonant DC converter system with a wide gain, including:

the system comprises a DC converter, a mode control module, a PI control module and a strategy modulation module, wherein the DC converter is a double half-bridge LLC resonant DC converter;

the input end of the primary side inversion module of the DC converter is connected with the input end of the modal control module, the output end of the secondary side voltage doubling rectifier bridge of the DC converter is connected with the input end of the PI control module, and the output end of the primary side inversion module is connected with the input end of the secondary side voltage doubling rectifier bridge;

the output end of the mode control module is connected with the input end of the PI control module, the output end of the PI control module is connected with the input end of the strategy modulation module, and the output end of the strategy modulation module is connected with the DC converter.

Preferably, the DC converter includes: the direct current input power supply, the bus input capacitor, the primary side inversion module, the secondary side voltage doubling rectifier bridge, the output capacitor group and the resistive load;

the positive electrode of the bus input capacitor and the positive electrode of the primary side inversion module are connected with the positive electrode of the direct current input power supply, and the negative electrode of the bus input capacitor and the negative electrode of the primary side inversion module are connected with the negative electrode of the direct current input power supply; the secondary side voltage doubling rectifier bridge, the output capacitor group and the resistive load are connected in parallel.

Preferably, the primary side inversion module includes: the first resonant bridge arm, the second resonant bridge arm, the first resonant cavity, the second resonant cavity, the primary side of the first transformer and the primary side of the second transformer;

the first resonant bridge arm comprises a first switching tube and a second switching tube, the drain electrode of the first switching tube is connected with the positive electrode of the direct current input power supply, the source electrode of the first switching tube is connected with the drain electrode of the second switching tube, and the source electrode of the second switching tube is connected with the negative electrode of the direct current input power supply;

the second resonance bridge arm comprises a third switching tube and a fourth switching tube, the drain electrode of the third switching tube is connected with the positive electrode of the direct current input power supply, the source electrode of the third switching tube is connected with the drain electrode of the fourth switching tube, and the source electrode of the fourth switching tube is connected with the negative electrode of the direct current input power supply;

the first resonant cavity comprises a first resonant inductor, a first excitation inductor and a first resonant capacitor, one end of the first resonant inductor is connected with the source electrode of the first switching tube, the other end of the first resonant inductor, the first excitation inductor and one end of the first resonant capacitor are connected in series, and the other end of the first resonant capacitor is connected with the negative electrode of the direct-current input power supply;

The second resonant cavity comprises a second resonant inductor, a second excitation inductor and a second resonant capacitor, one end of the second resonant inductor is connected with the source electrode of the third switching tube, the other end of the second resonant inductor, the second excitation inductor and one end of the second resonant capacitor are connected in series, and the other end of the second resonant capacitor is connected with the negative electrode of the direct-current input power supply;

the primary side of the first transformer is connected with the first excitation inductor in parallel;

the primary side of the second transformer is connected in parallel with the second excitation inductance.

Preferably, the secondary side voltage doubling rectifier bridge includes: a second side of the first transformer, a second side of the second transformer, a first diode, a second diode, a third diode and a fifth switching tube;

the second side synonym end of the first transformer is connected with the second side synonym end of the second transformer, the second side synonym end of the first transformer, the positive electrode of the first diode and the negative electrode of the second diode are connected, the second side synonym end of the second transformer, the negative electrode of the third diode and the source electrode of the fifth switch tube are connected, the negative electrode of the first diode is a first port of the secondary side voltage doubling rectifier bridge, the drain electrode of the fifth switch tube is a second port of the secondary side voltage doubling rectifier bridge, and the intersection point of the positive electrode of the second diode and the positive electrode of the third diode is a third port of the secondary side voltage doubling rectifier bridge.

Preferably, the output capacitor group includes: a first output capacitor and a second output capacitor;

the positive pole of first output electric capacity with the first port of secondary side voltage-multiplying rectifier bridge links to each other, the negative pole of first output electric capacity, the positive pole of second output electric capacity with the second port of secondary side voltage-multiplying rectifier bridge links to each other, the negative pole of second output electric capacity with the third port of secondary side voltage-multiplying rectifier bridge links to each other.

Preferably, the PI control module includes: an output voltage PI control module and an output current PI control module; the system also comprises an output current sampling module and an output voltage sampling module;

the input end of the output voltage PI control module is respectively connected with the output end of the output voltage sampling module and the output end of the modal control module, and the input end of the output current PI control module is respectively connected with the output end of the output current sampling module and the output end of the modal control module; the input end of the output current sampling module is connected between the positive electrode of the output capacitor group and the positive electrode of the resistive load, and the input end of the output voltage adoption module is connected with the resistive load in parallel.

Based on the same inventive concept, the present invention also provides, in a second aspect, a method of applying an LLC resonant DC converter system with a wide gain, to an LLC resonant DC converter system with a wide gain as described above, the method including:

after the resonance parameters and the modal interval of the DC converter of the system are determined, the input voltage, the output voltage and the output current of the DC converter are obtained;

determining a current mode of the DC converter according to the input voltage of the DC converter through a mode control module of the system;

obtaining modulation parameters according to the current mode, output voltage and output current of the DC converter through a PI control module of the system;

and controlling the on-off state of each switching tube of the DC converter according to the current mode of the DC converter and the modulation parameters through a strategy modulation module of the system.

Preferably, the determining the resonance parameter of the DC converter of the system includes:

and analyzing the current characteristics of the DC converter by a first harmonic approximation method to obtain the target frequency range and LLC resonance parameters of the DC converter.

Preferably, the determining the current mode of the DC converter according to the input voltage of the DC converter includes:

If the input voltage of the DC converter is in a first mode voltage threshold range, determining a first gain mode as the current mode, wherein the modulation mode of the first gain mode is phase-shift modulation;

and if the input voltage of the DC converter is in the second mode voltage threshold range, determining a compensation gain mode as the current mode, wherein the modulation mode of the compensation gain mode is frequency modulation.

and if the input voltage of the DC converter is in the third mode voltage threshold range, determining a second gain mode as the current mode, wherein the modulation mode of the second gain mode is pulse width modulation.

One or more technical solutions in the embodiments of the present invention at least have the following technical effects or advantages:

in an embodiment of the present invention, an LLC resonant type DC converter system with a wide gain includes: the system comprises a DC converter, a mode control module, a PI control module and a strategy modulation module, wherein the DC converter is a double half-bridge LLC resonant DC converter. Here, the DC converter of this embodiment includes a double half-bridge LLC resonant structure, and more circuit modes can be constructed through the double half-bridge LLC resonant structure to widen the output gain range of the DC converter, and achieve a wide output gain range of 0 to 2 times. When the independent resonant cavity structure is used for phase-shifting modulation, the problem of low load efficiency caused by overlarge phase-shifting angle can be effectively avoided.

The input end of the primary side inversion module of the DC converter is connected with the input end of the modal control module, the output end of the secondary side voltage doubling rectifier bridge of the DC converter is connected with the input end of the PI control module, and the output end of the primary side inversion module is connected with the input end of the secondary side voltage doubling rectifier bridge. The output end of the mode control module is connected with the input end of the PI control module, the output end of the PI control module is connected with the input end of the strategy modulation module, and the output end of the strategy modulation module is connected with the DC converter.

The mode control module is used for determining the current mode of the DC converter from preset modes of the DC converter according to the input voltage of the DC converter. The PI control module is used for obtaining modulation parameters according to the current mode and the output voltage and the output current of the DC converter, wherein the modulation parameters comprise modulation voltage and modulation current. And the strategy modulation module is used for controlling the on-off state of a switching tube of the DC converter according to the modulation parameters.

Therefore, the system can modulate the DC converter in different modes, widen the gain range of the DC converter, control the DC converter to output stable voltage, reduce the influence of load, improve the working efficiency of the DC converter and the system, and ensure the stable operation of the DC converter and the system.

Drawings

Various other advantages and benefits will become apparent to those of ordinary skill in the art upon reading the following detailed description of the preferred embodiments. The drawings are only for purposes of illustrating the preferred embodiments and are not to be construed as limiting the invention. Also throughout the drawings, like reference numerals are used to designate like parts. In the drawings:

fig. 1 shows a schematic configuration of an LLC resonant DC converter system with wide gain in accordance with an embodiment of the invention;

fig. 2 shows a circuit configuration diagram of a DC converter in an embodiment of the present invention;

fig. 3 is a schematic diagram showing a modulation mode corresponding to a mode of a DC converter in an embodiment of the present invention;

fig. 4 is a schematic diagram illustrating an operation principle between a mode control module and a PI control module of an LLC resonant DC converter system with a wide gain in accordance with an embodiment of the invention;

fig. 5a is a schematic diagram illustrating an operation principle of a PI control module in a low gain mode according to an embodiment of the present invention;

fig. 5b is a schematic diagram illustrating an operation principle of the PI control module in the compensation gain mode according to an embodiment of the present invention;

fig. 5c is a schematic diagram illustrating an operation principle of the PI control module in the high gain mode according to the embodiment of the present invention;

FIG. 6 is a schematic diagram illustrating the operation principle between strategy modulation modules of an LLC resonant DC converter system with wide gain in an embodiment of the invention;

fig. 7a is a schematic diagram illustrating an operation principle of a policy modulation module in a low gain mode according to an embodiment of the present invention;

fig. 7b is a schematic diagram illustrating an operation principle of the policy modulation module in the compensation gain mode according to an embodiment of the present invention;

fig. 7c is a schematic diagram illustrating an operation principle of the policy modulation module in the high gain mode according to the embodiment of the present invention;

fig. 8 is a schematic diagram illustrating an operation principle of a first carrier comparator in an embodiment of the present invention;

fig. 9 is a schematic diagram illustrating the working principle of the second carrier comparator in the embodiment of the present invention;

FIG. 10 shows waveforms obtained by modulating a DC converter with phase-shift modulation by the present system in an embodiment of the present invention;

FIG. 11 shows waveforms obtained by modulating a DC converter with frequency modulation by the present system in an embodiment of the present invention;

FIG. 12 shows waveforms obtained by modulating a DC converter with pulse width modulation by the present system in an embodiment of the present invention;

FIG. 13 shows a graph of the gain achieved with PFM modulation of a conventional LLC resonant converter in an embodiment of the present invention;

FIG. 14 shows a gain plot of the present DC converter implemented by the present system modulation in an embodiment of the present invention;

fig. 15 shows a schematic flow chart of steps of an application method of an LLC resonant DC converter system with wide gain in accordance with an embodiment of the invention.

Detailed Description

Exemplary embodiments of the present disclosure will be described in more detail below with reference to the accompanying drawings. While exemplary embodiments of the present disclosure are shown in the drawings, it should be understood that the present disclosure may be embodied in various forms and should not be limited to the embodiments set forth herein. Rather, these embodiments are provided so that this disclosure will be thorough and complete, and will fully convey the scope of the disclosure to those skilled in the art.

Embodiment one: a first embodiment of the present invention provides an LLC resonant type DC converter system with a wide gain, as shown in fig. 1, including: the system comprises a DC converter 100, a mode control module 200, a PI control module 300 and a strategy modulation module 400, wherein the DC converter 100 is a double half-bridge LLC resonant DC converter.

The input end of the primary side inversion module of the DC converter 100 is connected with the input end of the modal control module 200, the output end of the secondary side voltage doubling rectifier bridge of the DC converter 100 is connected with the input end of the PI control module 300, and the output end of the primary side inversion module is connected with the input end of the secondary side voltage doubling rectifier bridge.

The output end of the mode control module 200 is connected with the input end of the PI control module 300, the output end of the PI control module 300 is connected with the input end of the strategy modulation module 400, and the output end of the strategy modulation module 400 is connected with the DC converter 100.

The mode control module 200 is configured to determine a current mode of the DC converter 100 from preset modes of the DC converter 100 according to an input voltage of the DC converter 100;

the PI control module 300 is configured to obtain a modulation parameter according to the current mode, the output voltage and the output current of the DC converter 100, where the modulation parameter includes a modulation voltage and a modulation current;

the strategy modulation module 400 is used for controlling the on-off state of the switching tube of the DC converter 100 according to the current mode and the modulation parameter.

Next, a specific structure of the LLC resonant DC converter system with wide gain according to the present embodiment will be described in detail with reference to fig. 1:

the DC converter 100 of the present embodiment includes: the power supply comprises a direct current input power supply 110, a bus input capacitor 120, a primary side inverter module 130, a secondary side voltage-doubling rectifier bridge 140, an output capacitor set 150 and a resistive load 160.

The positive electrode of the bus input capacitor 120 and the positive electrode of the primary side inversion module 130 are both connected with the positive electrode of the direct current input power supply 110, and the negative electrode of the bus input capacitor 120 and the negative electrode of the primary side inversion module 130 are both connected with the negative electrode of the direct current input power supply 110; the secondary side voltage doubler rectifier bridge 140, the output capacitor bank 150, and the resistive load 160 are connected in parallel. The DC input power 110 provides DC power to the system and the DC converter 100, and the DC bus capacitor is used to stabilize the input voltage of the DC input power 110.

Specifically, as shown in fig. 2, the primary-side inverter module 130 includes: a first resonant bridge arm, a second resonant bridge arm, a first resonant cavity, a second resonant cavity and a first transformer T ₁ Primary side of (d) and second transformer T ₂ Is a primary side of (a).

The first resonant bridge arm comprises a first switch tube S ₁ And a second switching tube S ₂ First switch tube S ₁ A first switch tube S connected with the positive electrode of the DC input power supply 110 ₁ Source electrode of (a) and second switch tube S ₂ Is connected with the drain electrode of the second switch tube S ₂ Is connected to the negative pole of the dc input power source 110.

The second resonant bridge arm comprises a third switch tube S ₃ And a fourth switching tube S ₄ Third switch tube S ₃ A third switch tube S connected with the positive electrode of the DC input power supply 110 ₃ Source electrode and fourth switch tube S ₄ Is connected with the drain electrode of the fourth switching tube S ₄ Is connected to the negative pole of the dc input power source 110.

The first resonant cavity comprises a first resonant inductor L _r1 First excitation inductance L _m1 And a first resonance capacitor C _r1 First resonant inductance L _r1 One end of (a) is connected with the first switch tube S ₁ Is connected with the source electrode of the first resonant inductor L _r1 Is the other end of the first excitation inductance L _m1 And a first resonance capacitor C _r1 Is connected in series with one end of a first resonance capacitor C _r1 And the other end of the capacitor is connected to the negative electrode of the dc input power source 110.

The second resonant cavity comprises a second resonant inductance L _r2 Second excitation inductance L _m2 And a second resonance capacitor C _r2， Second resonant inductance L _r2 One end of (a) is connected with a third switch tube S ₃ A second resonant inductor L connected to the source electrode _r2 And the other end of the second excitation inductance L _m2 And a second resonance capacitor C _r2 A second resonance capacitor C connected in series with one end of _r2 And the other end of the power supply is connected with a direct current input power supply110 are connected.

First transformer T ₁ Primary side of (1) and first excitation inductance L _m1 Connected in parallel, a second transformer T ₂ Primary side of (2) and second excitation inductance L _m2 Connected in parallel.

The secondary side voltage doubler rectifier bridge 140 includes: first transformer T ₁ Secondary side of (2), second transformer T ₂ Secondary side of (a), first diode D ₁ Second diode D ₂ Third diode D ₃ And a fifth switching tube S _5。

First transformer T ₁ Is a second transformer T ₂ Is connected with the same name end of the secondary side of the first transformer T ₁ Second-side homonymous terminal of first diode D ₁ Positive electrode of (D) and second diode D ₂ Is connected with the negative pole of the second transformer T ₂ Second-side synonym terminal of (D), third diode D ₃ Is connected with the negative electrode of the fifth switch tube S ₅ Is connected with the source of the first diode D ₁ The negative pole of (a) is the first port of the secondary side voltage-doubling rectifier bridge 140, and the fifth switch tube S ₅ The drain of (a) is the second port of the secondary side voltage-multiplying rectifier bridge 140, the second diode D ₂ Positive electrode of (D) and third diode D ₃ The intersection of the positive poles of (2) is the third port of the secondary side voltage doubler rectifier bridge 140.

The output capacitor bank 150 includes: first output capacitor C _o1 And a second output capacitor C _o2 . First output capacitor C _o1 The positive pole of the (B) is connected with the first port of the secondary side voltage-doubling rectifier bridge 140, namely a first output capacitor C _o1 Positive electrode of (D) and first diode D ₁ Is connected to the negative electrode of the battery. First output capacitor C _o1 Negative electrode of (C) and second output capacitor (C) _o2 The positive pole of the (B) is connected with the second port of the secondary side voltage-doubling rectifier bridge 140, namely a first output capacitor C _o1 Negative electrode of (C) and second output capacitor (C) _o2 Positive electrode of (a) and fifth switching tube S ₅ Is connected to the drain of the transistor. Second output capacitor C _o2 Is connected to the third port of the secondary-side voltage-doubler rectifier bridge 140, i.e., the second diode D ₂ Positive electrode of (D), third diode D ₃ Positive electrode of (C) and second output capacitance C _o2 Is connected to the negative electrode of the battery.

Since the secondary side voltage-doubling rectifier bridge 140, the output capacitor set 150 and the resistive load 160 are connected in parallel, one end of the resistive load 160 is connected with the first output capacitor C _o1 Is connected with the positive electrode of the first diode D ₁ Is connected to the negative electrode of the resistive load 160 and the second diode D ₂ Positive electrode of (D), third diode D ₃ Positive electrode of (C) and second output capacitance C _o2 Is connected to the negative electrode of the battery.

Note that, the first switching tube S of the present embodiment ₁ Second switch tube S ₂ Third switch tube S ₃ Fourth switching tube S ₄ And a fifth switching tube S ₅ Are all MOS transistors. As shown in fig. 2, the switching transistors are all MOS transistors with a switching transistor junction capacitance and a diode.

First switching tube S ₁ To a fifth switching tube S ₅ Is related to the voltage level of the DC input power source 110, and a first switch S ₁ To a fifth switching tube S ₅ A margin should be maintained. Fifth switch tube S ₅ Is related to the resonant voltage of the resonant cavity of the DC converter 100 and should be kept with margin. First switching tube S ₁ To a fifth switching tube S ₅ The withstand voltage value of each switching tube should be 1.5 times or more the output voltage value of the dc input power supply 110. In order to ensure a margin for each switching tube, the present embodiment selects a power MOSFET having a withstand voltage 2 times that of the output voltage of the dc input power supply 110.

Fifth switch tube S ₅ Is selected in relation to the output voltage of the secondary side voltage-doubling rectifying bridge 140 of the DC converter 100, and the induced voltage of the secondary side of the DC converter 100 is supplied to the first output capacitor C through the secondary side voltage-doubling rectifying bridge 140 _o1 And a second output capacitor C _o2 And (5) charging. Thus, the fifth switching tube S ₅ The voltage stress at two ends is the second output capacitor C _o2 Half of the voltage across it. To make the fifth switch tube S ₅ The margin is maintained, and the MOSFET having the withstand voltage value of the maximum output voltage value of the DC converter is selected in this embodiment.

The principle of operation of the DC converter 100: the DC input power 110 provides DC power to the system and the DC converter 100, and as shown in fig. 1 and 2, the primary inverter module 130 includes two sets of relatively independent half-bridge LLC resonant structures, which are a first resonant bridge arm, a first resonant cavity, a second resonant bridge arm, and a second resonant cavity, respectively. The phase difference of the first resonant leg and the second resonant leg provides a phase-shift modulated mode of operation for the circuitry of the DC converter 100. When the phase difference between the first resonant bridge arm and the second resonant bridge arm is 0, the working frequency of the switching tube is changed to provide a frequency modulation working mode for the circuit. The secondary side voltage-multiplying rectifier bridge 140 comprises a set of controllable voltage-multiplying rectifiers, and when the phase difference between the first resonant bridge arm and the second resonant bridge arm of the primary side inverter module 130 is 0 and the working frequency of the switching tube is fixed and equal to the resonant frequency, the on pulse width of the switching tube in the secondary side voltage-multiplying rectifier bridge 140 is changed to provide a pulse width modulation mode for the circuit. In summary, the primary inverter module 130 and the secondary voltage-doubling rectifier bridge 140 provide three switchable modes, i.e. three degrees of freedom, for the circuit, which effectively widens the output gain range of the DC converter 100.

The DC converter 100 of this embodiment includes a double half-bridge LLC resonant structure, through which more circuit modes can be constructed to widen the output gain range of the DC converter 100, and a wide output gain range of 0-2 times is realized. When the independent resonant cavity structure is used for phase-shifting modulation, the problem of low load efficiency caused by overlarge phase-shifting angle can be effectively avoided. And, compared with the traditional full-bridge LLC resonant converter structure based on phase-shifting modulation, the traditional full-bridge LLC resonant converter based on phase-shifting modulation loses soft switching performance when the phase-shifting angle is larger. Because the dual half-bridge LLC resonant structure of the DC converter 100 is relatively independent in operation and does not affect each other, namely, the relative independence of the two sets of resonant cavities, the potential ZVS failure problem of the full-bridge LLC resonant converter based on phase-shift modulation can be effectively solved.

The voltage square wave input into the resonant cavity is in [0,U ] by the double half-bridge LLC resonant structure of the embodiment _dc ]Change between U _dc Is the voltage of the dc input power 110. After passing through the resonant cavity, the voltage stress of each resonant element is equal to that of the traditional full-bridge LLC resonant converter (in the same direct-current voltage input stripUnder the part), the DC converter 100 of the present embodiment is suitable for medium-high voltage and high power applications, such as widely applied in the fields of high power, high power density and high reliability requirements of electric automobile charging piles, vehicle-mounted chargers, electric automobiles and the like.

The mode control module 200 of the present embodiment includes a mode control module 201 and a mode selection module 202, and the system of the present embodiment further includes an input voltage sampling module 501, where the input voltage sampling module 501 is actually a voltage sensor. The input end of the input voltage sampling module 501 is connected with two ends of the direct current input power supply 110 of the DC converter 100, the input end of the mode control module 201 is connected with the output end of the input voltage sampling module 501, the output end of the mode control module 201 is connected with the input end of the module selection module, and the output end of the mode selection module 202 is connected with the input end of the PI control module 300.

The input voltage sampling module 501 is configured to collect an input voltage of the DC input power source 110 of the DC converter 100, that is, the input voltage of the DC converter 100, and then transmit the input voltage of the DC converter 100 to the mode control module 201.

The mode control module 201 is configured to receive the input voltage of the DC converter 100 transmitted by the input voltage sampling module 501, compare the input voltage of the DC converter 100 with a first preset voltage threshold and a second preset voltage threshold, and obtain an input voltage comparison result; and then, according to the input voltage comparison result, determining the current mode of the DC converter 100 from the preset modes of the DC converter 100, and transmitting the current mode of the DC converter 100 to the mode selection module 202. The first preset voltage threshold and the second preset voltage threshold are set according to actual requirements, and the first preset voltage threshold is smaller than the second preset voltage threshold.

The mode selection module 202 is configured to output a control signal corresponding to the current mode of the DC converter 100 to the PI control module 300 and the strategy modulation module 400 according to the received current mode of the DC converter 100, so that the PI control module 300 modulates the DC converter 100 according to the corresponding control signal and the output voltage and output current of the DC converter 100, and the strategy modulation module 400 also modulates the DC converter according to the corresponding control signal.

The PI control module 300 of the present embodiment includes: an output voltage PI control module 301 and an output current PI control module 302. The PI control modules in this embodiment are PI controllers, i.e., proportional-integral controllers. The system of this embodiment further includes an output current sampling module 502 and an output voltage sampling module 503, where the output current sampling module 502 is actually a current sensor, and the output voltage sampling module 503 is actually a voltage sensor.

The input end of the output voltage PI control module 301 is connected to the output end of the output voltage sampling module 503 and the output end of the mode control module 200 (i.e., the output end of the mode selection module 202), and the input end of the output current PI control module 302 is connected to the output end of the output current sampling module 502 and the output end of the mode control module 200 (i.e., the output end of the mode selection module 202). An input of the output current sampling module 502 is connected between the positive pole of the output capacitor bank 150 and the positive pole of the resistive load 160, and an input of the output voltage application module is connected in parallel with the resistive load 160.

The output voltage sampling module 503 is configured to collect the voltage of the resistive load 160, that is, the output voltage of the secondary side voltage-doubler rectifier bridge 140, which is also the output voltage of the DC converter 100, and then output the output voltage of the DC converter 100 to the output voltage PI control module 301.

The output current sampling module 502 is configured to collect the current of the resistive load 160, that is, the output current of the secondary side voltage-doubler rectifier bridge 140, which is also the output current of the DC converter 100, and then output the output current of the DC converter 100 to the output current PI control module 302.

The output voltage PI control module 301 is configured to obtain a modulated current according to an output voltage of the secondary side voltage-doubler rectifier bridge 140. Specifically, under the condition of determining the current mode of the DC converter 100, in the process of outputting the output voltage of the secondary side voltage-multiplying rectifier bridge 140 to the output voltage PI control module 301 through the output voltage sampling module 503, the output voltage of the secondary side voltage-multiplying rectifier bridge 140 and the reference voltage need to be subjected to a difference operation to obtain a voltage difference value, and then the voltage difference value is input to the output voltage PI control module 301 to obtain the modulation current. The reference voltage is set according to actual requirements.

The output current PI control module 302 is configured to obtain a modulation parameter according to an output current of the secondary side voltage-doubler rectifier bridge 140. Specifically, under the condition of determining the current mode of the DC converter 100, in the process of outputting the output current of the secondary side voltage-multiplying rectifier bridge 140 to the output current PI control module 302 through the output current sampling module 502, the output current of the secondary side voltage-multiplying rectifier bridge 140 and the modulation current need to be subjected to a difference operation to obtain a current difference value, the current difference value is input to the output current PI control module 302 to obtain a modulation parameter, and the modulation parameter is input to the strategy modulation module 400, so that the strategy modulation module 400 modulates the DC converter 100 in real time. The modulation parameters comprise a first modulation reference voltage, a second modulation reference voltage and a third modulation reference voltage.

The LLC resonant DC converter system with wide gain of the embodiment, based on the characteristics of the dual half-bridge LLC resonant DC converter of the embodiment, modulates the DC converter 100 with different modes according to different input voltages of the DC converter 100, improves more degrees of freedom for the mode conversion of the DC converter 100, and widens the output gain range of the DC converter 100 and the system, i.e., the wide voltage gain output is realized by means of the mode conversion of the DC converter 100; and the influence of load conditions on output gain can be effectively reduced.

The procedure for determining the LLC resonant parameters and frequency ranges of the DC converter 100 of this embodiment is: the current characteristics of the DC converter 100 are analyzed by a first harmonic approximation method to obtain the target frequency range and LLC resonant parameters of the DC converter 100. The target frequency range is the maximum frequency range of the DC converter 100, i.e. the frequency range.

It should be noted that, since the two half-bridge structures of the DC converter 100 are consistent and relatively independent, only any one half-bridge structure is analyzed to design the resonance parameters, in determining the LLC resonance parameters and the frequency conversion range of the DC converter 100 of this embodiment, the resonant cavity, the transformer, and the like involved refer to only the resonant cavity, the transformer, and the like in one half-bridge structure.

Specifically, assume the familyThe working frequency of the system is f _s The working frequency variation range is [ f ] _s.min ，f _s.max ]The voltage of the DC input power supply 110 is U _dc The voltage of the DC input power source 110 varies in a range of [ U ] _dc.min ，U _dc.max ]. Voltage U _dc The square wave u is output through the primary side inversion module 130 _i.sq (t) the value of the first harmonic approximation of the primary side is

. Setting a resonant current and u flowing into a resonant cavity of the DC converter 100 _i.FHA The phase difference between them is theta ₀ The root mean square value of the resonant current is I _r.rms， It is possible to obtain a resonance current of +.>

Further T in one switching cycle of the switching tube in the DC converter _s Can obtain the average value of the current of the direct current input power supply 110

。

Let the resistance of the resistive load 160 of the secondary-side voltage-doubler rectifier bridge 140 be R ₀ The voltage of the resistive load 160 is U _0， The resistance value R is calculated under the condition of First Harmonic Approximation (FHA) ₀ Converted to the primary side, the converted value is

. Due to clamping effect of exciting inductance of resonant cavity of DC converter 100, voltage of secondary side of transformer is changed in square wave, and square wave and u outputted from secondary side of transformer are set _i.FHA The phase angle difference of (t) is ψ _0， The value of the first harmonic approximation of the secondary side is

. Correspondingly, the current and the voltage of the secondary side of the transformer are in the same phase, and the root mean square value of the current of the secondary side of the transformer is set as I _rt.rms The current of the secondary side of the transformer is

。

Assume that the resonant inductance, transformer transformation ratio, excitation inductance and resonant capacitance of the primary side of the transformer are L respectively _r 、n、L _m And C _r . According to the law of conservation of power, the forward transfer function formula of the resonant cavity can be obtained as follows:

。

the forward transfer function of the resonant cavity is taken to obtain the gain of the resonant cavity as follows:

where j is an imaginary unit.

Introducing resonant frequency

Characteristic impedance->

Figure of merit->

Inductance ratio λ=l _r /L _m Normalized frequency f _n =f _s /f _r Obtaining a voltage gain expression:

。

to ensure that the gain of the resonant cavity can meet the output, the minimum gain M during no-load is ensured _min The boundary conditions are satisfied:

。

the loading conditions under no-load conditions at this time are:

，

wherein t is _D For dead time selected according to switching tube time characteristic and junction capacitance characteristic, C _ZVS The capacitance value formed by the stray capacitance of the MOSFET junction capacitance, namely the switch tube junction capacitance, can be regarded as normalA number.

The maximum gain of the resonant cavity is determined by the minimum voltage input when the circuit is operating in the compensation gain mode, and the gain required for power output at the highest quality factor:

，

wherein Q is a quality factor, Q and f _n Related to lambda according to

It is possible to derive Q and the introduced f _n And a function Q (f of lambda _n， λ)。

The maximum load condition that satisfies LLC zero voltage switch is:

。

the set frequency conversion range is obtained by solving according to the gain boundary value:

。

according to the load conditions

The parameter value of the resonant element of the resonant cavity can be obtained by inverse calculation: />

，/>

，/>

，L _m =L _r /λ。

In this embodiment, a first harmonic approximation method (FHA) is used to analyze the voltage-current characteristics of the LLC resonant device, analyze the mechanism of voltage gain generated by resonance, and find the gain boundary and soft switching conditions of the DC converter 100 to obtain the required gain space. And sequentially obtains the inductance ratio, the maximum frequency range, the LLC resonant parameter, and the adapted load condition range of the DC converter 100.

Based on the characteristics of the DC converter 100 of the present embodiment, the operation principle of the present system is explained:

before the system is formally operated, the mode section of the DC converter 100 needs to be divided according to the operating characteristics of the DC converter 100, that is, according to the output voltage of the DC converter 100 and the gain between the output voltages. The divided modal sections of the DC converter 100 are stored in the modal control module 201.

Specifically, a first preset voltage threshold U is set according to actual requirements _th1 And a second preset voltage threshold U _th2 Combined with the minimum voltage U of the DC input power source 110 _dc.min And maximum voltage value U _dc.max And the output voltage of the DC converter 100 and the gain between the output voltages, to define a first modal voltage threshold range [ U ] _th2， U _dc.max ) Threshold range of second mode voltage [ U ] _th1， U _th2 ) And third mode voltage threshold range [ U ] _dc.min， U _th1 )。

In the case where the input voltage of the DC converter 100 is within the first mode voltage threshold range [ U ] _th2， U _dc.max ) The DC converter 100 operates in a first gain mode, which is phase-shift modulated, wherein the first gain mode is a low gain mode. In the case where the input voltage of the DC converter 100 is within the second mode voltage threshold range [ U ] _th1， U _th2 ) The DC converter 100 operates in a compensation gain mode, which may also be referred to as a transition gain mode, in which the modulation mode is frequency modulation. In the case where the input voltage of the DC converter 100 is within the third mode voltage threshold range [ U ] _dc.min， U _th1 ) The DC converter 100 operates in a second gain mode, which is pulse width modulated, wherein the second gain mode is a high gain mode. The system automatically performs the mode division and mode switching functions of the DC converter 100 according to the mode characteristics of the DC converter 100, as follows:

。

a modulation scheme corresponding to the mode of the DC converter 100 is shown in fig. 3.

In the process of the formal operation of the present system, the input voltage, output voltage, and output current of the DC converter 100 are first acquired.

Specifically, the input voltage of the DC converter 100 is collected by an input voltage sampling module 501. The output voltage of the DC converter 100 is collected by the output voltage sampling module 503, and the output current of the DC converter 100 is collected by the output current sampling module 502.

Next, the mode control module 200 of the present system determines the current mode of the DC converter 100 from the input voltage of the DC converter 100.

Specifically, as shown in fig. 4 and 6, the mode control module 201 in the mode control module 200 receives the input voltage of the DC converter 100 transmitted by the input voltage sampling module 501, and determines the input voltage of the DC converter 100. The judgment is as follows:

if the input voltage of the DC converter 100 is within the first mode voltage threshold range, the first gain mode is determined as the current mode, wherein the modulation mode of the first gain mode is phase shift modulation.

If the input voltage of the DC converter 100 is within the second mode voltage threshold range, the compensation gain mode is determined as the current mode, wherein the modulation mode of the compensation gain mode is frequency modulation.

If the input voltage of the DC converter 100 is within the third mode voltage threshold range, the second gain mode is determined as the current mode, wherein the modulation mode of the second gain mode is pulse width modulation.

After determining the current mode of the DC converter 100, a modulation scheme, i.e. a modulation strategy, corresponding to the DC converter 100 is selected by the mode selection module 202. Specifically, if the current mode is the first gain mode, i.e., the low gain mode, the mode selection module sends a low gain mode enabling signal to the PI control module 300 and the strategy modulation module 400, so that the PI control module 300 and the strategy modulation module 400 modulate the DC converter 100 according to the low gain mode.

If the current mode is the compensation gain mode, the mode selection module sends a compensation gain mode enabling signal to the PI control module 300 and the strategy modulation module 400, so that the PI control module 300 and the strategy modulation module 400 modulate the DC converter 100 according to the compensation gain mode.

If the current mode is the second gain mode, i.e. the high gain mode, the mode selection module sends a high gain mode enabling signal to the PI control module 300 and the strategy modulation module 400, so that the PI control module 300 and the strategy modulation module 400 modulate the DC converter 100 according to the high gain mode.

Then, the PI control module 300 of the system obtains modulation parameters according to the current mode, output voltage and output current of the DC converter 100.

Specifically, as shown in fig. 1, 5 a-5 c, based on the current mode of the DC converter 100, the output voltage U of the DC converter 100 is output through the output voltage sampling module 503 _o.fb In the process of reaching the output voltage PI control module 301, the output voltage U of the DC converter 100 is required to be first _o.fb With reference voltage U _o.ref Performing difference operation to obtain a voltage difference U _o.err Then the voltage difference U is calculated _o.err Is input into an output voltage PI control module 301 to obtain a modulation current I _o.ref . Then, the output current I of the DC converter 100 is output through the output current sampling module 502 _o.fb In the process of the output current PI control module 302, the output current I of the DC converter 100 is required to be first _o.fb And modulating current I _o.ref Performing difference operation to obtain a current difference I _o.err Then the current difference I is calculated _o.err The modulation parameters are input to the output current PI control module 302, so that the output current PI control module 302 controls the strategy modulation module 400 in real time. Wherein the modulation parameter comprises a first modulation reference voltage V _Φ Second modulated reference voltage V _f Third modulated reference voltage V _D 。

As shown in fig. 5a, in the case that the current mode is the low gain mode, the modulation parameter outputted by the output current PI control module 302 includes a first modulation reference voltage V _Φ A second modulation reference voltage V with a constant value _f Third with value zeroModulating reference voltage V _D . Wherein the third modulation reference voltage V _D Is not actually generated.

As shown in fig. 5b, in the case that the current mode is the compensation gain mode, the modulation parameter outputted by the output current PI control module 302 includes a first modulation reference voltage V with a value of zero _Φ Second modulated reference voltage V _f、 The value is the second modulation reference voltage V _f Half of the third modulation reference voltage V _D。

As shown in fig. 5c, in the case that the present mode is the high gain mode, the modulation parameter outputted by the output current PI control module 302 includes a first modulation reference voltage V with a value of zero _Φ A second modulation reference voltage V with a constant value _f Third modulated reference voltage V _D。

Then, the on-off state of each switching tube of the DC converter 100 is controlled according to the current mode and modulation parameters of the DC converter 100 by the strategy modulation module 400 of the system.

Specifically, the modulation is implemented by using a carrier comparison method according to the current mode and modulation parameters of the DC converter 100. As shown in fig. 6, the policy modulation module 400 first modulates the first modulation reference voltage V sent by the PI control module 300 _Φ Second modulated reference voltage V _f Third modulated reference voltage V _D Signal reconstruction is carried out to obtain the corresponding carrier wave count value, namely V _Φ、 V _f、 V _D Reconstructing a count value of a counter usable by the digital controller. Wherein the count value of the corresponding carrier is the first modulation reference voltage V _Φ Second modulated reference voltage V _f Third modulated reference voltage V _D Reconstructing a count value of the formed carrier wave. The count value of the corresponding carrier wave comprises a first count value, a second count value and a third count value.

Based on the current mode of the DC converter 100 and the count value of the corresponding carrier, the current mode and the count value of the corresponding carrier are logically compared with the count value of the first carrier comparator and the count value of the second carrier comparator respectively to obtain the switching signal of each switching tube in the DC converter 100, so as to realize the on-off state of each switching tube of the DC converter 100 through the switching signal of each switching tube, and further obtain the driving signal of the secondary side voltage-multiplying rectifier bridge 140. The first carrier comparator is a carrier comparator that compares, in the current mode, a carrier count value generated by the primary side inversion mode 130 with a first count value and a second count value in the count values of the corresponding carriers, and the count value of the first carrier comparator is a carrier count value generated by the primary side inversion mode 130. The second carrier comparator is a carrier comparator that compares a carrier count value generated by the secondary side voltage-multiplying rectifier bridge 140 with a third count value in the count values of the corresponding carriers in the current mode, and the count value of the second carrier comparator is the carrier count value generated by the secondary side voltage-multiplying rectifier bridge 140.

Specifically, as shown in FIG. 6, the V _Φ The count value CNT of the phase-shift counter of the first and second resonant legs of the primary-side inverter module 130 of the DC converter 100 is reconstructed _Φ . From V _f Reconstructing a count value CNT of a switching tube carrier counter _f . From V _D The count value CNT of the pulse width PWM comparison counter of the secondary side voltage-doubler rectifier bridge 140 is reconstructed _D 。

CNT _f Determining the operating frequency and carrier frequency of the system, CNT _Φ、 CNT _D The switching tube S can be obtained after the logic operation and comparison with the carrier counter ₁ ~S ₅ Switch signal of (a), switch tube S ₁ ~S ₅ Comprises: PWM_1 is the first switch tube S ₁ Is a driving signal G of (2) _s1、 PWM_2 is the second switching tube S ₂ Is a driving signal G of (2) _s2、 PWM_3 is the second switching tube S ₃ Is a driving signal G of (2) _s3、 PWM_4 is the second switching tube S ₄ Is a driving signal G of (2) _s4、 PWM_5 is the second switching tube S ₅ Drive signal G _s5。 Wherein, pwm_1 and pwm_2 are pulse width signals with duty ratio constant 50% and complementary, pwm_3 and pwm_4 are pulse width signals with duty ratio constant 50% and complementary, and a phase difference phi exists between rising edges of pwm_1 and pwm_3; the PWM_5 signal has a frequency equal to f _s A pulse width signal with a duty cycle D.

Based on the current mode,CNT _Φ、 CNT _f And CNTs _D And respectively comparing the value with the count value of the first carrier comparator and the count value of the second carrier comparator, generating a switching signal of each switching tube in the primary side inversion module 130 by phase shifting and frequency mixing modulation, and generating a driving signal of the secondary side voltage-multiplying rectifier bridge 140 by pulse width modulation.

As shown in fig. 7a, in the case that the current mode is the low gain mode, the modulation mode of the system is phase shift modulation, and the PI control module 300 outputs a modulation parameter (a first modulation reference voltage V _Φ And a second modulation reference voltage V of a fixed value _f Does not generate V _D ) To the policy modulation module 400. The policy modulation module 400 performs signal reconstruction on the received modulation parameters to generate corresponding carrier count values, which are the first count values CNT respectively _Φ And a second count value CNT _f . And then the second count value CNT is added _f The primary side inversion module 130 generates a primary side carrier wave to obtain a carrier wave count value of the primary side carrier wave. Then, the carrier count value of the primary side carrier and 0.5CNT are used _f And CNTs _Φ The first carrier comparator input to the strategy modulation module 400 obtains a first switch tube S under a low gain mode ₁ To fourth switching tube S ₄ Is provided. Fifth switch tube S ₅ Switch tube signal and second switch tube S ₂ And a fourth switching tube S ₄ Is aligned with the switching tube signal of the (c).

Wherein the DC converter 100 operates at a fixed frequency in a low gain mode, i.e., in phase-shift modulation, V _f Is a certain constant value related to the resonance frequency. V (V) _f Is reconstructed into a load count value between a carrier generated on the primary side and a carrier generated on the secondary side, namely CNT _f And should satisfy the CNT _f Is constant. At this time, V is not generated _D I.e. V _D Not taking part in the fifth switching tube S ₅ And then fifth switch tube S ₅ Is composed of S ₂ And S is equal to ₄ Decision, satisfy S ₅ =S ₂ &S ₄ 。

As shown in fig. 7b, the current mode is compensationIn the case of the gain mode, the modulation mode of the system is frequency modulation, and the PI control module 300 outputs a modulation parameter (a first modulation reference voltage V with a value of 0 _Φ Second modulated reference voltage V _f And a third modulation reference voltage V _D =0.5V _f ) To the policy modulation module 400. The policy modulation module 400 performs signal reconstruction on the received modulation parameters to generate corresponding carrier count values, which are the first count values CNT respectively _Φ Second count value CNT _f And a third count value CNT _D . And then the second count value CNT is added _f The primary side inversion module 130 generates a primary side carrier wave to obtain a carrier wave count value of the primary side carrier wave. Then, the carrier count value of the primary side carrier and 0.5CNT are used _f And CNTs _Φ The first carrier comparator is input to the strategy modulation module 400 to obtain a first switch tube S under the compensation gain mode ₁ To fourth switching tube S ₄ Is provided. At the same time, the second count value CNT is counted _f The secondary side voltage-doubling rectifying bridge 140 generates a secondary side carrier wave to obtain a carrier wave count value of the secondary side carrier wave. Then, the carrier count value and CNT of the secondary side carrier are calculated _D The fifth switching tube S is obtained in the compensation gain mode by inputting the fifth switching tube S into the second carrier comparator of the strategy modulation module 400 ₅ Is provided.

Wherein in the compensation gain mode, i.e. in frequency modulation, V _Φ =0，V _f Is reconstructed into a load count value between a carrier generated on the primary side and a carrier generated on the secondary side, namely CNT _f The switching frequency of the system follows the CNT _f Changing and changing. V (V) _D The count value CNT of the switching tube carrier frequency counter is reconstructed through data _D And should satisfy the CNT _D =0.5CNT _f。

As shown in fig. 7c, when the current mode is the high gain mode, the modulation mode of the system is pulse width modulation, and the PI control module 300 outputs a modulation parameter (a first modulation reference voltage V with a value of 0 _Φ A second modulation reference voltage V with a constant value _f And a third modulation reference voltage V _D ) To the policy modulation module 400. Strategy ofThe slightly modulating module 400 firstly performs signal reconstruction on the received modulating parameters to generate corresponding carrier count values, which are respectively the first count value CNT _Φ Second count value CNT _f And a third count value CNT _D . And then the second count value CNT is added _f The primary side inversion module 130 generates a primary side carrier wave to obtain a carrier wave count value of the primary side carrier wave. Then, the carrier count value of the primary side carrier and 0.5CNT are used _f And CNTs _Φ The first carrier comparator input to the strategy modulation module 400 obtains a first switch tube S under a high gain mode ₁ To fourth switching tube S ₄ Is provided. At the same time, the second count value CNT is counted _f The secondary side voltage-doubling rectifying bridge 140 generates a secondary side carrier wave to obtain a carrier wave count value of the secondary side carrier wave. Then, the carrier count value and CNT of the secondary side carrier are calculated _D The fifth switching tube S under the high gain mode is obtained by inputting the fifth switching tube S into the second carrier comparator of the strategy modulation module 400 ₅ Is provided.

Wherein in high gain mode, i.e. in pulse width modulation, V _Φ =0, dc converter 100 operates at a constant frequency, V _f Is a certain constant value related to the resonance frequency. V (V) _f Is reconstructed into a load count value between a carrier generated on the primary side and a carrier generated on the secondary side, namely CNT _f And should satisfy the CNT _f Is constant. V (V) _D The count value CNT of the switching tube carrier frequency counter is reconstructed through data _D Fifth switch tube S ₅ Duty cycle of (c) is dependent on CNT _D Is changed by a change in (a).

It should be further noted that the working principle of the first carrier comparator is: as shown in fig. 8, the first Carrier comparator receives the Carrier count value Carrier of the primary side Carrier ₁ (i.e., carrier signal of primary side carrier in FIG. 8), CNT _f And CNTs _Φ。 When 0.5CNT _f＞ Carrier ₁ At the time S ₁ Gate signal G of (2) _s1 After a dead time t _d Is later set to 1, S ₂ Gate signal G of (2) _s2 Set to 0. When 0.5CNT _f ≤Carrier ₁ At the time S ₁ Gate signal G of (2) _s1 Is set as 0,S ₂ After a dead time t is passed by the gate signal of (2) _d And then set to 1. When CNT is present _Φ ＜Carrier ₁ Logic flag bit l ₁ After a dead time t _d And then set to 1. When CNT is present _Φ ≥Carrier ₁ Logic flag bit l ₁ =0. When 0.5CNT _f +CNT _Φ ＞Carrier ₁ Logic flag bit l ₂ After a dead time t _d And then set to 1; when 0.5CNT _f +CNT _Φ ≤Carrier ₁ Logical flag bit l ₂ =0。S ₃ Gate signal G of (2) _s3 By logical flag bit l ₁ And/l ₂ Logic operation is obtained, and G is satisfied _s3 ＝l ₁ &l _2； S ₄ Gate signal G of (2) _s4 And a switch tube S ₃ Gate signal G of (2) _s3 Complementary.

The working principle of the second carrier comparator is as follows: as shown in fig. 9, the second Carrier comparator receives the Carrier count value Carrier of the Carrier on the secondary side ₂ (i.e., carrier signal of carrier on secondary side in FIG. 8) and CNT _D . The carrier signal on the secondary side is delayed by half a switching period compared to the wave signal on the primary side. When CNT is present _D ＞Carrier ₂ At the time S ₅ Gate signal G of (2) _s5 After a dead time t _d And then set to 1; when CNT is present _D ≤Carrier ₂ At the time S ₅ Gate signal G of (2) _s5 Set to 0.

Each generated switching signal is input to the primary side inverter module 130 and the secondary side voltage-multiplying rectifier bridge 140, and the switching actions of the switching tubes are controlled, so that each switching tube is in an on state and/or an off state. The direct current input power supply 110 generates a square wave after the switching action of the primary side inversion module 130, and then forms a resonant current and a resonant voltage through the resonant cavity to transmit electric energy to the secondary side voltage-multiplying rectifier bridge 140. After the secondary side of the isolation transformer senses electricity, electric energy is output to the resistive load 160 through the secondary side voltage doubling rectifier bridge 140. The on-off state of each switch tube and the working condition of each resonant bridge arm are shown in the following table 1.

In table 1, "1" indicates that the switching tube of the resonant arm is in an on state, "0" indicates that the switching tube of the resonant arm is in an off state, "x" indicates an indefinite state, that is, "0" may be possible, and "1" may be possible, and the determination of "x" is made by the switching tube S ₅ Duty cycle determination of (2). First switching tube S in phase-shift modulation mode ₁ A second switch tube S in a conducting state ₂ To a fifth switching tube S ₅ All in the off state, for example, in which the first resonant arm begins to operate with the input clamped to U _dc The first resonant cavity operating frequency is equal to the resonant frequency, and the bridge arm of the secondary side voltage doubling rectifier bridge 140 does not operate.

As shown in fig. 10, the present system modulates the waveform obtained by the DC converter 100 with phase-shift modulation. In fig. 10, a first switching tube S ₁ And a third switching tube S ₃ Is consistent with the waveform of the second switch tube S ₂ And a fourth switching tube S ₄ Is identical in waveform. First switching tube S ₁ And a second switching tube S ₂ Between and third switching tube S ₃ And a fourth switching tube S ₄ Phase-shifting staggering angles phi exist between the two. Fifth switch tube S ₅ Is aligned with the second switch tube S ₂ And a fourth switching tube S ₄ And extend a waveform hold to the first switching tube S ₁ And a third switching tube S ₃ Is included in the waveform of (a).

As shown in fig. 11, the present system modulates the waveform obtained by the DC converter 100 with frequency modulation. In fig. 11, a first switching tube S ₁ Waveform of (2) and second switching tube S ₂ Is complementary to the waveform of the first switch tube S ₁ Waveform and second switching tube S ₂ Is 0. Third switch tube S ₃ Waveform of (d) and fourth switching tube S ₄ Is complementary to the waveform of the third switch tube S ₃ Waveform of (d) and fourth switching tube S ₄ Is 0. Fifth switch tube S ₅ Is aligned with the second switch tube S ₂ And a fourth switching tubeS ₄ Switching tube signal of (a), i.e. fifth switching tube S as shown in FIG. 11 ₅ Is positioned by a second switching tube S ₂ And a fourth switching tube S ₄ Is within the range of the switching tube signal formation.

As shown in fig. 12, the present system modulates the waveform obtained by the DC converter 100 with pulse width modulation. In fig. 12, three different operating frequencies (f ₁ 、f ₂ And f ₃ ) Lower first switch tube S ₁ To a fifth switching tube S ₅ Is a waveform of (a). Under the same working frequency, a first switching tube S ₁ And a third switching tube S ₃ Is consistent with the waveform of the second switch tube S ₂ And a fourth switching tube S ₄ Is identical in waveform. First switching tube S ₁ And a second switching tube S ₂ Between and third switching tube S ₃ And a fourth switching tube S ₄ The phase angles between the two are all 0. Fifth switch tube S ₅ Is aligned with the second switch tube S ₂ And a fourth switching tube S ₄ Is provided. Wherein the abscissa of fig. 10-12 represents time.

Based on the structure and the working principle of the system, the current mode of the DC converter 100 is determined according to the input voltage of the DC converter 100, so that different modes of the DC converter 100 are caused by the input voltage of different DC converters 100, and the DC converter 100 is controlled to perform real-time mode switching. And then, according to the current mode adaptation modulation strategy, calculating the deviation between the output voltage of the DC converter 100 and the reference voltage to obtain a modulation current, and then, calculating the deviation between the modulation current and the DC converter 100 to generate a phase shift angle of the primary side inversion module 130, a switching tube switching frequency and a pulse width duty ratio of the secondary side voltage doubling rectifier bridge 140, and generating a counter carrier modulation comparison value through data reconstruction. And then, according to the numerical carrier comparison, the phase shift and the frequency mixing modulation generate a driving signal of the primary side inversion module 130, and the pulse width modulation generates a driving signal of the secondary side voltage doubling rectifier bridge 140. Thus, the DC converter 100 is modulated in different modes, the gain range of the DC converter 100 is widened, the DC converter 100 is controlled to output stable voltage, the influence of load is reduced, the working efficiency of the DC converter 100 and the system is improved, and the stable operation of the DC converter 100 and the system is ensured.

The result of the operation achieved by the present system is compared with the result of the operation achieved by PFM modulation of a conventional LLC resonant converter as shown in fig. 13, which is a gain profile achieved by PFM modulation of a conventional LLC resonant converter, as shown in fig. 14, which is a gain profile of a DC converter 100 achieved by the present system modulation. The abscissa of fig. 13 and 14 represents the normalized frequency, and the ordinate represents the gain. The gain range achieved by conventional LLC resonant converters with PFM modulation is [0.8,1.4]. On the other hand, based on the characteristics of the DC converter 100, the low gain region in fig. 14 is the gain region of the DC converter 100 in the low gain mode, the transitional gain region in fig. 14 is the gain region of the DC converter 100 in the complementary gain mode, and the high gain region in fig. 14 is the gain region of the DC converter 100 in the high gain mode. The gain range of the DC converter 100 is thus 0, 2. More circuit modes are built through the DC converter 100 and the system to widen the gain range, a wide output gain range of 0-2 times is realized, the influence of load conditions on the output gain is effectively reduced, and the output stability of the system and the DC converter 100 is ensured.

In this embodiment, an LLC resonant type DC converter system having a wide gain includes: the system comprises a DC converter, a mode control module, a PI control module and a strategy modulation module, wherein the DC converter is a double half-bridge LLC resonant DC converter. Here, the DC converter of this embodiment includes a double half-bridge LLC resonant structure, and more circuit modes can be constructed through the double half-bridge LLC resonant structure to widen the output gain range of the DC converter, and achieve a wide output gain range of 0 to 2 times. When the independent resonant cavity structure is used for phase-shifting modulation, the problem of low load efficiency caused by overlarge phase-shifting angle can be effectively avoided.

Embodiment two: based on the same inventive concept, the second embodiment of the present invention further provides an application method of an LLC resonant DC converter system with wide gain, as shown in fig. 15, applied to an LLC resonant DC converter system with wide gain according to embodiment one, the method including:

s501, after the resonance parameters and the modal interval of a DC converter of the system are determined, the input voltage, the output voltage and the output current of the DC converter are obtained;

s502, determining the current mode of the DC converter according to the input voltage of the DC converter through a mode control module of the system;

s503, obtaining modulation parameters according to the current mode, the output voltage and the output current of the DC converter through a PI control module of the system;

s504, controlling the on-off state of each switching tube of the DC converter according to the current mode of the DC converter and the modulation parameters through a strategy modulation module of the system.

As an alternative embodiment, the determining the resonance parameter of the DC converter of the system includes:

As an alternative embodiment, the determining the current mode of the DC converter according to the input voltage of the DC converter includes:

if the input voltage of the DC converter is in the second mode voltage threshold range, determining a compensation gain mode as the current mode, wherein the modulation mode of the compensation gain mode is frequency modulation;

Since the application method of the LLC resonant DC converter system with wide gain described in this embodiment is a method adopted to implement the LLC resonant DC converter system with wide gain described in embodiment one of the present application, those skilled in the art will be able to understand the specific implementation of the application method of the LLC resonant DC converter system with wide gain and various modifications thereof based on the LLC resonant DC converter system with wide gain described in embodiment one of the present application, and therefore how to implement the application method of the LLC resonant DC converter system with wide gain in embodiment one of the present application will not be described in detail here. The method adopted by those skilled in the art to implement the LLC resonant DC converter system with wide gain in the first embodiment of the present application falls within the scope of protection intended by the present application.

It will be appreciated by those skilled in the art that embodiments of the present invention may be provided as a method, system, or computer program product. Accordingly, the present invention may take the form of an entirely hardware embodiment, an entirely software embodiment or an embodiment combining software and hardware aspects. Furthermore, the present invention may take the form of a computer program product embodied on one or more computer-usable storage media (including, but not limited to, disk storage, CD-ROM, optical storage, and the like) having computer-usable program code embodied therein.

The present invention is described with reference to flowchart illustrations and/or block diagrams of methods, apparatus (systems) and computer program products according to embodiments of the invention. It will be understood that each flow and/or block of the flowchart illustrations and/or block diagrams, and combinations of flows and/or blocks in the flowchart illustrations and/or block diagrams, can be implemented by computer program instructions. These computer program instructions may be provided to a processor of a general purpose computer, special purpose computer, embedded processor, or other programmable data processing apparatus to produce a machine, such that the instructions, which execute via the processor of the computer or other programmable data processing apparatus, create means for implementing the functions specified in the flowchart flow or flows and/or block diagram block or blocks.

These computer program instructions may also be stored in a computer-readable memory that can direct a computer or other programmable data processing apparatus to function in a particular manner, such that the instructions stored in the computer-readable memory produce an article of manufacture including instruction means which implement the function specified in the flowchart flow or flows and/or block diagram block or blocks.

These computer program instructions may also be loaded onto a computer or other programmable data processing apparatus to cause a series of operational steps to be performed on the computer or other programmable apparatus to produce a computer implemented process such that the instructions which execute on the computer or other programmable apparatus provide steps for implementing the functions specified in the flowchart flow or flows and/or block diagram block or blocks.

While preferred embodiments of the present invention have been described, additional variations and modifications in those embodiments may occur to those skilled in the art once they learn of the basic inventive concepts. It is therefore intended that the following claims be interpreted as including the preferred embodiments and all such alterations and modifications as fall within the scope of the invention.

It will be apparent to those skilled in the art that various modifications and variations can be made to the present invention without departing from the spirit or scope of the invention. Thus, it is intended that the present invention also include such modifications and alterations insofar as they come within the scope of the appended claims or the equivalents thereof.

Claims

1. An LLC resonant DC converter system with a wide gain, comprising: the system comprises a DC converter, a mode control module, a PI control module and a strategy modulation module, wherein the DC converter is a double half-bridge LLC resonant DC converter;

the output end of the mode control module is connected with the input end of the PI control module, the output end of the PI control module is connected with the input end of the strategy modulation module, and the output end of the strategy modulation module is connected with the DC converter;

The DC converter includes: the direct current input power supply, the bus input capacitor, the primary side inversion module, the secondary side voltage doubling rectifier bridge, the output capacitor group and the resistive load;

the positive electrode of the bus input capacitor and the positive electrode of the primary side inversion module are connected with the positive electrode of the direct current input power supply, and the negative electrode of the bus input capacitor and the negative electrode of the primary side inversion module are connected with the negative electrode of the direct current input power supply; the secondary side voltage doubling rectifier bridge, the output capacitor group and the resistive load are connected in parallel;

the secondary side voltage doubling rectifier bridge comprises: a second side of the first transformer, a second side of the second transformer, a first diode, a second diode, a third diode and a fifth switching tube;

the second side synonym end of the first transformer is connected with the second side synonym end of the second transformer, the second side synonym end of the first transformer, the positive electrode of the first diode and the negative electrode of the second diode are connected, the second side synonym end of the second transformer, the negative electrode of the third diode and the source electrode of the fifth switch tube are connected, the negative electrode of the first diode is a first port of the second side voltage-multiplying rectifier bridge, the drain electrode of the fifth switch tube is a second port of the second side voltage-multiplying rectifier bridge, and the intersection point of the positive electrode of the second diode and the positive electrode of the third diode is a third port of the second side voltage-multiplying rectifier bridge;

The DC converter is used for firstly controlling the DC converter to switch between a first state of the first gain mode and a second state of the first gain mode if the current mode is the first gain mode, and then controlling the DC converter to switch between a third state of the first gain mode and a fourth state of the first gain mode, wherein the DC converter comprises a first switch tube, a second switch tube, a third switch tube, a fourth switch tube and a fifth switch tube, the first state of the first gain mode is that the first switch tube is in a conducting state, the second switch tube to the fifth switch tube are all in a disconnecting state, the second state of the first gain mode is that the second switch tube is in a conducting state, the first switch tube and the third switch tube to the fifth switch tube are all in a disconnecting state, the third state of the first gain mode is that the first switch tube and the third switch tube are all in a conducting state, the second switch tube and the fourth switch tube are all in a signal conducting state, and the fourth switch tube are all in a signal conducting state;

In a first state of the first gain mode, clamping a voltage of a first resonant bridge arm of the DC converter to a voltage of a direct current input power supply of the system, wherein a first resonant cavity working frequency of the DC converter is equal to a resonant frequency of the DC converter; in a third state of the first gain mode and a fourth state of the first gain mode, the first resonant bridge arm and the second resonant bridge arm of the DC converter work simultaneously and have phase-shifting staggered angles, and the working frequencies of the first resonant cavity of the DC converter and the second resonant cavity of the DC converter are equal to the resonant frequency;

if the current mode is a compensation gain mode, the DC converter is controlled to switch between a first state of the compensation gain mode and a second state of the compensation gain mode, wherein the first state of the compensation gain mode is that the first resonant bridge arm and the second resonant bridge arm are both in a conducting state, the second switching tube, the fourth switching tube and the fifth switching tube are both in a disconnecting state, the second state of the compensation gain mode is that the first switching tube and the third switching tube are both in a disconnecting state, the second switching tube, the fourth switching tube and the fifth switching tube are both in a conducting state, in the first state of the compensation gain mode and the second state of the compensation gain mode, the first resonant bridge arm and the second resonant bridge arm are simultaneously operated, the phase angle of the first resonant cavity and the second resonant cavity are both in an unfixed state, the working frequency of the first resonant cavity and the fifth resonant cavity is limited to be in a target frequency range, the switching tube signal of the fifth switching tube is aligned with the second switching tube and the fourth switching tube is in a target frequency range, and the frequency of the DC converter is in a maximum frequency range of 5.0;

If the current mode is a second gain mode, the DC converter is controlled to switch between a first state of the second gain mode and a second state of the second gain mode, wherein the first state of the second gain mode is that the first switching tube and the third switching tube are in an on state, the second switching tube and the fourth switching tube are in an off state, the on-off state of the fifth switching tube is determined by the duty ratio of the fifth switching tube, the second state of the second gain mode is that the first switching tube and the third switching tube are in an off state, the second switching tube and the fourth switching tube are in an on state, switching tube signals of the fifth switching tube are aligned with switching tube signals of the second switching tube and the fourth switching tube, in the second gain, the first resonant cavity and the second resonant cavity are simultaneously operated, the phase angles of the first resonant cavity and the second resonant cavity are zero, the second resonant cavity and the fourth resonant cavity are equal to the duty ratio of the fourth switching tube and the fourth switching tube is equal to or more than the fifth switching tube, and the fourth switching tube is in an on state of the fourth switching tube is aligned with the fourth switching tube, and the fourth switching tube is equal to or more than the duty ratio of the fourth switching tube is equal to 0.

2. The system of claim 1, wherein the primary side inverter module comprises: the first resonant bridge arm, the second resonant bridge arm, the first resonant cavity, the second resonant cavity, the primary side of the first transformer and the primary side of the second transformer;

3. The system of claim 1, wherein the output capacitor bank comprises: a first output capacitor and a second output capacitor;

4. The system of claim 1, wherein the PI control module comprises: an output voltage PI control module and an output current PI control module; the system also comprises an output current sampling module and an output voltage sampling module;

5. A method of applying an LLC resonant DC converter system with a wide gain, characterized in that it is applied to an LLC resonant DC converter system with a wide gain as claimed in any of claims 1 to 4, the method comprising:

determining, by a mode control module of the system, a current mode of the DC converter according to an input voltage of the DC converter, comprising:

if the input voltage of the DC converter is in a third mode voltage threshold range, determining a second gain mode as the current mode, wherein the modulation mode of the second gain mode is pulse width modulation;

and controlling the on-off state of each switching tube of the DC converter according to the current mode of the DC converter and the modulation parameters by a strategy modulation module of the system, wherein the strategy modulation module comprises the following steps:

if the current mode is the first gain mode, the DC converter is controlled to switch between a first state of the first gain mode and a second state of the first gain mode, and then the DC converter is controlled to switch between a third state of the first gain mode and a fourth state of the first gain mode, wherein the DC converter comprises a first switch tube, a second switch tube, a third switch tube, a fourth switch tube and a fifth switch tube, the first state of the first gain mode is the on state of the first switch tube, the second switch tube to the fifth switch tube are all in the off state, the second state of the first gain mode is the on state of the second switch tube, the third switch tube to the fifth switch tube are all in the off state, the third state of the first gain tube is the on state of the first switch tube and the third switch tube, the second switch tube and the fourth switch tube are all in the on state of the fourth switch tube and the fourth switch tube, the fourth switch tube is the off state of the fourth switch tube and the fourth switch tube is the fourth switch tube, the fourth switch tube is the signal tube;

if the current mode is the compensation gain mode, the DC converter is controlled to switch between a first state of the compensation gain mode and a second state of the compensation gain mode, wherein the first state of the compensation gain mode is that the first resonant arm and the second resonant arm are both in a conducting state, the second switching tube, the fourth switching tube and the fifth switching tube are both in a disconnecting state, the second state of the compensation gain mode is that the first switching tube and the third switching tube are both in a disconnecting state, the second switching tube, the fourth switching tube and the fifth switching tube are both in a conducting state, in the first state of the compensation gain mode and the second state of the compensation gain mode, the first resonant arm and the second resonant arm are simultaneously operated, the phase angle of the first resonant cavity and the second resonant cavity is zero, the operating frequency of the first resonant cavity and the second resonant cavity is not fixed, the operating frequency of the second resonant cavity is limited to be within a target frequency range, the switching tube is aligned with the second switching tube and the fourth switching tube is a maximum frequency range, and the switching tube is a maximum frequency of the DC converter is 5;

If the current mode is the second gain mode, the DC converter is controlled to switch between a first state of the second gain mode and a second state of the second gain mode, wherein the first state of the second gain mode is that the first switching tube and the third switching tube are both in an on state, the second switching tube and the fourth switching tube are both in an off state, the on-off state of the fifth switching tube is determined by the duty ratio of the fifth switching tube, the second state of the second gain mode is that the first switching tube and the third switching tube are both in an off state, the second switching tube and the fourth switching tube are both in an on state, switching tube signals of the fifth switching tube are aligned with switching tube signals of the second switching tube and the fourth switching tube, in the second gain mode, the first resonant bridge arm and the second resonant tube work simultaneously, the phase angles of the first resonant cavity and the second resonant cavity are zero, the second resonant cavity and the fourth switching tube are equal to the duty ratio of the fourth switching tube and the fourth switching tube is equal to the duty ratio of the fourth switching tube to the fourth switching tube, and the fourth switching tube is in an on state of the fourth switching tube is equal to or more than 0.

6. The method of claim 5, wherein determining the resonant parameters of the DC converter of the system comprises: