CN114465489A - Full-half-bridge resonant converter and voltage balance control method thereof - Google Patents

Full-half-bridge resonant converter and voltage balance control method thereof Download PDF

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CN114465489A
CN114465489A CN202210104033.5A CN202210104033A CN114465489A CN 114465489 A CN114465489 A CN 114465489A CN 202210104033 A CN202210104033 A CN 202210104033A CN 114465489 A CN114465489 A CN 114465489A
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voltage
bridge
full
normalized
resonant converter
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胡松
汪锐
钟黎萍
杨浩东
康乐
杨晴晴
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Changshu Institute of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The invention discloses a full-half-bridge resonant converter and a voltage balance control method thereof, wherein a primary side of the full-half-bridge resonant converter is a full bridge and is provided with 4 switching tubes, a resonant inductor and a capacitor, a secondary side of the full-half-bridge resonant converter is a half bridge and is provided with 2 switching tubes, and the total number of the switching tubes is 6. The invention adopts a fundamental wave approximation method to analyze a power model and soft switching conditions, and provides a voltage balance control strategy, so that the converter can realize the voltage balance of the primary side and the secondary side under a wide voltage range, and simultaneously ensures that all switch tubes realize soft switching, thereby improving the efficiency of the converter.

Description

Full-half-bridge resonant converter and voltage balance control method thereof
Technical Field
The invention relates to a control technology of an isolated high-frequency resonant converter, in particular to a topology of a full-half-bridge resonant converter and a voltage balance control method thereof.
Background
In the composition of a new energy electric automobile, an On-Board-Charger OBC (On-Board-Charger) is one of core devices and consists of a power factor correction PFC and an isolation bidirectional DC-DC. Isolated bidirectional DC-DC converters are receiving more and more attention due to their advantages of high power density, electrical isolation, bidirectional energy transmission, etc. However, when the voltage gain deviates from 1, problems such as loss of soft switching, increased circulating current, transformer saturation, etc. occur, resulting in a decrease in converter efficiency.
Resonant dual active bridge converters have a wide soft switching range and some phase/shift control schemes have been proposed. Traditionally, dual active bridge converters employ a Single Phase Shift (SPS) modulation strategy; by adopting the strategy, the original secondary bridge of the double-active-bridge converter is required to work under 50% duty ratio at the same time, and the phase shift of the original secondary bridge is adjustable. In addition, the condition that the normalized voltage gain of the dual-active-bridge converter is equal to 1 is defined as a voltage balance condition. In this condition, the DAB converter has the lowest circulating current and the widest Zero Voltage Switching (ZVS) range. However, SPS-based dual active bridge converters suffer from increased circulating current and loss of ZVS when the voltage conversion ratio deviates from 1.
To solve this problem, different improved modulation strategies have been proposed by the scholars. Wherein the Extended Phase Shift (EPS) modulation is to reduce the duty cycle of the output voltage of a full bridge below 0.5. This is usually achieved by introducing an additional intra-bridge phase shift between the drive signals of a full bridge, wherein the internal phase shift angle is determined by the actual voltage conversion ratio. Therefore, the output voltage of one full bridge becomes a three-level waveform, while the output voltage of the other full bridge is still a two-level square wave. Compared with SPS, the EPS scheme not only reduces the circulating current, but also expands the ZVS range. However, when the converter state is switched between buck and boost mode, the control signals of the two full bridges need to be interchanged.
In order to maintain good dynamic performance, researchers have proposed Double Phase Shift (DPS) modulation. Like EPS, DPS significantly reduces the cycle power, expanding the ZVS range. Furthermore, unlike EPS, DPS simultaneously produces one and the same intra-bridge phase shift on two full bridges. Thus, the output voltages of both full bridges are symmetrical three-level waveforms. However, both EPS and DPS have only two degrees of freedom of control, which prevents further performance gains.
Therefore, the scholars propose Triple Phase Shift (TPS) modulation. Unlike DPS, there are two unequal intra-bridge phase shifts in TPS. However, due to the additional degree of control freedom, the optimal modulation strategy in the form of an analytical solution is difficult to determine. One approach to solving this problem is to derive an optimal strategy using fourier analysis. However, because the result is an expression with infinite term coefficients, the analysis results tend to be complex and not intuitive. Another approach is to use a convex optimization tool to derive the results. In many cases, however, the mathematical expression being solved is a non-convex problem, and thus there is no globally optimal solution. In addition, it is also highly dependent on the transducer parameters, which makes it difficult to apply in practice. Further, in EPS, DPS, and TPS, the output voltage of the full bridge always has a zero level period. During this time no active power transfer takes place by the converter, which impairs the power transfer capability of DAB.
Disclosure of Invention
The invention provides a voltage balance control strategy of a full-half-bridge resonant converter to meet the requirement of a wide voltage range, and the voltage balance control strategy is used for expanding the soft switching range of the full-half-bridge resonant converter and improving the overall efficiency.
The technical solution for realizing the purpose of the invention is as follows: aiming at a high-frequency resonant converter, a full-half-bridge resonant circuit structure and a voltage balance control strategy for expanding the range of a Zero Voltage Switch (ZVS), reducing energy consumption and improving the overall efficiency are provided.
The invention discloses a full half-bridge resonant converter, which comprises a full bridge at the primary side, a half bridge at the secondary side and a resonant capacitor CpResonant inductor LsThe turn ratio is 1: n high-frequency transformer Tr;
wherein the primary side full-bridge comprises a switch tube S1～S4Body diode ds1～ds4Parasitic capacitance Cs1～Cs4(ii) a The secondary side half-bridge comprises a switching tube S5～S6Body diode ds5～ds6Parasitic capacitance Cs5～Cs6Voltage equalizing capacitor Co1And Co2；
VinAnd VoutInput voltage and output voltage, iLAnd ioRespectively a resonant current and an output current;
and the positive half-cycle area of the primary side fundamental wave voltage and the positive half-cycle area of the secondary side fundamental wave voltage are adjusted to be equal, so that the voltage is balanced.
Preferably, by setting S1～S4The pulse widths of the four switches regulate the gate control signal of the pulse width controller to generate a three-level PWM voltage waveform.
Preferably, switch S1And S2The duty cycle of (2) is 50%; switch S4Is reduced to δ, S3The pulse width is increased to 2 pi-delta, switch S1And S4The on-time of (c) is synchronized.
Preferably, by setting S5～S6The pulse widths of the two switches are adjusted by the gate control signal of the pulse width controller to generate a secondary AC voltage vcdThe waveform of (2).
Preferably, S is adjusted5-S6Has a duty cycle of 50%, S1And S5Between them produce a phase shift angle
Figure BDA00034932324200000311
Is S5Hysteresis S1The phase shift angle δ of (1) is the pulse width of the switching tube S4.
Preferably, the primary alternating voltage v is determined from the midpoint by steady state analysisabAnd a secondary alternating voltage vcdObtaining the resonant current i from the waveform diagramLThe waveform of (2).
The invention provides a voltage balance control method of a full-half-bridge resonant converter, which adopts a fundamental wave approximation method to perform steady-state analysis due to the resonant operation of the converter:
obtaining an equivalent circuit diagram of the converter in a phasor domain FHA by the circuit structure of a full half-bridge resonant converter, wherein v is two voltage sources respectivelyabAnd vcdNormalized fundamental phasor of/n, obtaining vabAnd vcdNormalized phasor expression for/n:
Figure BDA0003493232420000031
Figure BDA0003493232420000032
Figure BDA0003493232420000033
is vabIn the form of a normalized vector representation of (c),
Figure BDA0003493232420000034
is vcdM is the voltage gain, in particular
Figure BDA0003493232420000035
Further, obtaining a voltage gain M of the converter according to the turn ratio of the transformer; according to the normalized switching frequency F ═ omegas/ωNQuality factor Q ═ omegaNLs/ZNObtaining the normalized impedance of the capacitor: QF-Q/F, where F is the normalized switching frequency, Q is the quality factor, L issThe sum of the external inductor and the leakage inductance of the transformer; omegasFor switching angular frequency omegaNIs a standard resonance angular frequency, in particular
Figure BDA0003493232420000036
ZNIs a standard load resistance, specifically ZN＝RL/n2，RLIs a load resistor;
and obtaining a normalized resonance current expression by using an equivalent circuit:
Figure BDA0003493232420000037
wherein
Figure BDA0003493232420000038
And IpAre respectively provided withIs the resonant current and vABPhase shift angle and peak current of;
further obtaining the normalized output power PpuAbout pulse width delta and phase shift angle
Figure BDA0003493232420000039
The expression of (c):
Figure BDA00034932324200000310
further analyzing the range of ZVS and obtaining the switch S1～S6Each corresponding to a ZVS condition.
Furthermore, when the positive half-cycle areas of the primary side fundamental voltage and the secondary side fundamental voltage are equal, the voltages are considered to be balanced; the relational expression between δ and M can be obtained by calculation:
Figure BDA0003493232420000041
further, when M is 0.5, the primary side full bridge operates in the half bridge state, S3Is always on and S4Is always turned off, at this time vabAnd vcdThe positive half-cycle area of/n is equal; when M is equal to 1, the primary side works in a full-bridge state, the voltage balance control modulation strategy is equivalent to the traditional single phase-shift control, v is equal toabAnd vcdThe positive half-cycle area of/n is equal; when 0.5<M<1, the primary side operates in an intermediate state between the full bridge and the half bridge, vabAnd vcdThe/n positive half-cycle area remains equal.
Compared with the prior art, the invention has the following remarkable advantages:
(1) compared with the existing double-bridge resonant converter, the full-half-bridge resonant converter topological structure provided by the invention reduces the number of switching tubes and effectively reduces the cost.
(2) The invention uses the resonant circuit, has the advantages of low switching loss, easy control, approximate sine wave current and the like, and has higher efficiency compared with a non-resonant converter.
(3) A voltage balance control strategy is provided, so that the converter can realize the voltage balance of the primary side and the secondary side under a wide voltage range, soft switching of all switch tubes is guaranteed, and the efficiency of the converter is improved.
(4) The modulation strategy of the invention does not need phase shift in the bridge, thereby improving the power transmission capability.
Drawings
FIG. 1 is a schematic diagram of a full half-bridge resonant converter;
FIG. 2 shows a combination of switch S1～S6Control method, by controlling switch S1～S6A generated voltage waveform diagram and a generated output current waveform diagram;
fig. 3 is an equivalent circuit of a full half-bridge resonant converter in the phasor domain FHA;
FIG. 4 is a diagram of an implementation of a voltage balancing control strategy;
FIG. 5 is a graph showing the relationship between V and Vin＝75V，VoutWhen M is 1 and P is 200W, V is 100Vab、vcd、iLWaveform and each switching tube current;
FIG. 6 shows the equation Vin＝125V，VoutWhen M is 0.6 and P is 200W, V is 100Vab、vcd、iLWaveform and each switching tube current;
FIG. 7 shows the equation Vin＝150V，VoutWhen M is 0.5 and P is 200W, V is 100Vab、vcd、iLWaveform and switching tube current.
Detailed Description
The technical solutions in the examples of the present invention are clearly and completely described below with reference to the drawings in the examples of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments of the present invention without inventive step, are within the scope of the present invention.
The present invention will be described in further detail with reference to the accompanying drawings.
Example 1
A full half-bridge resonant converter is schematically shown in FIG. 1, where VinAnd VoutInput voltage and output voltage, iLAnd ioRespectively input and output currents, CpIs a resonant capacitor, Co1、Co2Is a voltage-sharing capacitor, LsIs the sum of external inductor and leakage inductor of transformer S1～S4Switching elements of primary side, S5～S6The 6 switching elements are secondary side switching elements each formed by a body diode (d)S1～dS6) And parasitic capacitance (C)S1～CS6) And n is the transformer transformation ratio.
Firstly, the pulse width of each switch is adjusted to obtain a gating signal scheme for a high-pulse-width controller, and then a midpoint alternating voltage V is generatedabA waveform diagram of (a). As shown in fig. 2, switch S1And S2The duty cycle of (2) is 50%. Switch S4Is reduced to δ, S3The pulse width increases to 2 pi-delta. Thus, a three-level PWM voltage waveform is generated. Since the width of the positive pulse is delta and the width of the negative pulse is always equal to pi. Regulating S5～S6Has a duty cycle of 50% and generates a symmetrical square-wave signal vcd。S1And S5Between them produce a phase shift angle
Figure BDA0003493232420000051
By varying the width delta and phase-shift angle of the positive pulse
Figure BDA0003493232420000052
The normalized resonance current i can be controlledL,NAnd normalized power Ppu。
By vabAnd vcdThe waveform diagram is subjected to steady-state analysis to obtain a resonance current iLThe waveform of (2). In order to obtain phasor expressions for respective correlation quantities by changing the positive pulse width in correspondence with the phase shift angle, since the resonance current approximates to a sine wave, a Fundamental Harmonic Approximation (FHA) is used for steady-state analysis. For the sake of convenience in the art,all the quantities are normalized according to a base value, and an FHA equivalent circuit diagram of the converter in a phasor domain is obtained through the circuit structure of the full-half-bridge series resonant converter. Switch S1And switch S5With a phase delay therebetween, i.e., the phase angle. In one cycle, switch S5Closing, wherein the pulse width is pi; switch S6Closed, the pulse width is also pi. Thus, a secondary alternating voltage v is generatedcdThe waveform of (2).
Since the resonance current is approximate to sine wave, the steady state analysis can be performed by adopting a Fundamental Harmonic Approximation (FHA) method, and the normalized resonance current i can be obtained by performing the steady state analysis on the oscillogramLIs described in (1).
For convenience, all formulas are normalized by normalization to a base value (V)NIs a standard voltage, ZNIs a standard load resistance, omegaNStandard resonance angular frequency):
VN＝Vin
ZN＝RL/n2
Figure BDA0003493232420000061
FIG. 3 shows the FHA equivalent circuit of the converter in the phasor domain, where the two voltage sources are vabAnd vcdNormalized fundamental phasor of/n, can be obtained:
Figure BDA0003493232420000062
Figure BDA0003493232420000063
then, the voltage gain M of the converter is obtained according to the turns ratio of the transformer
M＝0.5Vout/(nVin)
According to the normalized switching frequency F ═ omegas/ωNSwitching angular frequencyRate omegasQuality factor Q ═ omegaNLt/ZNThe normalized impedance of the resonator can be found:
QF-Q/F
by using the equivalent circuit, a normalized resonance current expression i can be obtainedL,N(t)：
iL,N(t)＝IPcos(ωSt+Φi)
Wherein the phase angle phiiAnd peak current IpThe method comprises the following steps:
Figure BDA0003493232420000064
Figure BDA0003493232420000065
according to current IpEffective value vABThe effective voltage value can be calculated to obtain a normalized output power expression:
Figure BDA0003493232420000066
then, according to the definition of ZVS, when the current value passing through the opening moment of the switching tube is negative, the realization of ZVS is indicated. From FIG. 2, the ZVS condition for each switch can be derived
Figure BDA0003493232420000071
Further in accordance with fig. 4, a voltage balance control strategy (VBT) is implemented to equalize the positive half-cycles of the primary-side fundamental voltage and the secondary-side fundamental voltage, as expressed below:
Figure BDA0003493232420000072
further, simplifying it, we get the expression as follows:
Figure BDA0003493232420000073
from the range of δ being [0, π ], we can get cos δ between [ -1,1], thus the range of values for M is as follows:
1/2≤M≤1
by this point, voltage balancing has been completed and the relationship of M and δ and the range of M is obtained.
So far, the sum of delta under the voltage balance control strategy can be calculated according to the normalized power and different values of M
Figure BDA0003493232420000074
The value of (a).
Simulations were performed according to the designed input, output and power, at which time soft switching in the full power range can be achieved for all switching tubes.
To verify the correctness of the theory, a simulation test is performed in the PSIM:
(1) when V isin＝75V，VoutWhen M is 1 and P is 200W, V is 100Vab、vcd、iLThe waveforms and the switching tube currents are shown in fig. 5;
(2) when V isin＝125V，VoutWhen M is 0.6 and P is 200W, V is 100Vab、vcd、iLThe waveforms and the switching tube currents are shown in fig. 6;
(3) when V isin＝150V，VoutWhen M is 0.5 and P is 200W, V is 100Vab、vcd、iLThe waveforms and the currents of the switching tubes are shown in FIG. 7;
after the simulation waveform verification is combined, the theory is found to be consistent with the reality, and the feasibility of the method is proved.
The above description is only for the preferred embodiment of the present invention, but the scope of the present invention is not limited thereto, and any changes or substitutions that can be easily conceived by those skilled in the art within the technical scope of the present invention are included in the scope of the present invention. Therefore, the protection scope of the present invention shall be subject to the protection scope of the claims.

Claims (10)

1. A full half-bridge resonant converter, characterized by: comprises a full bridge at the primary side, a half bridge at the secondary side, and a resonant capacitor CpResonant inductor LsThe turn ratio is 1: n high-frequency transformer Tr;
wherein the primary side full-bridge comprises a switch tube S1～S4Body diode ds1～ds4Parasitic capacitance Cs1～Cs4(ii) a The secondary side half-bridge comprises a switching tube S5～S6Body diode ds5～ds6Parasitic capacitance Cs5～Cs6Voltage equalizing capacitor Co1And Co2；
VinAnd VoutInput voltage and output voltage, iLAnd ioRespectively a resonant current and an output current;
and the positive half-cycle area of the primary side fundamental wave voltage and the positive half-cycle area of the secondary side fundamental wave voltage are adjusted to be equal, so that the voltage is balanced.
2. The full half-bridge resonant converter of claim 1, wherein: by setting S1～S4The pulse widths of the four switches regulate the gate control signal of the pulse width controller to generate a three-level PWM voltage waveform.
3. The full half-bridge resonant converter of claim 2, wherein: switch S1And S2The duty cycle of (2) is 50%; switch S4Is reduced to δ, S3The pulse width is increased to 2 pi-delta, switch S1And S4The on-time of (c) is synchronized.
4. Full half-bridge resonant converter according to claim 2, characterized in that: by setting S5～S6Pulse width controller for two switchesGenerating a secondary ac voltage vcdThe waveform of (2).
5. Full half-bridge resonant converter according to claim 4, characterized in that: regulating S5-S6Has a duty cycle of 50%, S1And S5Between them produce a phase shift angle
Figure FDA0003493232410000013
Figure FDA0003493232410000014
Is S5Hysteresis S1The phase shift angle δ of (1) is the pulse width of the switching tube S4.
6. Full half-bridge resonant converter according to claim 4, characterized in that: by steady state analysis, from the midpoint primary AC voltage vabAnd a secondary alternating voltage vcdObtaining the resonant current i from the waveform diagramLThe waveform of (2).
7. A method of voltage balancing control using a full half-bridge resonant converter according to any of claims 1 to 6, characterized by using a fundamental approximation for steady state analysis:
obtaining an equivalent circuit diagram of the converter in a phasor domain FHA by the circuit structure of a full half-bridge resonant converter, wherein v is two voltage sources respectivelyabAnd vcdNormalized fundamental phasor of/n, obtaining vabAnd vcdNormalized phasor expression for/n:
Figure FDA0003493232410000011
Figure FDA0003493232410000012
Figure FDA0003493232410000021
is vabIn the form of a normalized vector representation of (c),
Figure FDA0003493232410000022
is vcdM is the voltage gain, in particular
Figure FDA0003493232410000023
8. The voltage balance control method according to claim 7, wherein a voltage gain M of the converter is obtained from a turns ratio of the transformer; according to the normalized switching frequency F ═ omegas/ωNQuality factor Q ═ omegaNLs/ZNObtaining the normalized impedance of the capacitor: QF-Q/F, where F is the normalized switching frequency, Q is the quality factor, L issThe sum of the external inductor and the leakage inductance of the transformer; omegasFor switching angular frequency omegaNOmega is the standard resonance angular frequency, in particular
Figure FDA0003493232410000024
ZNIs a standard load resistance, specifically ZN＝RL/n2，RLIs a load resistor;
and obtaining a normalized resonance current expression by using an equivalent circuit:
Figure FDA0003493232410000025
wherein
Figure FDA0003493232410000026
And IpRespectively, the resonant current and vABPhase shift angle and peak current of;
further obtaining the normalized output power PpuAbout pulse width delta and shiftPhase angle
Figure FDA0003493232410000027
Expression (c):
Figure FDA0003493232410000028
further analyzing the range of ZVS and obtaining the switch S1～S6Each corresponding to a ZVS condition.
9. The voltage balance control method according to claim 8, characterized in that:
when the positive half-cycle areas of the primary side fundamental voltage and the secondary side fundamental voltage are equal, the voltages are considered to be balanced; the relational expression between δ and M can be obtained by calculation:
Figure FDA0003493232410000029
10. the voltage balance control method according to claim 7 or 9, characterized in that: when M is 0.5, the primary side full bridge works in the half bridge state, S3Is always on and S4Is always turned off, at this time vabAnd vcdThe positive half-cycle area of/n is equal; when M is equal to 1, the primary side works in a full-bridge state, the voltage balance control modulation strategy is equivalent to the traditional single phase-shift control, v is equal toabAnd vcdThe positive half-cycle area of/n is equal; when 0.5<M<1, the primary side operates in an intermediate state between the full bridge and the half bridge, vabAnd vcdThe/n positive half-cycle area remains equal.
CN202210104033.5A 2022-01-28 2022-01-28 Full-half-bridge resonant converter and voltage balance control method thereof Pending CN114465489A (en)

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN115955122A (en) * 2022-12-27 2023-04-11 常熟理工学院 Backflow-free modulation method and system for double-bridge series resonant converter

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN115955122A (en) * 2022-12-27 2023-04-11 常熟理工学院 Backflow-free modulation method and system for double-bridge series resonant converter
CN115955122B (en) * 2022-12-27 2023-11-03 常熟理工学院 Reflux-free modulation method and system for double-bridge series resonant converter

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