WO2009049501A1 - Procédé et dispositif de commande de moteur utilisant une modulation de largeur d'impulsion à vecteur spatial - Google Patents

Procédé et dispositif de commande de moteur utilisant une modulation de largeur d'impulsion à vecteur spatial Download PDF

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Publication number
WO2009049501A1
WO2009049501A1 PCT/CN2008/070145 CN2008070145W WO2009049501A1 WO 2009049501 A1 WO2009049501 A1 WO 2009049501A1 CN 2008070145 W CN2008070145 W CN 2008070145W WO 2009049501 A1 WO2009049501 A1 WO 2009049501A1
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WIPO (PCT)
Prior art keywords
phase
motor
current
value
carrier cycle
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PCT/CN2008/070145
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English (en)
Chinese (zh)
Inventor
Meijuan Xie
Weiyi Lin
Yunzhou Fang
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Chery Automobile Co., Ltd.
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Publication of WO2009049501A1 publication Critical patent/WO2009049501A1/fr

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control

Definitions

  • the present invention relates to motor control techniques, and more particularly to a motor control method and apparatus employing space vector pulse width modulation.
  • the following describes the method of space vector control by taking a three-phase bridge voltage type inverter as an example.
  • the basic function of the three-phase bridge voltage type inverter is to convert the DC bus voltage Ud into a three-phase AC voltage for driving a three-phase motor, and the rotating magnetic field generated by the alternating three-phase AC voltage causes the motor to rotate at a certain speed. .
  • FIG. 1 shows the basic circuit structure of a three-phase bridge voltage type inverter.
  • the inverter mainly comprises a three-phase motor having a three-phase winding Z, and the three-phase winding has three pairs of bridge arms, respectively labeled as A phase, B phase, and C phase; each pair of bridge arms includes an upper arm and a lower bridge.
  • the arm and each arm are controlled by the controllable high-power switching device to control the opening and closing of the bridge arm.
  • the midpoints a, b and c of the upper and lower arms of each phase are connected to one end of the corresponding phase winding of the motor, and the motor phase windings
  • the other end point is the common junction n of the three-phase winding.
  • the DC bus voltage U d can be converted into an AC voltage having a certain frequency, which flows into the three-phase motor to rotate the three-phase motor.
  • the DC bus voltage Ud is expressed as two Ud/2, and the midpoint is zero.
  • the vector control method is used to control the three-phase motor, that is, each arm of the inverter is controlled to be turned on according to a certain frequency and sequence; in order to indicate different working states of the inverter, the on-off state of each bridge arm is used in a three-dimensional manner.
  • Space vector to represent Since the upper arm and the lower arm of each pair of bridge arms cannot be simultaneously turned on, the three-dimensional space vector is sufficient to indicate the working state of all the bridge arms, and further indicates the working state of the inverter, which indicates the inverter.
  • the three-dimensional space vector of the working state of the device is called a voltage space vector.
  • the motion of the three-dimensional space vector corresponding to the three-phase windings of A, B, and C is represented on the three-dimensional coordinates, and the three coordinate axes of the three-dimensional coordinates are respectively A, B corresponding to the three phases of A, B, and C,
  • the C-axis is 120 degrees from each other on the graph.
  • the voltage vectors corresponding to the respective axes are ⁇ , u b , and u c , respectively .
  • the three-phase inverter will have a total of eight working states, and the eight working states include six effective vectors.
  • S VPWM space vector pulse width modulation
  • SPWM space vector pulse width modulation
  • the use of space vector theory to control the inverter can reduce the switching times of switching devices by one-third, and the DC voltage utilization rate is increased by one percent. Fifteen, can get better harmonic suppression effect, and easy to achieve digital control.
  • the space vector pulse width modulation method provided by the prior art directly uses the switching state hexagon based on the calculation of the space vector modulation signal in FIG. 2, which requires complicated online sine function and inverse tangent function operation, resulting in The computational complexity is large, and its complex algorithm has a negligible impact on high-precision real-time control.
  • some methods for simplifying the space vector pulse width modulation have also appeared. For example, the method of patent number US6, 819, 078 B2 "SPACE VECTOR PWM MODULATOR FOR PERMANENT MAGNET MOTOR DRIVE", although Some improvements are conventional, but since the modulation based on the hexagonal state of the switch state is still inevitable, the modulation steps are inevitably complicated and complicated.
  • the space vector pulse width modulation software is implemented based on a single chip microcomputer or a digital signal processor (DSP). Many instructions need to be executed. The code length, especially the execution time of software instructions, cannot meet the high performance control system in some applications. Design requirements.
  • the switching state of the power transistor is usually driven by an interrupt.
  • the microprocessor or DSP considering the execution delay time of the CPU and the code execution time in the interrupt, the above-mentioned voltage space vector control requires a higher performance microprocessor or DSP. The above problems cause the cost of the space vector control device to be increased, and the high-performance real-time control requirements cannot be met.
  • the technical problem to be solved by the present invention is to provide a motor control method and apparatus using space vector pulse width modulation to simplify the prior art space vector pulse width modulation calculation process and meet the requirements of high performance real time control.
  • the present invention provides a motor control method using space vector pulse width modulation, including:
  • a PWM driving signal corresponding to the upper and lower arms of each phase of the inverter is generated, and the on and off of the respective arms of the inverter in the current carrier cycle are controlled.
  • the method of calculating the SPWM vector modulated signal comprises the steps of:
  • the calculating the sinusoidal pulse width modulation SPWM vector modulation signal of each phase of the current carrier cycle is:
  • the motor operating state detection data and the command data are obtained by triggering an interrupt update sampling value at a midpoint of the carrier cycle.
  • the motor running state detection data includes: a motor rotor position angle, a motor rotor angular velocity, and a motor current; the motor rotor position angle is detected by a rotor position detector to obtain a rotor rotor position angle, and the motor rotor angular velocity is according to the adjacent The rotor position angle difference is obtained by dividing the sampling time.
  • ⁇ The zero sequence ⁇ * is recorded by the following formula
  • M * - :max ⁇ *, M *, M *)-(l- :).min ⁇ *, M *, M *)+(2 :-l);
  • ⁇ is a zero-order component;
  • ⁇ is The SPWM vector modulation signal of each phase in the carrier cycle;
  • K is a constant greater than or equal to zero and less than or equal to 1.
  • the obtaining a duty ratio of each phase in the carrier period according to the SVPWM vector modulation signal is specifically calculated by using the following formula:
  • ⁇ ⁇ ⁇ is the on-time of each phase
  • M d is the DC bus voltage, and is the sampling period
  • T s is the corresponding phase The duty Than.
  • the generating according to the determined duty ratio of each phase in the current carrier cycle, generating a PWM control signal corresponding to each phase, specifically: counting a carrier period by using a counting register, where the counting register value is a time base period The value stored in the register TBPRD; a carrier cycle includes the value of the count register increasing from 0 to the value saved by TBPRD, and then counting down from the value held by TBPRD to 0 symmetrical process; calculating according to the duty ratio of each phase A count compares the value of register A, where the calculated formula is:
  • the invention also provides a motor control device using space vector pulse width modulation, comprising: an SPWM vector modulation signal calculation unit, configured to receive motor running state detection data and instruction data obtained in a current carrier cycle, and calculate correspondingly according to the calculation SPWM vector modulated signal of carrier period;
  • An SVPWM vector modulation signal calculation unit configured to receive an SPWM vector modulation signal output by the SPWM vector modulation signal calculation unit, and add a zero sequence component corresponding to a carrier period to the vector modulation signal to obtain an SVPWM vector modulation in the carrier period Signal
  • a duty ratio calculation unit configured to receive the output of the SVPWM vector modulation signal calculation unit
  • the SVPWM vector modulates the signal, and calculates a duty ratio of each phase in the current carrier cycle according to the SVPWM vector modulation signal;
  • a PWM control signal generating unit configured to receive a duty ratio of each phase in a current carrier period output by the duty ratio calculating unit, and accordingly generate a PWM control signal corresponding to each phase duty ratio in the carrier period;
  • the SPWM vector modulation signal calculation unit includes:
  • the current command value determining subunit is configured to receive the motor rotor angular velocity of the current carrier cycle obtained by the detection, and the current motor torque command value, and obtain the current command value i d x of the synchronous rotating coordinate system d-axis and the q-axis through the motor characteristic table. , i q x ;
  • a fixed/synchronous coordinate converter configured to receive a motor current detection value of a current carrier cycle, and a rotor position detection value, and calculate an actual current value i d , iq of the d-axis and the q-axis of the synchronous rotating coordinate system according to the above value;
  • a current controller for receiving the current command value /, of the d-axis and the q-axis, and the actual current value i q of the d-axis and the q-axis, and calculating the d-axis voltage of the synchronous rotating coordinate system in combination with the motor rotor angular velocity obtained by the detection Command value and q-axis voltage command value V d *; V*;
  • a synchronous or fixed coordinate converter for receiving the synchronous rotating coordinate system d-axis voltage command value and the q-axis voltage command value, and converting the same into a three-phase voltage command value output in a stationary coordinate system; the stationary coordinate system
  • the lower three-phase voltage command value is the desired SPWM vector modulation signal.
  • the motor rotor angular velocity, the motor current, and the current motor torque command value are all sampled at the midpoint of the carrier cycle.
  • a duty ratio calculation unit is obtained by calculation using the following equation ;; h and the phases in the current duty cycle of the carrier cycle:
  • the PWM control signal generating unit includes:
  • the counting register is configured to perform frequency counting on each carrier cycle to implement time measurement of the carrier cycle;
  • the register value of the counting register is a value saved by TBPRD, and the timing of one carrier cycle includes counting from 0 to TBPRD. Value, then two symmetrical processes that count down from the value held by TBPRD to 0;
  • the comparison value calculation unit is configured to receive each phase duty ratio in the current carrier cycle output by the duty ratio calculation unit, and calculate a comparison value according to each phase duty ratio; the specific calculation formula is:
  • CMPA TBPRD * J1 / 2 -
  • comparison result output unit configured to receive the comparison value of each phase output by the comparison value calculation unit, and compare the comparison value with the current count value of the count register, and generate the result according to the comparison result
  • the rotor position detector is a resolver or a Hall position sensor.
  • the method and the device provided by the invention add the zero-sequence component of the corresponding carrier period to the SPWM vector modulation signal of each carrier cycle, that is, obtain the SVPWM vector modulation signal, and the method for obtaining the SVPWM signal has fewer steps and is simple to calculate.
  • Real-time control of the motor can be realized with a cheaper control chip.
  • the prior art uses an SVPWM vector modulation signal based on a switch state hexagon to calculate each carrier cycle, and the calculation method requires calculation using a plurality of trigonometric functions. The process is complicated, if the control chip is used, the control chip will not achieve good real-time control effect.
  • the method and device provided by the invention simplify the calculation process of the SVPWM vector modulation signal, thereby reducing the requirement of the control chip required for implementing the SVPWM control, and expanding the application range of the SVPWM vector modulation signal motor control mode. .
  • 1 is a basic circuit structure of a three-phase bridge voltage type inverter in the prior art
  • FIG. 3 is a flow chart of a motor control method using space vector pulse width modulation according to a first embodiment of the present invention
  • FIG. 4 is a flow chart showing a typical method for calculating an SPWM vector modulated signal w a *, u b M c * in the present invention
  • FIG. 5 is a schematic diagram of the space vector required for combining the effective vector and the zero vector in the present invention
  • FIG. 6 is a method for obtaining a required space vector using the SVPWM modulated signal in the present invention
  • FIG. 7 is a timing chart of the carrier period in the present invention. Schematic diagram of implementation principle
  • Figure 8 is a schematic view showing the control of the three-phase bridge arm on and off using the comparative value CMPA in the present invention
  • Figure 9 is a block diagram showing the structure of the second embodiment of the present invention.
  • the first embodiment of the present invention provides a motor control method using space vector pulse width modulation, which is used to provide a PWM control signal to the three-phase bridge inverter shown in Fig. 1.
  • the PWM control signal is provided to enable the three-phase bridge inverter to output three-phase alternating current, which obtains a circular rotating magnetic field of a desired speed on the stator of the motor, so that the rotor of the motor outputs a corresponding speed.
  • the control method provided by this embodiment is as follows: First, the three-phase vector modulation signal Ma *, u b u c ⁇ is calculated according to the current motor running detection data and the command data to the three-phase vector modulation signal Ma *, u b Me * zero sequence component was added, to obtain the cycle carrier-phase SVPWM modulation signal u * u b * u **; according to the three-phase SVPWM modulation signal w a **, u * is obtained in the respective phases of the carrier cycle Duty cycle; according to the obtained duty cycle of each phase, each arm of the control inverter is turned on and off in turn.
  • the method of implementing the above various steps will be specifically described below.
  • FIG. 3 illustrates a space vector pulse width modulation according to a first embodiment of the present invention.
  • Flowchart of the motor control method The following is a detailed introduction with the figure.
  • the motor driven in this embodiment is a three-phase 7jC magnetic synchronous motor.
  • Step S301 calculating, according to the motor running state detection data and the command data of the current carrier cycle, each phase SPWM vector modulation signal w a *, u b u c corresponding to the carrier cycle.
  • the vector modulation signal SPWM w a *, u b M c * and the motor control requirements of the state detector is obtained by calculation obtained.
  • the prior art has provided a variety of specific calculation methods, and a typical calculation method is briefly described below.
  • FIG. 4 there is shown a flow diagram of a typical method of calculating SPWM vector modulated signals w a *, u b u c *.
  • step 401 the three-phase current, the DC bus voltage, and the rotor running speed of the motor are detected.
  • the three-phase current of the motor is obtained by detecting a current sensor mounted on any two phases of the motor. Since the sum of the currents flowing into the same node is zero, the current value of the other phase can be calculated according to the two-phase current obtained by the detection.
  • the DC bus voltage is obtained by detecting a DC bus by a voltage sensor.
  • the rotor running speed is obtained by detecting the rotor position angle of the adjacent sampling time and then calculating it.
  • the rotor position angle can be obtained by a resolver or Hall element detection.
  • the rotor angular difference value of the adjacent sampling interval is divided by the sampling time to obtain the rotor angular velocity ⁇ ; the above calculation formula is expressed as follows: , where 0 represents the rotor position obtained at the current sampling time K detection
  • 0 7 represents the rotor position angle obtained at the previous sampling time (K-1); it is the sampling interval.
  • Step 402 Receive a command value for the motor torque.
  • the motor torque command value is a motor torque command value determined according to the demand of the load, and the motor torque command value is related to the magnitude of the external load and the rotational speed requirement of the motor, and is calculated according to the basic torque formula.
  • Step 403 According to the rotor position angle of the motor, the rotor angular velocity, and the motor torque command value, the current command value of the d-axis and the q-axis is obtained by providing a maximum torque characteristic table by the unit current of the motor, that is, id i q
  • the d-axis and the q-axis are coordinate axes of the synchronously rotated coordinate system of the transformed motor, and the transformation process
  • the motor stationary coordinate system is transformed into a synchronous synchronous coordinate system of the motor, and the static three-axis coordinate is transformed into two-axis coordinates, which is called 3/2 transformation or fixed/synchronous coordinate transformation.
  • the motor torque command value reflects the expectation of the motor torque
  • the rotor position angle and the rotor angular speed of the motor reflect the actual running condition of the motor.
  • the current required for the d-axis and the q-axis when the motor is operated as needed can be known.
  • the current is the current command values i d x , iq x of the d-axis and the q-axis.
  • Step S404 using the current detection value of the step S401, calculating an actual current value of the synchronous rotating coordinate system.
  • the three-phase current obtained by the detection can be converted into the actual current value of the coordinate axis on the synchronous rotating coordinate system, that is, the d-axis current value i d and the q-axis on the synchronous rotating coordinate system.
  • Step S405 Calculate the d-axis voltage command value and the q-axis voltage command value according to the calculation results of the above steps.
  • step S404 obtains the d-axis current value q-axis current value iq on the synchronous rotating coordinate system, the value
  • the actual current values of the d-axis and the q-axis are expressed, and based on the above results, the d-axis voltage command value M and the q-axis voltage command value M can be calculated, and the above values represent expected values for the d-axis and q-axis voltages.
  • the specific calculation method is as follows:
  • the d-axis voltage vector is the product of the PI control output value of the pair / i d minus the motor pole pair, the rotor angular velocity ", the q-axis inductance ⁇ and the q-axis voltage vector u is the pair i and
  • Step S406 converting the d-axis voltage command value M and the q-axis voltage command value in the synchronous rotating coordinate system into a three-phase voltage command value in a stationary coordinate system: u a u h u * , that is, SPWM three-phase vector modulation
  • the signal command value that is, the SPWM vector modulation signal.
  • the transformation process of this step is the inverse of the above 3/2 transformation process, called 2/3 transformation, or synchronous or fixed coordinate transformation.
  • the above method of obtaining the SPWM vector modulation signals M a *, M e * has existed in the prior art, and there are various ways in the prior art to obtain the above SPWM vector modulation based on the speed or torque command value and the detected rotor operating speed.
  • the method of the signals w a *, u b M c * since no special improvement is made in this process in the present invention, the above process will not be described in detail herein.
  • a set of SPWM vector modulated signals w a *, u b u c can be obtained by the prior art or even various methods that may be generated in the future.
  • CTR (Counter value) 0 is used.
  • Step 302 Add a zero sequence component corresponding to the carrier period to the SPWM vector modulation signals u a *, u b *, u e *, respectively, to obtain each phase SVPWM vector modulation signal Ma **, u b in the carrier cycle. * c .
  • the three-phase vector modulation signals M a *, u b M e * are three-phase vector modulated signals obtained according to the SPWM principle, the number of switchings controlled using the above three-phase vector modulated signals is more than that of using SVPWM three-phase vector modulated signals.
  • the utilization of the DC voltage is correspondingly low, and more harmonic components are generated.
  • Purpose of this example is the three-phase vector modulation signals * ⁇ * into three-phase space vector, the specific method used is the three-phase vector modulation signal M a A zero sequence component is added to M C * to obtain an SVPWM vector modulation signal corresponding to the SPWM vector modulation signal.
  • PWM modulation a vector required for synthesis, such as the space vector shown in Fig. 2, is obtained.
  • FIGS. 5 and 2 For the space vector shown in Fig.
  • the above-mentioned action time is symmetrically distributed on the carrier period T s , that is, a time distribution diagram corresponding to the effective vector 2 and the zero vector ⁇ sum shown in FIG. 5 is obtained, and the horizontal axis of the graph represents the effective vector, 2 and zero vectors ⁇
  • the space vector can be synthesized; the above-mentioned effective vector, u 2 and zero vector ⁇ /.
  • the action time of ⁇ and ⁇ needs to be implemented to the A, B, C three-phase conduction time to achieve control.
  • the action time T ⁇ . 7 is shown symmetrically in Figure 5, which is for ease of calculation. The above is obtained by theoretical derivation
  • the three-phase conduction of A, B, and C required for the inter-vector ab it can be seen that the three-phase conduction time of A, B, and C can be expressed as follows:
  • Fig. 6 shows a method of obtaining the above-described space vector using the SVPWM three-phase modulation signals C/ a **, C/ b **, C/ c **. Due to the carrier period; very short, the voltage value of C/ a **, U b * in the period can be considered to be fixed, that is, the line parallel to the horizontal axis shown in Fig. 6, the above SVPWM three-phase modulation signal U *
  • the action time ⁇ ⁇ , T b , 7 of U b * C/ c ** is the action time of the three phases A, B and C calculated according to formula (3).
  • K is a constant greater than or equal to 0 and less than or equal to 1, and the constant can be taken as needed.
  • the value of ⁇ is often taken as 0.5 to obtain the effect of simplifying the equation (6).
  • Step S303 obtaining duty ratios of the phases in the current carrier cycle according to the SVPWM three-phase modulation signals C/ a **, U*U*.
  • step S303 the current SVPWM three-phase modulation signals C/ a **, U b * U * have been obtained and brought into the formula (4), and the duty ratios of the three phases A, B, and C can be obtained:
  • the duty cycle is the duty cycle corresponding to one carrier cycle.
  • Step S304 ⁇ is determined by the duty ratio of each phase, and a PWM control signal corresponding to each phase is generated.
  • FIG. 7 shows the implementation principle of timing the carrier period. Since it is necessary to control the duty ratio of the on-time of each phase in one carrier cycle, it is necessary to have a time unit capable of measuring the carrier cycle, and specifically, the clock frequency of the control system can be used.
  • the carrier frequency of the PWM signal is ⁇ , that is, one carrier period is ⁇ , and the carrier period can be timed by the clock frequency of the control system of 100 Mhz.
  • the minimum count time step T TBCLK value is 0. 01 ⁇ 8 , that is, one carrier cycle contains 10000 time units.
  • TBPRD a register value of TBPRD
  • Fig. 8 shows a method of controlling the on and off of a three-phase bridge arm using the comparison value CMPA. In conjunction with the figure, the phase A is taken as an example to illustrate the control process of the phase.
  • the design number is directly CTR.
  • CTR 0, the current carrier cycle starts. At this time, CTR ⁇ CMPA, PWM control signal & keeps low.
  • CTR CMPA, and the count register is in the up counting phase, PWM is generated.
  • Step S305 generating a driving signal for the upper and lower arms of each phase of the inverter according to the PWM control signal.
  • the above process causes the PM control signal output to obtain the required duty cycle in the carrier cycle, most preferably into the space vector required in the carrier cycle.
  • the process of implementing the duty cycle control is a relatively simple implementation in the prior art. In fact, other methods can be used to obtain the required duty cycle. Those skilled in the art can perform the steps according to this step. Control the requirements, design other forms of control, so that the PWM control signal The duty cycle output.
  • a second embodiment of the present invention provides a motor control apparatus that implements space vector pulse width modulation.
  • FIG. 9 there is shown a block diagram of the unit composition of the second embodiment of the present invention.
  • the motor control apparatus using space vector pulse width modulation includes an SPWM vector modulation signal calculation unit 91, an SVPWM vector modulation signal calculation unit 92, a duty ratio calculation unit 93, a PWM control signal generation unit 94, a drive signal generation unit 95, and a rotor. Position detector 96 and speed calculator 97.
  • the figure also shows the controlled object motor 90, and the inverter 98.
  • the motor 90 is specifically a 7 j synchronous electrode.
  • the SPWM vector modulation signal calculation unit 91 is configured to receive motor running state detection data and command data obtained in a current carrier cycle, and calculate an SPWM vector modulation signal corresponding to a carrier cycle accordingly. Since there are various calculation methods for calculating the SPWM vector modulation signal in the prior art, different calculation methods may require different motor state detection parameters, and thus the motor running state detection data is different according to different SPWM vector modulation signals, generally
  • the motor running state detection data mainly includes data such as a motor rotor position angle, a motor rotor angular speed, and a motor current. As shown in FIG. 9, the rotor position detector 96 is used to detect the rotor position angle of the motor.
  • the rotor position detector 96 generally uses a spin-on transformer or a Hall position sensor, and the direct detection result is the sine value of the rotor position angle.
  • the cosine value can be obtained by trigonometric function calculation and the rotor position angle can be obtained.
  • the speed calculator 97 receives the rotor position angle detection result output from the rotor position detector 96, and calculates the motor rotor angular speed ⁇ using the formula ⁇ , which is explained in the first embodiment.
  • the current of the two phases of the three-phase input of the motor 90 can be detected by various methods, as shown in FIG. , i c .
  • the SPWM vector modulation signal calculation unit 91 includes a current command value determination subunit 911, a fixed/synchronous coordinate converter 912, a current controller 913, and a synchronous or fixed coordinate transformation subunit 914.
  • the current command value determining subunit 911 is configured to receive the motor rotor angular velocity ⁇ of the current carrier cycle obtained by the detection, and the current motor torque command value ⁇ , and the maximum torque characteristic table can be provided by the unit current of the motor 90 to obtain synchronous rotation.
  • the motor torque command value comes from the demand of the main control unit for the motor torque, and the command value determines the working demand for the motor.
  • the unit current can provide a maximum torque characteristic table which is a data table reflecting the characteristics of the motor, and each permanent magnet synchronous motor has a corresponding data table.
  • the fixed/synchronous coordinate converter 912 is configured to receive a motor current detection value of a current carrier cycle, and use the current detection value, and the rotor position detector 96 detects the rotor position angle of the motor, and obtains a synchronous rotation coordinate system d.
  • a current controller 913 configured to receive current command values /, i q of the d-axis and q-axis and actual current values i d , iq of the d-axis and the q-axis, and calculate a synchronous rotating coordinate system d-axis voltage command accordingly Value and q-axis voltage command value ⁇ , ⁇ M , d-axis voltage vector is the product of the PI control output value of the pair/and minus the motor pole pair, the rotor angular velocity, and the q-axis inductance; the q-axis voltage vector u is the pair The sum of the PI control output value of i and the product of the motor pole pair, rotor angular velocity, q-axis inductance and id, and the motor pole pair P, ⁇ , permanent magnet flux m.
  • the synchronous or fixed coordinate converter 914 is configured to receive the synchronous rotating coordinate system d-axis voltage command value and the q-axis voltage command value M / u q and convert it into a three-phase voltage command value output M in a stationary coordinate system.
  • the three-phase voltage command value ⁇ ⁇ *, ' M C * in the stationary coordinate system is the desired SPWM vector modulation signal.
  • a vector modulation signal SPWM 91 receives the output from the vector modulation signal SPWM calculation unit ⁇ , ⁇ u b u should be added to the carrier cycle c to vector modulation signal Obtaining the SVPWM vector modulation signal ⁇ *, ⁇ u b * u c * in the carrier period.
  • the value calculated by the SPWM vector modulation signal calculation unit 91 is brought into the above zero sequence component calculation formula to obtain zero for the carrier cycle.
  • Order component Since the SPWM vector modulation signal u of the current carrier cycle; u b M c * depends on the calculation result of the detection signal obtained by sampling in the current carrier cycle, the sampling of the current carrier cycle has not been performed just after entering the carrier cycle, so actually, The SPWM vector modulation signal ⁇ , ⁇ u b u * used for the above calculation is performed using the sample value obtained in the previous carrier cycle. Since the state of the motor in the adjacent carrier cycle does not change much, the result of such calculation can satisfy the demand. ,
  • the duty ratio calculation unit 93 is configured to receive the SVPWM vector modulation signal calculation unit
  • the output SVPWM vector modulated signal u a ** / u b * u * and the duty cycle of each phase in the current carrier cycle is calculated according to the SVPWM vector modulation signal ⁇ *, ⁇ u b * O.
  • the duty ratio of each phase can be calculated according to the following formula: ' a. , b, c:
  • Nf + represents the SVPWM three-phase modulation signal at the sampling time.
  • the PWM control signal generating unit 94 is configured to receive the duty ratios of the phases in the current carrier cycle output by the duty ratio calculating unit 93, and generate PWM control signals &, S h corresponding to the respective phases according to the PWM control signals S c .
  • the unit includes a count register 941, a comparison value calculation unit 942, and a comparison result output unit 943.
  • the counting register 941 is configured to perform frequency division counting on each carrier cycle to implement time measurement of the carrier cycle. There are many ways to time meter the carrier cycle. A typical approach is provided below.
  • the register value of the count register 941 is set to a value saved by TBPRD. From the time of entering the carrier cycle, the counter is incremented from 0 to the value saved by TBPRD, and then subtracted from the value saved by TBPRD. Counting to 0, just ending a carrier cycle. During this timing, one carrier cycle includes two symmetric counting processes.
  • the counter unit of the counter generally adopts the minimum clock frequency of the system. After the minimum clock frequency is determined, the value saved by the time base period register TBPRD can be calculated according to the value of the carrier period;
  • the comparison value calculation unit 942 is configured to receive each phase duty ratio in the current carrier cycle output by the duty ratio calculation unit 93; 7 / a, b, c , and calculate corresponding to each phase duty ratio
  • the comparison value of the phase is as follows: ⁇ ⁇ t / 2.
  • the comparison result output unit 943 is configured to receive the comparison values of the phases output by the comparison value calculation unit 942, compare the comparison value with the current count value of the count register 941, and generate corresponding PWM control according to the comparison result. signal.
  • Corresponding to each phase is &, S b , S c .
  • the comparison value of the corresponding A phase duty is the comparison comparison register A (CMPA, A counter compare register).
  • CMPA comparison comparison register
  • the driving signal generating unit 95 receives the PWM control signal generating unit 94 to output a PWM control signal corresponding to each phase, thereby generating complementary two PWM driving signals corresponding to the respective phases, respectively driving the upper and lower arms of the phase.
  • the PWM drive signal includes PWM a , PWM a Down, PWM b , PWM b , PWM C , PWM C , corresponding to the upper and lower arms of A, B, C three phases.
  • the dead zone is not set, the upper and lower arms of each phase are complementarily turned on.
  • the dead zone setting is adopted, the lower arm has a delay or a complementary stagnation of the upper arm.
  • the motor control device provided in this embodiment can be implemented by using a DSP TMS320f2808 chip. It should be noted that the register value and the value saved by TBPRD in the present invention are generally equivalent.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

La présente invention concerne un procédé et un dispositif de commande de moteur utilisant une modulation de largeur d'impulsion à vecteur spatial. Le procédé comprend : le calcul d'un signal de modulation de vecteur SPWM de chaque phase dans la période de porteuse actuelle ; l'ajout du signal de modulation de vecteur SPWM de chaque phase au composant de séquence zéro correspondant à la période de porteuse de manière à obtenir un signal de modulation de vecteur SVPWM de chaque phase dans la période de porteuse ; l'obtention du rapport de charge de chaque phase dans la période de porteuse actuelle en fonction du signal de modulation de vecteur SVPWM de chaque phase ; la génération d'un signal de commande PWM de chaque phase conformément au rapport de charge déterminé de chaque phase dans la période de porteuse actuelle ; enfin la génération de signaux d'entraînement PWM pour les bras supérieur et inférieur correspondants de chaque phase d'un inverseur en fonction du signal de commande PWM de manière à commander la marche et l'arrêt de chaque bras de l'inverseur dans la période de porteuse actuelle.
PCT/CN2008/070145 2007-10-19 2008-01-21 Procédé et dispositif de commande de moteur utilisant une modulation de largeur d'impulsion à vecteur spatial WO2009049501A1 (fr)

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