WO2009049501A1

WO2009049501A1 - A motor control method and device by using space vector pulse width modulation

Info

Publication number: WO2009049501A1
Application number: PCT/CN2008/070145
Authority: WO
Inventors: Meijuan Xie; Weiyi Lin; Yunzhou Fang
Original assignee: Chery Automobile Co., Ltd.
Priority date: 2007-10-19
Filing date: 2008-01-21
Publication date: 2009-04-23
Also published as: CN101174811A; CN101174811B

Abstract

A motor control method and device by using space vector pulse width modulation. The method comprises computing a SPWM vector modulation signal of each phase in the current carrier period; adding the SPWM vector modulation signal of each phase to the zero sequence component corresponding to the carrier period in order to obtain a SVPWM vector modulation signal of each phase in the carrier period; obtaining duty ratio of each phase in the current carrier period according to the SVPWM vector modulation signal of each phase; generating a PWM control signal of each phase according to the determined duty ratio of each phase in the current carrier period; and generating PWM drive signals for the corresponding upper and lower arms of each phase of an inverter according to the PWM control signal in order to control on and off of each arm of the inverter in the current carrier period.

Description

A motor control method and device using space vector pulse width modulation. The present application claims to be submitted to the Chinese Patent Office on October 19, 2007, the application number is 200710163379.8, and the invention name is "a motor control method using space vector pulse width modulation". The priority of the Chinese Patent Application, the entire disclosure of which is incorporated herein by reference. Technical field

The present invention relates to motor control techniques, and more particularly to a motor control method and apparatus employing space vector pulse width modulation.

Background technique

In recent years, the space vector theory of electric motors has been introduced into inverters and their control. The basic principle is to use the different combinations of inverter bridge switch control signals to make the output trajectory of the output voltage space vector of the inverter. As close as possible to the circle, use this output voltage to power the motor so that the motor runs smoothly at the required speed.

The following describes the method of space vector control by taking a three-phase bridge voltage type inverter as an example. The basic function of the three-phase bridge voltage type inverter is to convert the DC bus voltage Ud into a three-phase AC voltage for driving a three-phase motor, and the rotating magnetic field generated by the alternating three-phase AC voltage causes the motor to rotate at a certain speed. .

Please refer to FIG. 1, which shows the basic circuit structure of a three-phase bridge voltage type inverter. The inverter mainly comprises a three-phase motor having a three-phase winding Z, and the three-phase winding has three pairs of bridge arms, respectively labeled as A phase, B phase, and C phase; each pair of bridge arms includes an upper arm and a lower bridge. The arm and each arm are controlled by the controllable high-power switching device to control the opening and closing of the bridge arm. The midpoints a, b and c of the upper and lower arms of each phase are connected to one end of the corresponding phase winding of the motor, and the motor phase windings The other end point is the common junction n of the three-phase winding. By controlling the on and off of the six bridge arms, the DC bus voltage U _d can be converted into an AC voltage having a certain frequency, which flows into the three-phase motor to rotate the three-phase motor. In the figure, the DC bus voltage Ud is expressed as two Ud/2, and the midpoint is zero.

The vector control method is used to control the three-phase motor, that is, each arm of the inverter is controlled to be turned on according to a certain frequency and sequence; in order to indicate different working states of the inverter, the on-off state of each bridge arm is used in a three-dimensional manner. Space vector to represent. Since the upper arm and the lower arm of each pair of bridge arms cannot be simultaneously turned on, the three-dimensional space vector is sufficient to indicate the working state of all the bridge arms, and further indicates the working state of the inverter, which indicates the inverter. The three-dimensional space vector of the working state of the device is called a voltage space vector. In the voltage space vector, "1" is used to indicate the conduction of the upper arm, and the lower arm is closed; using "0" The upper arm is terminated and the lower arm is turned on; the respective components of the three-dimensional space vector respectively represent the bridge arms corresponding to the three phases of the motor, B, and C. Thus, in the process of performing AC inverter, the working process of the inverter can be expressed as a space vector hexagon as shown in FIG. 2.

In FIG. 2, the motion of the three-dimensional space vector corresponding to the three-phase windings of A, B, and C is represented on the three-dimensional coordinates, and the three coordinate axes of the three-dimensional coordinates are respectively A, B corresponding to the three phases of A, B, and C, The C-axis is 120 degrees from each other on the graph. The voltage vectors corresponding to the respective axes are ^, u _b , and u _{c , respectively} . The three-phase inverter will have a total of eight working states, and the eight working states include six effective vectors.

( ) and two zero vectors ( ^{U o} , ^{U 7} ), we can see the six effective vectors of Figure 2.

(U - U e ) forms a switch state space hexagon. In the vector space defined above, an arbitrary space vector can be represented by the voltage vector corresponding to each axis:

9

U _r = —( ua + e ^π u _b + e ^] " ^π U c )

³ ( 1 ) performing space vector pulse width modulation (S VPWM ), which is a series of rotation vectors obtained by the above vector expression and varying at a certain rate according to the switch state hexagon, the rotation vector The end point forms a circle, and the actual effect is to form a circular rotating magnetic field on the motor stator to stabilize the motor at the required rate. Compared with the traditional Sine pulse width modulation (SPWM), the use of space vector theory to control the inverter can reduce the switching times of switching devices by one-third, and the DC voltage utilization rate is increased by one percent. Fifteen, can get better harmonic suppression effect, and easy to achieve digital control.

The space vector pulse width modulation method provided by the prior art directly uses the switching state hexagon based on the calculation of the space vector modulation signal in FIG. 2, which requires complicated online sine function and inverse tangent function operation, resulting in The computational complexity is large, and its complex algorithm has a negligible impact on high-precision real-time control. At present, some methods for simplifying the space vector pulse width modulation have also appeared. For example, the method of patent number US6, 819, 078 B2 "SPACE VECTOR PWM MODULATOR FOR PERMANENT MAGNET MOTOR DRIVE", although Some improvements are conventional, but since the modulation based on the hexagonal state of the switch state is still inevitable, the modulation steps are inevitably complicated and complicated.

Usually, the space vector pulse width modulation software is implemented based on a single chip microcomputer or a digital signal processor (DSP). Many instructions need to be executed. The code length, especially the execution time of software instructions, cannot meet the high performance control system in some applications. Design requirements. The switching state of the power transistor is usually driven by an interrupt. In the microprocessor or DSP, considering the execution delay time of the CPU and the code execution time in the interrupt, the above-mentioned voltage space vector control requires a higher performance microprocessor or DSP. The above problems cause the cost of the space vector control device to be increased, and the high-performance real-time control requirements cannot be met.

Summary of the invention

In view of the above drawbacks, the technical problem to be solved by the present invention is to provide a motor control method and apparatus using space vector pulse width modulation to simplify the prior art space vector pulse width modulation calculation process and meet the requirements of high performance real time control.

In order to solve the above technical problem, the present invention provides a motor control method using space vector pulse width modulation, including:

Calculating each phase of the current carrier cycle SPWM vector modulation signal;

Adding each phase SPWM vector modulation signal to a zero sequence component corresponding to a carrier period, and obtaining each phase SVPWM vector modulation signal in the carrier period;

Obtaining a duty ratio of each phase in a current carrier cycle according to each phase SVPWM vector modulation signal; generating a PWM control signal corresponding to each phase according to the determined duty ratio of each phase in the current carrier cycle;

According to the PWM control signal, a PWM driving signal corresponding to the upper and lower arms of each phase of the inverter is generated, and the on and off of the respective arms of the inverter in the current carrier cycle are controlled.

Preferably, the method of calculating the SPWM vector modulated signal comprises the steps of:

Detecting the three-phase current, the DC bus voltage, and the rotor running speed of the motor in the current carrier cycle, and calculating the rotor angular velocity of the motor according to the rotor operating speed;

Receiving the current command value for the motor torque;

Obtaining current command values i _d ^x , iq ^x of the d-axis and q-axis of the synchronous rotating coordinate system by the motor characteristic table according to the rotor angular velocity and the motor torque command value; Using the current carrier cycle motor current detection value, calculating the actual current values i _d , iq of the synchronous rotating coordinate system d-axis and q-axis;

Calculating the synchronous rotating coordinate system d-axis voltage command value and the q-axis voltage command value according to the current command values i _d ^x , iq ^{x of the} d-axis and the q-axis and the actual current values i _d , i _q of the d-axis and the q-axis;

Converting the d-axis voltage command value and the q-axis voltage command value in the synchronous rotating coordinate system into a three-phase voltage command value in a stationary coordinate system; and the three-phase voltage command value is the SPWM vector modulated signal of each phase .

Preferably, the calculating the sinusoidal pulse width modulation SPWM vector modulation signal of each phase of the current carrier cycle is:

Calculating each phase sinusoidal pulse width modulation SPWM vector modulation signal of the current carrier cycle according to the motor running state detection data and the command data receiving the current carrier cycle;

The motor operating state detection data and the command data are obtained by triggering an interrupt update sampling value at a midpoint of the carrier cycle.

Preferably, the motor running state detection data includes: a motor rotor position angle, a motor rotor angular velocity, and a motor current; the motor rotor position angle is detected by a rotor position detector to obtain a rotor rotor position angle, and the motor rotor angular velocity is according to the adjacent The rotor position angle difference is obtained by dividing the sampling time. ^ The zero sequence ^ * is recorded by the following formula

_M * =- :max^*, _M *, _M *)-(l- :).min^*, _M *, _M *)+(2 :-l); where ^ is a zero-order component; ^ is The SPWM vector modulation signal of each phase in the carrier cycle; K is a constant greater than or equal to zero and less than or equal to 1.

Preferably, the obtaining a duty ratio of each phase in the carrier period according to the SVPWM vector modulation signal is specifically calculated by using the following formula:

**

T ⁺⁰ - ⁵ U _d

Where Τ _α ^ is the on-time of each phase, M _d is the DC bus voltage, and is the sampling period; the on-time Τ _α of each phase in one sampling period, divided by the sampling period T _{s is} the corresponding phase The duty Than.

Preferably, the generating, according to the determined duty ratio of each phase in the current carrier cycle, generating a PWM control signal corresponding to each phase, specifically: counting a carrier period by using a counting register, where the counting register value is a time base period The value stored in the register TBPRD; a carrier cycle includes the value of the count register increasing from 0 to the value saved by TBPRD, and then counting down from the value held by TBPRD to 0 symmetrical process; calculating according to the duty ratio of each phase A count compares the value of register A, where the calculated formula is:

CMPA 2 TBPRD " II ; When the value of the count register CTR = 0, start the current carrier cycle, when CTR < CMPA, the phase PWM control signal remains low; when CTR = CMPA, and the count register is incremented In the phase, the phase PWM control signal transitions to a high level output; when CTR = CMPA, and the count register is in the down counting phase, the phase PWM control signal transitions to a low level output.

The invention also provides a motor control device using space vector pulse width modulation, comprising: an SPWM vector modulation signal calculation unit, configured to receive motor running state detection data and instruction data obtained in a current carrier cycle, and calculate correspondingly according to the calculation SPWM vector modulated signal of carrier period;

An SVPWM vector modulation signal calculation unit, configured to receive an SPWM vector modulation signal output by the SPWM vector modulation signal calculation unit, and add a zero sequence component corresponding to a carrier period to the vector modulation signal to obtain an SVPWM vector modulation in the carrier period Signal

a duty ratio calculation unit, configured to receive the output of the SVPWM vector modulation signal calculation unit

The SVPWM vector modulates the signal, and calculates a duty ratio of each phase in the current carrier cycle according to the SVPWM vector modulation signal;

a PWM control signal generating unit, configured to receive a duty ratio of each phase in a current carrier period output by the duty ratio calculating unit, and accordingly generate a PWM control signal corresponding to each phase duty ratio in the carrier period;

a driving signal generating unit, configured to receive a PWM control signal output by the PWM control signal generating unit, and generate two driving signals according to the PWM control signals of the respective phases, and the driving signal is output to the upper and lower bridges of each phase of the inverter The arm controls the turning on and off of each arm of the inverter. Preferably, the SPWM vector modulation signal calculation unit includes:

The current command value determining subunit is configured to receive the motor rotor angular velocity of the current carrier cycle obtained by the detection, and the current motor torque command value, and obtain the current command value i _d ^{x of the} synchronous rotating coordinate system d-axis and the q-axis through the motor characteristic table. , i _q ^x ;

a fixed/synchronous coordinate converter, configured to receive a motor current detection value of a current carrier cycle, and a rotor position detection value, and calculate an actual current value i _d , iq of the d-axis and the q-axis of the synchronous rotating coordinate system according to the above value;

a current controller for receiving the current command value /, of the d-axis and the q-axis, and the actual current value i _{q of the} d-axis and the q-axis, and calculating the d-axis voltage of the synchronous rotating coordinate system in combination with the motor rotor angular velocity obtained by the detection Command value and q-axis voltage command value V _d *; V*;

a synchronous or fixed coordinate converter for receiving the synchronous rotating coordinate system d-axis voltage command value and the q-axis voltage command value, and converting the same into a three-phase voltage command value output in a stationary coordinate system; the stationary coordinate system The lower three-phase voltage command value is the desired SPWM vector modulation signal.

Preferably, the motor rotor angular velocity, the motor current, and the current motor torque command value are all sampled at the midpoint of the carrier cycle.

Preferably, the zero sequence component of the current carrier cycle used by the SVPWM vector modulation signal calculation unit is specifically obtained by: u] = -km _a x{ _u , _u , _u l)-(lk)*mm{ _u , _u , _u l) + (2k - l); wherein ^ is the zero sequence component; is the SPWM vector modulation signal of each phase in the carrier cycle; K is a constant greater than or equal to zero and less than or equal to 1.

Preferably, a duty ratio calculation unit is obtained by calculation using the following equation ;; _h and the phases in the current duty cycle of the carrier cycle:

Τ ^{+ 0} - ⁵ U _d;

7" TT _d ' where Τ. „ is the conduction time of each phase, M _d is the DC bus voltage; is the sampling period; 3⁄4^ That is, the duty ratio corresponding to each phase; 7 _be . Preferably, the PWM control signal generating unit includes:

The counting register is configured to perform frequency counting on each carrier cycle to implement time measurement of the carrier cycle; the register value of the counting register is a value saved by TBPRD, and the timing of one carrier cycle includes counting from 0 to TBPRD. Value, then two symmetrical processes that count down from the value held by TBPRD to 0;

The comparison value calculation unit is configured to receive each phase duty ratio in the current carrier cycle output by the duty ratio calculation unit, and calculate a comparison value according to each phase duty ratio; the specific calculation formula is:

CMPA = TBPRD * J1 / 2 -

/ a, b, c ' comparison result output unit, configured to receive the comparison value of each phase output by the comparison value calculation unit, and compare the comparison value with the current count value of the count register, and generate the result according to the comparison result Different output signals are used as PWM control signals; when CTR = 0, 3⁄4 new carrier period, when CTR<CMPA, the PM control signal output by the unit is low; when CTR = CMPA, and the counting register is increasing In the counting phase, the PWM control signal corresponding to the phase output of the unit jumps to a high level; when CTR = CMPA, and the counting register is in the down counting phase, the PWM control signal corresponding to the phase output of the unit jumps to low power. level.

Preferably, the electrical line state detection data includes a motor rotor position angle, a motor rotor angular velocity, and a motor current; the motor rotor position angle is detected by a rotor position detector to obtain a rotor rotor position angle, and the motor rotor angular velocity is determined according to The detected value of the adjacent rotor position angle is obtained by the following formula: _ω = ^ .

Δί

Preferably, the rotor position detector is a resolver or a Hall position sensor.

The method and the device provided by the invention add the zero-sequence component of the corresponding carrier period to the SPWM vector modulation signal of each carrier cycle, that is, obtain the SVPWM vector modulation signal, and the method for obtaining the SVPWM signal has fewer steps and is simple to calculate. Real-time control of the motor can be realized with a cheaper control chip. In contrast, the prior art uses an SVPWM vector modulation signal based on a switch state hexagon to calculate each carrier cycle, and the calculation method requires calculation using a plurality of trigonometric functions. The process is complicated, if the control chip is used, the control chip will not achieve good real-time control effect.

In summary, the method and device provided by the invention simplify the calculation process of the SVPWM vector modulation signal, thereby reducing the requirement of the control chip required for implementing the SVPWM control, and expanding the application range of the SVPWM vector modulation signal motor control mode. .

DRAWINGS

1 is a basic circuit structure of a three-phase bridge voltage type inverter in the prior art;

2 is a space vector hexagon formed by an operating state of an inverter in the prior art;

3 is a flow chart of a motor control method using space vector pulse width modulation according to a first embodiment of the present invention;

4 is a flow chart showing a typical method for calculating an SPWM vector modulated signal w _a *, u _b M _c * in the present invention;

5 is a schematic diagram of the space vector required for combining the effective vector and the zero vector in the present invention; FIG. 6 is a method for obtaining a required space vector using the SVPWM modulated signal in the present invention; FIG. 7 is a timing chart of the carrier period in the present invention. Schematic diagram of implementation principle;

Figure 8 is a schematic view showing the control of the three-phase bridge arm on and off using the comparative value CMPA in the present invention; and Figure 9 is a block diagram showing the structure of the second embodiment of the present invention.

detailed description

The first embodiment of the present invention provides a motor control method using space vector pulse width modulation, which is used to provide a PWM control signal to the three-phase bridge inverter shown in Fig. 1. The PWM control signal is provided to enable the three-phase bridge inverter to output three-phase alternating current, which obtains a circular rotating magnetic field of a desired speed on the stator of the motor, so that the rotor of the motor outputs a corresponding speed.

The specific steps of the control method provided by this embodiment are as follows: First, the three-phase vector modulation signal _Ma *, u _b u _c \ is calculated according to the current motor running detection data and the command data to the three-phase vector modulation signal _Ma *, u _{b Me} * zero sequence component was added, to obtain the cycle carrier-phase SVPWM modulation signal u * u _b * u **; according to the three-phase SVPWM modulation signal w _a **, u * is obtained in the respective phases of the carrier cycle Duty cycle; according to the obtained duty cycle of each phase, each arm of the control inverter is turned on and off in turn. The method of implementing the above various steps will be specifically described below.

Please refer to FIG. 3, which illustrates a space vector pulse width modulation according to a first embodiment of the present invention. Flowchart of the motor control method. The following is a detailed introduction with the figure. The motor driven in this embodiment is a three-phase 7jC magnetic synchronous motor.

Step S301, calculating, according to the motor running state detection data and the command data of the current carrier cycle, each phase SPWM vector modulation signal w _a *, u _b u _c corresponding to the carrier cycle.

The vector modulation signal SPWM _{_{_{w a *, u b M c}}} * and the motor control requirements of the state detector is obtained by calculation obtained. The prior art has provided a variety of specific calculation methods, and a typical calculation method is briefly described below.

Referring to Figure 4, there is shown a flow diagram of a typical method of calculating SPWM vector modulated signals w _a *, u _b u _c *.

In step 401, the three-phase current, the DC bus voltage, and the rotor running speed of the motor are detected. The three-phase current of the motor is obtained by detecting a current sensor mounted on any two phases of the motor. Since the sum of the currents flowing into the same node is zero, the current value of the other phase can be calculated according to the two-phase current obtained by the detection. The DC bus voltage is obtained by detecting a DC bus by a voltage sensor. The rotor running speed is obtained by detecting the rotor position angle of the adjacent sampling time and then calculating it. The rotor position angle can be obtained by a resolver or Hall element detection. After setting the speed sampling frequency, the rotor angular difference value of the adjacent sampling interval is divided by the sampling time to obtain the rotor angular velocity ω; the above calculation formula is expressed as follows: , where 0 represents the rotor position obtained at the current sampling time K detection

Angle; 0 7 represents the rotor position angle obtained at the previous sampling time (K-1); it is the sampling interval.

Step 402: Receive a command value for the motor torque.

The motor torque command value is a motor torque command value determined according to the demand of the load, and the motor torque command value is related to the magnitude of the external load and the rotational speed requirement of the motor, and is calculated according to the basic torque formula.

Step 403: According to the rotor position angle of the motor, the rotor angular velocity, and the motor torque command value, the current command value of the d-axis and the q-axis is obtained by providing a maximum torque characteristic table by the unit current of the motor, that is, id i _q

The d-axis and the q-axis are coordinate axes of the synchronously rotated coordinate system of the transformed motor, and the transformation process The motor stationary coordinate system is transformed into a synchronous synchronous coordinate system of the motor, and the static three-axis coordinate is transformed into two-axis coordinates, which is called 3/2 transformation or fixed/synchronous coordinate transformation. Since the motor torque command value reflects the expectation of the motor torque, the rotor position angle and the rotor angular speed of the motor reflect the actual running condition of the motor. According to the above two values, the current required for the d-axis and the q-axis when the motor is operated as needed can be known. The current is the current command values i _d ^x , iq ^{x of} the d-axis and the q-axis. The above calculations and coordinate transformation processes are well known to those skilled in the art and will not be described in detail herein.

Step S404, using the current detection value of the step S401, calculating an actual current value of the synchronous rotating coordinate system.

Through the 3/2 transformation process mentioned in the previous step, the three-phase current obtained by the detection can be converted into the actual current value of the coordinate axis on the synchronous rotating coordinate system, that is, the d-axis current value i _d and the q-axis on the synchronous rotating coordinate system. Current value iq.

Step S405: Calculate the d-axis voltage command value and the q-axis voltage command value according to the calculation results of the above steps.

Since the current command value /, iq ^x is obtained in step S403, the value represents an expected value for the d-axis and the q-axis current; and step S404 obtains the d-axis current value q-axis current value iq on the synchronous rotating coordinate system, the value The actual current values of the d-axis and the q-axis are expressed, and based on the above results, the d-axis voltage command value M and the q-axis voltage command value M can be calculated, and the above values represent expected values for the d-axis and q-axis voltages. The specific calculation method is as follows: The d-axis voltage vector is the product of the PI control output value of the pair / i _d minus the motor pole pair, the rotor angular velocity ", the q-axis inductance ^ and the q-axis voltage vector u is the pair i and The sum of the PI control output value and the motor pole pair, the rotor angular velocity, the q-axis inductance, and the product of the motor pole pair, ω, and the permanent magnet flux ^ ^ . Step S406, converting the d-axis voltage command value M and the q-axis voltage command value in the synchronous rotating coordinate system into a three-phase voltage command value in _a stationary coordinate system: u _a u _h u * , that is, SPWM three-phase vector modulation The signal command value, that is, the SPWM vector modulation signal.

The transformation process of this step is the inverse of the above 3/2 transformation process, called 2/3 transformation, or synchronous or fixed coordinate transformation. The above method of obtaining the SPWM vector modulation signals M _a *, M _e * has existed in the prior art, and there are various ways in the prior art to obtain the above SPWM vector modulation based on the speed or torque command value and the detected rotor operating speed. The method of the signals w _a *, u _b M _c *, since no special improvement is made in this process in the present invention, the above process will not be described in detail herein. In summary, a set of SPWM vector modulated signals w _a *, u _b u _c can be obtained by the prior art or even various methods that may be generated in the future.

In the step of calculating the SPWM vector modulation signals w _a *, u _b M _e * in the above steps, for the convenience of subsequent processing, the modulus of the detected data is triggered when the start time of the carrier cycle, that is, CTR (Counter value) = 0 is used. At the midpoint, the value stored in the 00=Time base module period register (TMPRD) triggers the PWM interrupt, and the sampled value after the analog-to-digital conversion is updated in the interrupt, thereby obtaining the detected sample data.

Step 302: Add a zero sequence component corresponding to the carrier period to the SPWM vector modulation signals u _a *, u _b *, u _e *, respectively, to obtain each phase SVPWM vector modulation signal _Ma **, u _{b in} the carrier cycle. * c .

Since the three-phase vector modulation signals M _a *, u _b M _e * are three-phase vector modulated signals obtained according to the SPWM principle, the number of switchings controlled using the above three-phase vector modulated signals is more than that of using SVPWM three-phase vector modulated signals. The utilization of the DC voltage is correspondingly low, and more harmonic components are generated. Purpose of this example is the three-phase vector modulation signals * ^ * into three-phase space vector, the specific method used is the three-phase vector modulation signal M _a

A zero sequence component is added to M _C * to obtain an SVPWM vector modulation signal corresponding to the SPWM vector modulation signal. The zero sequence component is: u] = -km^ii^[, _u l, _u l)-(lk)*mmi^[, _u l, _u l)+(2k-l). The principle of adopting this method will be described in detail below. First of all, the prior art is to adopt the switch state hexagon shown in FIG. 2, and determine the sector according to the vector instruction, and determine the on-time of each phase, that is, the duty ratio, through a series of calculations according to the sector information. Thus, by PWM modulation, a vector required for synthesis, such as the space vector shown in Fig. 2, is obtained. The above calculation process will be described below with reference to FIGS. 5 and 2. For the space vector shown in Fig. 2, it is located in the first quadrant, and the vector in this quadrant is synthesized by the effective vector, ₂ , and the zero vector ^ and the sum are also added. Therefore, in this carrier cycle, there are vectors such as the effective vector, u ₂ and zero vector ^ and ^;, where the zero vector ^ and optionally are selected. According to the principle of the switch state hexagon, it can be calculated that the valid, and zero vector ^ / in the carrier period. And ^ respective time T Τ. _Oh . The effective vector of the above action time, U ₂ and zero vector ^ /. And eventually can be synthesized into a space vector ^. The above-mentioned action time is symmetrically distributed on the carrier period T _s , that is, a time distribution diagram corresponding to the effective vector ₂ and the zero vector ^ sum shown in FIG. 5 is obtained, and the horizontal axis of the graph represents the effective vector, ₂ and zero vectors ^ The action time of ^ and ^;, since each vector is synthesized by the three-phase conduction time of A, B, C, the corresponding three-phase conduction time of A, B, C is plotted. In fact, the specific controllable quantity in the circuit is whether the B and C phases are turned on. Therefore, the space vector ^^ required for the carrier period is obtained. First, the effective vector and the zero vector ^/ are calculated. And the respective time; the appropriate action time of the carrier cycle, after the determination, the space vector can be synthesized; the above-mentioned effective vector, u ₂ and zero vector ^ /. The action time of ^ and ^; needs to be implemented to the A, B, C three-phase conduction time to achieve control. The above effective vector, and zero vector ^ /. And the action time T Γ. ₇ is shown symmetrically in Figure 5, which is for ease of calculation. The above is obtained by theoretical derivation The three-phase conduction of A, B, and C required for the inter-vector ab, it can be seen that the three-phase conduction time of A, B, and C can be expressed as follows:

The above equation shows the on-time and effective vector, ^/ ₂ and zero vector of the three phases A, B, and C. And the action time TT: Γ. ₇ relationship between. Fig. 6 shows a method of obtaining the above-described space vector using the SVPWM three-phase modulation signals C/ _a **, C/ _b **, C/ _c **. Due to the carrier period; very short, the voltage value of C/ _a **, U _b * in the period can be considered to be fixed, that is, the line parallel to the horizontal axis shown in Fig. 6, the above SVPWM three-phase modulation signal U * The action time Γ _α , T _b , 7 of U _b * C/ _c ** is the action time of the three phases A, B and C calculated according to formula (3).

According to the similar triangle in Figure 6, it can be known that the following formula holds:

c/ _a **, c/ _b **,

1⁄4.

Bring equation (3) into equation (4) and get

(2 1) ( ^{5 )}

The above formula (5) is obtained for the space vector ^ in Fig. 2, where M _z * is the zero sequence component corresponding to the space vector ^. The above derivation process, for any class on the switch hexagonal structure shown in FIG. For other space vectors like the space vector ^;, the result similar to equation (5) can be obtained by the above derivation. Corresponding to different zero-sequence components used by different spatial vectors, the zero-sequence component M _Z * can be expressed as a complete period

In the formula, K is a constant greater than or equal to 0 and less than or equal to 1, and the constant can be taken as needed. In practice, the value of Κ is often taken as 0.5 to obtain the effect of simplifying the equation (6).

Through the above derivation process, it can be known that only the corresponding zero sequence component ^ is added to _Ae , and the SPWM three-phase vector modulation signal command values t/ _a *, _Uh C/ _e * can be converted into corresponding SVPWM three-phase modulation signals. c/ _a **, c/ _b **, u; This method does not require complex triangulation as in the prior art, which saves the computation time of the controller and facilitates real-time control.

Step S303, obtaining duty ratios of the phases in the current carrier cycle according to the SVPWM three-phase modulation signals C/ _a **, U*U*.

By the step S303, the current SVPWM three-phase modulation signals C/ _a **, U _b * U * have been obtained and brought into the formula (4), and the duty ratios of the three phases A, B, and C can be obtained:

The duty cycle is the duty cycle corresponding to one carrier cycle.

Step S304, ^ is determined by the duty ratio of each phase, and a PWM control signal corresponding to each phase is generated.

After the duty ratio of each phase is determined, it is only necessary to output a PWM control signal according to the duty ratio to control the sequential turn-on and turn-off of the respective bridge arms of the inverter, so that SVPWM control can be realized. A method for realizing SVPWM control based on duty ratio is known in the prior art. One of the solutions is provided below. The scheme is based on the conduction of each phase of A, B, and C in one carrier cycle as shown in FIG. 5, that is, the duty ratio control is implemented in a center symmetric manner. The measurement of the length of time in the duty cycle is implemented by counting.

Please refer to FIG. 7, which shows the implementation principle of timing the carrier period. Since it is necessary to control the duty ratio of the on-time of each phase in one carrier cycle, it is necessary to have a time unit capable of measuring the carrier cycle, and specifically, the clock frequency of the control system can be used. In this embodiment The carrier frequency of the PWM signal is ΙΟΚΗζ, that is, one carrier period is ΙΟΟμε, and the carrier period can be timed by the clock frequency of the control system of 100 Mhz. At this time, the minimum count time step T _TBCLK value is _0. _{01 μ8} , that is, one carrier cycle contains 10000 time units. The carrier period is measured by counting the carrier period using a count register with a register value of TBPRD = 5000, incrementing from 0 to 5000, and then counting down to 5000 by 5000. In Fig. 7, an example in which eight pulse counts are used for the carrier cycle is shown.

In the above manner, according to the determined duty ratio of each phase in the carrier cycle, a comparison value corresponding to the duty ratio of each phase is stored in the comparison register, and the count value of the count register is compared with the comparison value. And determine the continuity of each phase based on the comparison result. Since the symmetrical waveform is generated by increasing or decreasing the value, the formula for calculating the comparative value is = βΤ^Ζ) *;; , ^{/ 2} . Fig. 8 shows a method of controlling the on and off of a three-phase bridge arm using the comparison value CMPA. In conjunction with the figure, the phase A is taken as an example to illustrate the control process of the phase.

The design number is directly CTR. When CTR = 0, the current carrier cycle starts. At this time, CTR<CMPA, PWM control signal & keeps low. When CTR = CMPA, and the count register is in the up counting phase, PWM is generated. The control signal & jumps to a high level output; when CTR = CMPA, and the count register is in the down counting phase, the PWM control signal & jumps to a low level output.

Step S305, generating a driving signal for the upper and lower arms of each phase of the inverter according to the PWM control signal.

Taking phase A as an example, when there is no dead time setting, the driving signal PWM _a and PWM _a are complementary actions; when & = 1, the PWM _a outputs a high level, and the A phase upper arm guide Pass, PWM _a output low level, the A phase lower arm is turned off; when & = 0, the PWM ^ output low level, the A phase upper arm is turned off, PWM _{a T} output is high Level, the A-phase lower arm is conducting; when the dead zone setting is adopted, the lower arm has a delay or a complementary lag with respect to the upper arm. In either case, make sure that the upper and lower arms are not turned on at the same time. The above process causes the PM control signal output to obtain the required duty cycle in the carrier cycle, most preferably into the space vector required in the carrier cycle.

The process of implementing the duty cycle control, that is, the PWM modulation, is a relatively simple implementation in the prior art. In fact, other methods can be used to obtain the required duty cycle. Those skilled in the art can perform the steps according to this step. Control the requirements, design other forms of control, so that the PWM control signal The duty cycle output.

Although the above embodiment is directed to a three-phase permanent magnet synchronous motor, substantially the same technical solution is applicable to other types of three-phase motors.

A second embodiment of the present invention provides a motor control apparatus that implements space vector pulse width modulation. Referring to Figure 9, there is shown a block diagram of the unit composition of the second embodiment of the present invention.

The motor control apparatus using space vector pulse width modulation includes an SPWM vector modulation signal calculation unit 91, an SVPWM vector modulation signal calculation unit 92, a duty ratio calculation unit 93, a PWM control signal generation unit 94, a drive signal generation unit 95, and a rotor. Position detector 96 and speed calculator 97. The figure also shows the controlled object motor 90, and the inverter 98. In this embodiment, the motor 90 is specifically a ₇ j synchronous electrode.

The SPWM vector modulation signal calculation unit 91 is configured to receive motor running state detection data and command data obtained in a current carrier cycle, and calculate an SPWM vector modulation signal corresponding to a carrier cycle accordingly. Since there are various calculation methods for calculating the SPWM vector modulation signal in the prior art, different calculation methods may require different motor state detection parameters, and thus the motor running state detection data is different according to different SPWM vector modulation signals, generally The motor running state detection data mainly includes data such as a motor rotor position angle, a motor rotor angular speed, and a motor current. As shown in FIG. 9, the rotor position detector 96 is used to detect the rotor position angle of the motor. The rotor position detector 96 generally uses a spin-on transformer or a Hall position sensor, and the direct detection result is the sine value of the rotor position angle. The cosine value can be obtained by trigonometric function calculation and the rotor position angle can be obtained. The speed calculator 97 receives the rotor position angle detection result output from the rotor position detector 96, and calculates the motor rotor angular speed ω using the formula ^, which is explained in the first embodiment. For the motor current, the current of the two phases of the three-phase input of the motor 90 can be detected by various methods, as shown in FIG. , i _c . Many specific current detecting methods have been provided in the prior art, for example, using an electric resistance method or using a current transformer or the like, and will not be described in detail herein. In the measurement. After the two-phase current, the current of the other phase is calculated by the relationship between the sum of the three-phase currents of the motor and zero. The above rotor position angle, motor current, etc. are obtained by using a certain frequency, wherein the optimal sampling frequency is taken as the carrier frequency, and the midpoint of the carrier period is taken as the sampling point, and the detection result obtained in one carrier period is calculated in the next carrier period. use. The SPWM vector modulation signal calculation unit 91 includes a current command value determination subunit 911, a fixed/synchronous coordinate converter 912, a current controller 913, and a synchronous or fixed coordinate transformation subunit 914.

The current command value determining subunit 911 is configured to receive the motor rotor angular velocity ω of the current carrier cycle obtained by the detection, and the current motor torque command value τ, and the maximum torque characteristic table can be provided by the unit current of the motor 90 to obtain synchronous rotation. The current command value of the d-axis and the q-axis of the coordinate system /, i _{q The} motor torque command value comes from the demand of the main control unit for the motor torque, and the command value determines the working demand for the motor. The unit current can provide a maximum torque characteristic table which is a data table reflecting the characteristics of the motor, and each permanent magnet synchronous motor has a corresponding data table.

The fixed/synchronous coordinate converter 912 is configured to receive a motor current detection value of a current carrier cycle, and use the current detection value, and the rotor position detector 96 detects the rotor position angle of the motor, and obtains a synchronous rotation coordinate system d. The actual current value of the axis and q axis ^ ^ and output.

a current controller 913, configured to receive current command values /, i _q of the d-axis and q-axis and actual current values i _d , iq of the d-axis and the q-axis, and calculate a synchronous rotating coordinate system d-axis voltage command accordingly Value and q-axis voltage command value Μ , · M , d-axis voltage vector is the product of the PI control output value of the pair/and minus the motor pole pair, the rotor angular velocity, and the q-axis inductance; the q-axis voltage vector u is the pair The sum of the PI control output value of i and the product of the motor pole pair, rotor angular velocity, q-axis inductance and id, and the motor pole pair P, ω, permanent magnet flux m.

The synchronous or fixed coordinate converter 914 is configured to receive the synchronous rotating coordinate system d-axis voltage command value and the q-axis voltage command value M / u _q and convert it into a three-phase voltage command value output M in a stationary coordinate system.

The three-phase voltage command value Μ _Α *, ' M _C * in the stationary coordinate system is the desired SPWM vector modulation signal.

Zero sequence component of the SVPWM modulation signal vector calculation unit 92, a vector modulation signal SPWM 91 receives the output from the vector modulation signal SPWM calculation unit Μ, · u _b u should be added to the carrier cycle _c to vector modulation signal Obtaining the SVPWM vector modulation signal 该 *, · u _b * u _c * in the carrier period. The zero sequence component of the current carrier period used by the SVPWM vector modulation signal calculation unit 92 is specifically obtained by the following equation: u] =

which is The zero sequence component; is the SPWM vector modulation signal of each phase in the carrier cycle; K is a constant greater than or equal to zero and less than or equal to 1. The zero sequence component is different with the carrier period. In each carrier cycle, the value calculated by the SPWM vector modulation signal calculation unit 91 is brought into the above zero sequence component calculation formula to obtain zero for the carrier cycle. Order component. Since the SPWM vector modulation signal u of the current carrier cycle; u _b M _c * depends on the calculation result of the detection signal obtained by sampling in the current carrier cycle, the sampling of the current carrier cycle has not been performed just after entering the carrier cycle, so actually, The SPWM vector modulation signal Μ , · u _b u * used for the above calculation is performed using the sample value obtained in the previous carrier cycle. Since the state of the motor in the adjacent carrier cycle does not change much, the result of such calculation can satisfy the demand. ,

The duty ratio calculation unit 93 is configured to receive the SVPWM vector modulation signal calculation unit

The output SVPWM vector modulated signal u _a ** / u _b * u * and the duty cycle of each phase in the current carrier cycle is calculated according to the SVPWM vector modulation signal Μ *, · u _b * O. According to the SVPWM vector signal, the duty ratio of each phase can be calculated according to the following formula: ' a. , b, c:

In the formula, Nf + represents the SVPWM three-phase modulation signal at the sampling time.

The PWM control signal generating unit 94 is configured to receive the duty ratios of the phases in the current carrier cycle output by the duty ratio calculating unit 93, and generate PWM control signals &, S _h corresponding to the respective phases according to the PWM control signals S _c . There are various implementation methods of the unit. In this embodiment, the unit includes a count register 941, a comparison value calculation unit 942, and a comparison result output unit 943.

The counting register 941 is configured to perform frequency division counting on each carrier cycle to implement time measurement of the carrier cycle. There are many ways to time meter the carrier cycle. A typical approach is provided below. The register value of the count register 941 is set to a value saved by TBPRD. From the time of entering the carrier cycle, the counter is incremented from 0 to the value saved by TBPRD, and then subtracted from the value saved by TBPRD. Counting to 0, just ending a carrier cycle. During this timing, one carrier cycle includes two symmetric counting processes. The counter unit of the counter generally adopts the minimum clock frequency of the system. After the minimum clock frequency is determined, the value saved by the time base period register TBPRD can be calculated according to the value of the carrier period;

The comparison value calculation unit 942 is configured to receive each phase duty ratio in the current carrier cycle output by the duty ratio calculation unit 93; 7 / a, b, c , and calculate corresponding to each phase duty ratio The comparison value of the phase is as follows: ΜΡΛ

η _t / 2. The comparison result output unit 943 is configured to receive the comparison values of the phases output by the comparison value calculation unit 942, compare the comparison value with the current count value of the count register 941, and generate corresponding PWM control according to the comparison result. signal. Corresponding to each phase is &, S _b , S _c .

The above comparison process is illustrated by an example of a specific comparison process. In a certain carrier cycle, the comparison value of the corresponding A phase duty is the comparison comparison register A (CMPA, A counter compare register). When the current count value CTR = 0, it indicates that the new carrier cycle is 3⁄4, at this time, CTR<CMPA, the A-phase PWM control signal & output of the unit is low, that is, & = 0; when CTR = CMPA, and the counting register is in the up counting phase, the unit corresponds to the PWM control signal output by the phase Jump to high level, that is, change from & = 0 to & = 1; When CTR = CMPA, and the count register is in the down counting phase, the unit's PWM control signal corresponding to the phase output jumps low, ie Change from & = 1 to & = 0 until the end of the carrier cycle. The above process can achieve a duty cycle of & = 1 in one carrier cycle. After the end of one carrier cycle, the next carrier cycle is entered, and the new CMPA value is calculated according to the duty cycle of each phase in the new carrier cycle, and the counting is restarted. This results in a complete motor control process from one carrier cycle to the next.

The driving signal generating unit 95 receives the PWM control signal generating unit 94 to output a PWM control signal corresponding to each phase, thereby generating complementary two PWM driving signals corresponding to the respective phases, respectively driving the upper and lower arms of the phase. As shown in Figure 9, the PWM drive signal includes PWM _a , PWM _a Down, PWM _b , PWM _b , PWM _C , PWM _C , corresponding to the upper and lower arms of A, B, C three phases. When the dead zone is not set, the upper and lower arms of each phase are complementarily turned on. Taking the phase A as an example, when there is no dead time setting, when the received PWM control signal & = 1, the PWM _a outputs a high level, so that the A-phase upper arm is turned on, and the PWM _{a T} outputs a low level, causing the A-phase lower arm to end. When the received PWM control signal & = 0, the PWM ^ outputs a low level, so that the A-phase upper arm is turned off, and the PWM _a outputs a high level, so that the A-phase lower arm is turned on. Of course, you can also interpret the & signal in reverse. When the dead zone setting is adopted, the lower arm has a delay or a complementary stagnation of the upper arm.

The motor control device provided in this embodiment can be implemented by using a DSP TMS320f2808 chip. It should be noted that the register value and the value saved by TBPRD in the present invention are generally equivalent.

The above is only a preferred embodiment of the present invention, and it should be noted that those skilled in the art can also make several improvements and retouchings without departing from the principles of the present invention. It should be considered as the scope of protection of the present invention.

Claims

Rights request

A motor control method using space vector pulse width modulation, comprising: calculating a sinusoidal pulse width modulation SPWM vector modulation signal of each phase of a current carrier cycle;

Adding each phase SPWM vector modulation signal to the zero sequence component of the carrier cycle to obtain each phase voltage space vector pulse width modulation SVPWM vector modulation signal in the carrier cycle;

Obtaining a duty ratio of each phase in a current carrier cycle according to each phase SVPWM vector modulation signal; generating a pulse width modulation PWM control signal corresponding to each phase according to the determined duty ratio of each phase in the current carrier cycle;

2. The method of claim 1 wherein calculating the SPWM vector modulated signal comprises the steps of:

Receiving the current command value for the motor torque;

Obtaining current command values i _d ^x , iq ^x of the d-axis and q-axis of the synchronous rotating coordinate system by the motor characteristic table according to the rotor angular velocity and the motor torque command value;

Using the current carrier cycle motor current detection value, calculating the actual current values i _d , iq of the synchronous rotating coordinate system d-axis and q-axis;

The d-axis voltage command value and the q-axis voltage command value are converted into a three-phase voltage command value in a stationary coordinate system; and the three-phase voltage command value is the SPWM vector modulation signal of each phase.

The motor control method according to claim 1 or 2, wherein the calculating the sinusoidal pulse width modulation SPWM vector modulation signal of each phase of the current carrier cycle is:

The motor running state detection data and the command data are obtained by triggering an interrupt update sampling value at a midpoint of a carrier cycle.

4. The motor control method according to claim 3, wherein

The motor running state detection data includes: a motor rotor position angle, a motor rotor angular velocity, and a motor current; the motor rotor position angle is detected by a rotor position detector to obtain a rotor rotor position angle, and the motor rotor angular velocity is based on an adjacent rotor position angle The difference is divided by the sampling time.

5. The method according to claim 1, wherein the zero sequence component is obtained by: z _z = - ^max (^, W:, (丄 - ) * ^min (w:, W:, 3⁄4 :) ⁺ ( ² — ; where ^ is a zero-order component; _{3⁄4 £} is the SPWM vector modulation signal of each phase in the carrier period; K is a constant greater than or equal to zero and less than or equal to 1.

The method according to claim 1, wherein the determining the duty ratio of each phase in the carrier period according to the SVPWM vector modulation signal is specifically calculated by using the following formula:

T ^{+ 0} - ⁵ U _d

Among them, T. ^ is the conduction time of each phase, M _d is the DC bus voltage, which is the sampling period; the conduction time T of each phase in one sampling period. The dividing of _c by the sampling period T _s is the duty ratio corresponding to each phase.

The method according to claim 1, wherein the generating, according to the determined duty ratio of each phase in the current carrier period, generating a PWM control signal corresponding to each phase, specifically:

The carrier period is counted by a count register, and the count register value is a value held by the time base period register TBPRD;

One carrier cycle includes the count register incrementing from 0 to the value saved by TBPRD, and then counting down from the value saved by TBPRD to 0 symmetrical process;

The value of the count comparison register A is calculated according to the duty ratio of each phase by the formula, and the formula is: CMPA = TBPRD / 2; when the count value CTR = 0, the current carrier cycle starts, and CTR <CMPA, The phase PWM control signal remains low; when CTR = CMPA, and the count register is in the up count phase, the phase PWM control signal transitions to a high level output; when CTR = CMPA, and the count register During the down counting phase, the phase PWM control signal transitions to a low level output.

8. The method according to claim 1, wherein the controlling the turning on and off of each arm of the inverter in the current carrier cycle is specifically:

When the PWM control signal corresponding to a certain phase is at a high level, a high level signal is output to the upper arm of the phase, so that the bridge arm is turned on, and a low level signal is output to the lower arm of the phase, so that the bridge arm is turned off. ;

When the phase PWM control signal is low, a low level signal is output to the upper arm of the phase, so that the upper arm is turned off, and a high level signal is output to the lower arm of the phase, so that the lower arm of the phase is turned on.

9. A motor control apparatus using space vector pulse width modulation, comprising: an SPWM vector modulation signal calculation unit, configured to receive motor running state detection data and instruction data obtained in a current carrier cycle, and calculate according to the same SPWM vector modulated signal corresponding to the carrier period;

a driving signal generating unit, configured to receive a PWM control signal output by the PWM control signal generating unit, and generate two driving signals according to the PWM control signals of the respective phases, and the driving signal is output to the upper and lower bridges of each phase of the inverter The arm controls the turning on and off of each arm of the inverter.

10. The motor control apparatus according to claim 9, wherein the SPWM vector modulation signal calculation unit comprises:

a fixed or synchronous coordinate converter for receiving motor current detection values for the current carrier cycle, and a rotor position detection value, and calculating an actual electric power of the d-axis and the q-axis of the synchronous rotating coordinate system according to the detected value: ^ value i _d , iq;

a current controller for receiving the current command value /, iq ^x of the d-axis and the q-axis, and an actual current value ^ i _{q of the} d-axis and the q-axis, and calculating a synchronous rotating coordinate system in combination with the obtained motor rotor angular velocity D-axis voltage command value and q-axis voltage command value V _d *; V *;

a synchronous or fixed coordinate converter for receiving the synchronous rotating coordinate system d-axis voltage command value and the q-axis voltage command value, and converting the d-axis voltage command value and the q-axis voltage command value into a stationary coordinate system The three-phase voltage command value output; the three-phase voltage command value in the static coordinate system is a desired SPWM vector modulation signal.

11. A motor control apparatus according to claim 9 or 10, wherein the motor rotor angular velocity, the motor current, and the current motor torque command value are all sampled at a midpoint of the carrier cycle.

12. The motor control apparatus according to claim 10, wherein the zero sequence component of the current carrier cycle used by the SVPWM vector modulation signal calculation unit is specifically obtained by:

( ^2k ~ ¹ ); wherein, the zero sequence component; is the SPWM vector modulation signal of each phase in the carrier cycle; Κ is a constant greater than or equal to zero and less than or equal to 1.

The motor control device according to claim 9, wherein the duty ratio calculation unit specifically calculates the duty ratio of each phase in the current carrier cycle by using the following formula; 7 _be : **

T = ^{+ 0} - ⁵ U _d ;

Among them, T. , for each phase conduction time, M _d is the DC bus voltage, and is the sampling period;

丄 s is the duty cycle corresponding to each phase;

The motor control device according to claim 9, wherein the PWM control signal generating unit comprises: The counting register is configured to perform frequency counting on each carrier cycle to implement time measurement of the carrier cycle; the register value of the counting register is a value saved by TBPRD, and the timing of one carrier cycle includes counting from 0 to TBPRD. Value, then two symmetrical processes that count down from the value held by TBPRD to 0;

CMPA = TBPRD *†1 1 2 -

I a, b , c ' comparison result output unit, configured to receive the comparison value of each phase output by the comparison value calculation unit, and compare the comparison value with the current count value CTR of the count register, according to the comparison result Generate different output signals as PWM control signals; when CTR = 0, 3⁄4 new carrier cycle, at this time CTR<CMPA, the PWM control signal of the unit output is low; when CTR = CMPA, and the counting register is In the up counting phase, the unit jumps to the high level of the PWM control signal corresponding to the phase output; when CTR = CMPA, and the counting register is in the down counting phase, the unit jumps to the PWM control signal corresponding to the phase output. Level.

The motor control device according to claim 9, wherein the motor running state detection data comprises a motor rotor position angle, a motor rotor angular velocity, and a motor current; and the motor rotor position angle is detected by a rotor position detector. The rotor position angle of the rotor, which is calculated according to the detected value of the adjacent rotor position angle by the following formula: , ø _(κ)

Indicates the rotor position angle obtained by the 采样 detection at the current sampling time; Θ -7 indicates the rotor position angle obtained at the previous sampling time (K-1); it is the sampling interval time.

The motor control device according to claim 15, wherein the rotor position detector is a resolver or a Hall position sensor.