WO2007119855A1 - Dispositif de conversion d'energie et procede de regulation correspondant - Google Patents

Dispositif de conversion d'energie et procede de regulation correspondant Download PDF

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Publication number
WO2007119855A1
WO2007119855A1 PCT/JP2007/058377 JP2007058377W WO2007119855A1 WO 2007119855 A1 WO2007119855 A1 WO 2007119855A1 JP 2007058377 W JP2007058377 W JP 2007058377W WO 2007119855 A1 WO2007119855 A1 WO 2007119855A1
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WIPO (PCT)
Prior art keywords
current
output
value
phase
command value
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PCT/JP2007/058377
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English (en)
Japanese (ja)
Inventor
Tomomichi Ito
Shoichiro Koseki
Yasuhiro Kiyofuji
Takashi Aihara
Shuji Kato
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Hitachi, Ltd.
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Priority to CN2007800133675A priority Critical patent/CN101421684B/zh
Priority to JP2008511024A priority patent/JP5051127B2/ja
Publication of WO2007119855A1 publication Critical patent/WO2007119855A1/fr

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/18—Arrangements for adjusting, eliminating or compensating reactive power in networks
    • H02J3/1821—Arrangements for adjusting, eliminating or compensating reactive power in networks using shunt compensators
    • H02J3/1835—Arrangements for adjusting, eliminating or compensating reactive power in networks using shunt compensators with stepless control
    • H02J3/1842—Arrangements for adjusting, eliminating or compensating reactive power in networks using shunt compensators with stepless control wherein at least one reactive element is actively controlled by a bridge converter, e.g. active filters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
    • Y02E40/20—Active power filtering [APF]

Definitions

  • the present invention relates to a power converter connected to an AC system, and more particularly, to a power converter that suppresses voltage fluctuation (fretting force) caused by load fluctuation.
  • Japanese Patent No. 2 6 7 5 2 0 6 proposes a flicker suppressing device for suppressing flicker.
  • the above device calculates active and reactive currents from the load current, calculates the normal phase by applying a single-pass filter, and calculates the reverse phase by applying a high-pass filter. By outputting a current in the opposite phase to the detected load current, the fluctuation of the current flowing in from the system is canceled, thereby suppressing the flick force.
  • the current command value contains harmonic components.
  • a power converter includes a current detector that detects an alternating current output current, and a current that controls the alternating current output current so that the output current command value matches the output value of the current detector.
  • a current command value calculator that includes a control means, expands the output value of a current detector that detects a current flowing into the load, and calculates the output current command value according to each output value of the Fourier series expansion. It is characterized by having a vessel.
  • a power converter includes a current detector that detects an AC output current, and controls an AC output current so that an output current command value matches the current detector output value. And a current control means for generating two sine wave generators that generate two sine waves having the same frequency and a phase difference of 90 degrees, and the output of the current detector that detects the output value of the sine wave generator and the current flowing into the load. It has a current command value calculator that multiplies the force value and calculates the output current command value from the result of moving average calculation of the multiplication result.
  • a power converter includes a current detector that detects an AC output current, and controls an AC output current so that an output current command value matches the current detector output value. And a current control means for generating two sine wave generators that generate two sine waves having the same frequency and a phase difference of 90 degrees, and the output of the current detector that detects the output value of the sine wave generator and the current flowing into the load.
  • a current command value calculator for calculating a cycle integral in which the multiplication result is obtained by multiplying the multiplication result by using a cycle of the sine wave output from the sine wave generator as an integration period, and calculating an output current command value from the cycle integral value It is characterized by having.
  • a power converter includes a current detector that detects an AC output current, and an AC output current so that an output current command value and the current detector output value match. Equipped with current control means to control The output value of the current detector for detecting the flowing-in current is expanded by a Fourier series, a phase compensation filter operation is performed to advance the phase to the Fourier series expansion output value, and an output current command value is calculated according to the result of the filter operation calculation. It has a current command value calculator.
  • the power conversion device further includes: a current command value calculator that performs a phase compensation filter calculation that advances a phase to a value obtained by moving average calculation of the multiplication result, and calculates an output current command value according to the filter calculation result It is characterized by having.
  • the power conversion device of the present invention is characterized in that the phase compensation filter is constituted by a first-order advance / delay filter.
  • the power conversion device of the present invention is characterized in that the phase compensation filter is configured by a first-order incomplete component.
  • the power conversion device of the present invention is characterized in that the frequency of the sine wave output from the sine wave generator is equal to the grid system frequency.
  • the power conversion device is characterized in that the frequency of the sine wave output from the sine wave generator is an integral multiple of the grid system frequency.
  • a power converter includes a current detector that detects an AC output current, and controls an AC output current so that an output current command value matches the current detector output value.
  • Current control means for detecting the current flowing into the load, and expanding the output value of the current detector to a Fourier series, and expanding the output value of the current series to a frequency twice that of the grid system frequency.
  • a current command value calculator for calculating an output current command value according to the filter calculation result.
  • a power converter according to the present invention includes a current detector that detects an AC output current, and controls an AC output current so that an output current command value matches the current detector output value.
  • a current control means for generating two sine wave generators that generate two sine waves having the same frequency and a phase difference of 90 degrees, and the output of the current detector that detects the output value of the sine wave generator and the current flowing into the load. Multiply the force value, calculate the period integral with the multiplication result as the integration period of the period of the sine wave output from the sine wave generator, and at a frequency twice that of the system frequency linked to the period integration result It has a current command value calculator for performing a filter operation with a gain smaller than the gain at the system frequency and calculating an output current command value according to the filter operation result.
  • a power converter includes a current detector that detects an AC output current, and controls an AC output current so that an output current command value matches the current detector output value.
  • Current control means for generating a sine wave generator for generating two sine waves having the same frequency and 90 degrees different phases, and an output of a current detector for detecting the sine wave generator output value and the current flowing into the load. Multiply the values, and perform a filter operation with a gain at a frequency twice that of the grid frequency connected to the grid frequency less than the gain at the system frequency to the value obtained by moving average calculation of the multiplication result, and output according to the filter operation result It has a current command value calculator for calculating a current command value.
  • the power conversion device is characterized in that a filter having a gain at a frequency twice as high as the system frequency is smaller than the gain at the system frequency is constituted by a notch filter.
  • the power converter according to the present invention is characterized in that a gain at a frequency twice the system frequency is smaller than a gain at the system frequency by subtracting the bandpass filter output value from the input signal. It is what.
  • the present invention provides an amplitude detection method for detecting an amplitude value from a signal including an alternating current component whose amplitude varies, from a value obtained by applying a phase compensation filter to an output value of Fourier series expansion. The amplitude value is calculated.
  • the present invention provides a method for controlling a power conversion device that changes operating conditions in accordance with the voltage of an AC system to be connected or the current flowing through the system.
  • the current is subjected to Fourier series expansion, and the AC output voltage of the power converter is changed using a value obtained by applying a phase compensation filter to the value obtained by the Fourier series expansion.
  • the present invention provides a phase compensation filter on the output value of the family series expansion in an amplitude detection method for detecting the amplitude value from a signal including an alternating current component whose amplitude varies.
  • the amplitude value is calculated from the value, and the current command value is changed using the calculated value.
  • the present invention provides a method for controlling a power conversion device that changes operating conditions in accordance with the voltage of an AC system to be connected or the current flowing through the system.
  • the present invention is characterized in that the AC output voltage of the power converter is changed using a value obtained by performing a Fourier series expansion of the current and applying a phase compensation filter to the value obtained by the Fourier series expansion.
  • a power converter includes a current detector that detects a load current flowing into a load, a current detector that detects an AC output current, and an AC output current according to a current command value.
  • Current control means to control And only the fundamental wave component included in the load current is compensated.
  • the means for extracting the fundamental wave component of the load current is a Fourier series expander.
  • a power converter includes a current detector that detects an AC output current, and controls an AC output current so that an output current command value matches the current detector output value.
  • a current control means for detecting the current flowing into the load as an input, and extracting a component whose amplitude fluctuates between 0.1 Hz and 30 Hz from the fundamental wave. It is characterized by having an output current command value calculator for adding a current in phase with the current component to the output current command value.
  • the power converter of the present invention is characterized in that a fundamental wave component whose amplitude fluctuates between 0.1 Hz and 30 Hz is calculated by a Fourier series expander.
  • the power conversion device of the present invention is characterized by including a phase correction calculator that performs a phase correction calculation on the extracted load current fundamental wave component.
  • the phase correction computing unit compensates for the phase delay caused by the computation of the Fourier series expander.
  • the power conversion device of the present invention is characterized in that the phase correction arithmetic unit performs phase compensation including the phase delay due to the operation of the Fourier series expander and the delay of the current controller.
  • a power converter includes a current detector that detects an AC output current, and controls an AC output current so that an output current command value matches the current detector output value.
  • a power converter having a current control means for detecting a current flowing into a load The ratio of the amplitude of the harmonic component in the output current of the power converter to the amplitude of the harmonic component in the load current is between the orders in the output current of the power converter with respect to the amplitude of the inter-order harmonic component in the load current. It is characterized by a smaller amplitude ratio of harmonic components.
  • the power conversion device of the present invention is characterized by detecting a system current instead of detecting a current flowing into a load.
  • the power converter of the present invention estimates a load current from the detected value of the grid current and the detected AC output current value output from the power converter, and uses the estimated load current as a detected value of the current flowing into the load.
  • the current command value is calculated.
  • a power converter includes a current detector that detects an AC output current, and controls an AC output current so that an output current command value matches the current detector output value.
  • the current command value is a composite value of the positive phase current command value and the negative phase current command value
  • the negative phase current command value is calculated from the rated current of the power conversion device. It is characterized in that it is limited to a small predetermined value or less, and the positive phase current command value is limited to fall within the range of the difference between the rated current and the negative phase current command value limited to the predetermined value or less. is there.
  • a power converter includes a current detector that detects an AC output current, and controls an AC output current so that an output current command value matches the current detector output value.
  • a power conversion device having a current control means configured to provide a current command value that is a composite value of a positive phase current command value and a negative phase current command value
  • the positive phase current command value is less than or equal to a rated current of the power converter.
  • the negative-phase current command value is limited to fall within the range of the difference between the rated current and the positive-phase current command value at the rated current.
  • the power conversion device of the present invention is characterized in that the negative-phase current command value falls within a range of a difference between the rated current and the positive-phase current command value and is limited to a predetermined value smaller than the rated current. Is.
  • the power converter of the present invention is characterized in that the negative phase current command value is limited by multiplying the negative phase current command value by a variable gain.
  • the power conversion device of the present invention is characterized in that the reverse-phase current is limited by a circular limiter.
  • FIG. 1 is an explanatory diagram of a first embodiment of the present invention.
  • FIG. 2 is an explanatory view of a second embodiment of the present invention.
  • FIG. 3 is an explanatory diagram of the gain characteristic improvement effect and the phase characteristic improvement effect by the phase correction filter of the second embodiment of the present invention.
  • FIG. 4 shows another example of realization of the phase correction filter according to the second embodiment of the present invention.
  • FIG. 5 is an explanatory diagram of the first embodiment of the present invention.
  • FIG. 6 is an explanatory view of a third embodiment of the present invention.
  • FIG. 7 is an explanatory diagram of a fundamental wave DFT calculator according to the third embodiment of the present invention.
  • FIG. 8 is an explanatory diagram of a secondary DFT calculator according to the third embodiment of the present invention.
  • FIG. 9 is a diagram for explaining the operation of the first embodiment of the present invention.
  • FIG. 10 is an explanatory view of a modification of the first embodiment of the present invention.
  • FIG. 11 is an explanatory diagram of a fourth embodiment of the present invention.
  • FIG. 12 shows another example of realizing the phase correction filter according to the fourth embodiment of the present invention.
  • FIG. 13 is an explanatory diagram of the fourth embodiment of the present invention. BEST MODE FOR CARRYING OUT THE INVENTION
  • the flicker suppressing device 1 of the present invention is connected in parallel with a load 3 and is connected to an AC power source 2 via a system impedance 7.
  • System impedance 7 means transmission line impedance and transformer impedance.
  • the flicker suppressing device of the present invention suppresses / reduces flicker due to load current fluctuation.
  • the flicker suppressing apparatus 1 includes a main circuit unit and a control calculation unit 100.
  • the main circuit is composed of a power converter 4 consisting of IGBT and diode, a filter reactor 5 and a DC capacitor 6.
  • the AC output terminal of the power converter 4 is connected to the AC system via the filter reactor 5.
  • a DC capacitor 6 is connected in parallel to the DC output terminal of the power converter 4.
  • the control calculation unit 1 0 0 receives the connection point voltage detection value, the AC current detection value, and the DC capacitor voltage detection value of the flicker suppression device 1 as inputs, and the current of the power converter 4 is maintained so as to maintain the DC capacitor voltage. Calculate the command value.
  • a current command value for suppressing flicker is calculated from the load current detection value and the connection point voltage detection value, and the sum of the current command value and the current command value for maintaining the DC capacitor voltage is newly added.
  • the AC output voltage command value of the power converter 4 is calculated so that the current command value of the power converter 4 matches the AC current detection value of the power converter 4.
  • the control calculation unit 1 0 0 compares the AC output voltage command value with the carrier wave. 0
  • the gate signal to I GB T of power converter 4 is calculated and output to power converter 4 to output an AC voltage following the command value.
  • the DC capacitor voltage is detected by the voltage detector 14, and the detection value VD C is input to the subtractor 1 0 4. .
  • the subtractor 1 0 4 calculates the difference between the DC capacitor voltage command value VD C REF and VD C and outputs it to the DC voltage controller 1 0 5.
  • the DC voltage controller 1 0 5 receives the DC capacitor voltage deviation as input, and calculates the effective current command value I AVR of the flicker suppression device 1 so that the DC capacitor voltage matches the command value VD C REF.
  • the effective current command value I AVR is input to the inverse d 1 Q converter 1 1 2.
  • the inverse d-Q converter 1 1 2 calculates the current command value of the flicker suppression device 1 from the flicker suppression current command value described below and the output of the DC voltage controller 1 0 5.
  • phase detector 1 0 1 calculates the fundamental phase 0.
  • phase 0 is Vcos 0 for the fundamental component of V u
  • Vcos ( ⁇ -4/3 ⁇ ) for the fundamental component of Vw Is the phase.
  • Phase 0 is input to sine wave tables 1 0 2 and 1 0 3, and sine wave tables 1 0 2 and 1 0 3 calculate cos ⁇ and sin 0, respectively.
  • cos0 and sin0 are output to the normal phase DFT calculator 1 07, the negative phase DFT calculator 1 0 8, the reverse dQ converter 1 1 2, and the dq converter 1 1 3.
  • the load current is detected by the current sensors 2 0, 2 1, 2 2, and the detected values ILU, ILV, I LW are output to the ⁇ — j6 converter 1 0 6.
  • the ⁇ - ⁇ converter 1 0 6 calculates I L ⁇ , I L / 3 by the calculation shown in the following equation.
  • the I L and I L ⁇ calculated by the j8 converter 10 6 are output to the positive phase D F arithmetic unit 10 07 and the negative phase D F T arithmetic unit 10 8.
  • the positive phase DFT calculator 1 07 calculates the amplitude value IL 1 RE of the positive phase real axis component and the amplitude value IL 1 IM of the positive phase imaginary axis component of the load current by the calculation shown in (Formula 2).
  • t is the current time
  • f s is the grid frequency of the interconnection
  • T l / f s .
  • I L Xsin0 and — I L Xsin ⁇ + I L j3 Xcos ⁇ is a component that oscillates at double the system frequency, but is integrated in the power cycle T, so the output value is zero.
  • the negative phase D F T calculator 1 0 9 calculates the amplitude value I L 2 R E of the negative phase real axis component of the load current and the amplitude value I L 2 I M of the negative phase imaginary axis component by the calculation shown in (Equation 3).
  • IL2RE 2f s ) t (IL Of xcos 0) dt- (iL ⁇ xsin 0) dt ⁇
  • ILQ; and IL ⁇ contain distortion components because they contain harmonics, but IL 1 RE, IL 1 IM, IL 2 RE and IL 2 IM do not generate distortion components, and harmonics can be eliminated. I understand.
  • IL 1 RE and IL 2 RE can be periodically integrated to eliminate anti-phase contamination in positive-phase computation and positive-phase contamination in anti-phase computation, and the values of IL 1 RE and IL 2 RE are respectively It turns out that the amplitude of the positive phase component and the amplitude of the negative phase component in the load current coincide with each other, and the amplitude calculation can be realized correctly.
  • IL 1 RE, IL 1 IM, IL 2 RE, and IL 2 IM are the positive-phase active component amplitude value, positive-phase invalid component amplitude value, negative-phase real-axis component amplitude value,
  • the load 3 changes the period integral value of the negative phase component in the positive phase DFT calculator 1 0 7 and the positive phase component in the negative phase DFT calculator 1 0 9 It is not completely zero.
  • the output IL 1 IM of the positive phase DFT calculator 1 0 7 is input to the notch filter 10 8 and the double frequency component of the system frequency is removed by the notch filter 1 0 8 IL 1 IM 2 To the inverse d — q converter 1 1 2 Four
  • the output IL 2 RE of the anti-phase DFT calculator 1 0 9 is input to the notch filter 1 1 0, IL 2 IM to the notch filter 1 1 1, and the value obtained by removing the double frequency component of the system frequency IL 2 RE 2 , Input IL 2 IM 2 to d-Q converter 1 1 3.
  • the inverse d-q converter 1 1 2 receives the output of the DC voltage controller 1 0 5 and the notch filter 1 0 8, the reference sine waves cos 0 and sinS, and performs the calculation shown in (Equation 5).
  • the phase current command values IL la and IL l jS are calculated.
  • the d-q converter 1 1 3 inputs the output values IL2RE2, IL 2 IM 2 of the notch filters 1 0 9 and 1 1 0 and the reference sine wave cos0 and sin0, and performs the calculation shown in (Equation 6).
  • Reverse phase current command value IL 2 a;, IL 2) 3 is calculated.
  • the output of the inverse d-q converter 1 1 2 and the d-q converter 1 1 3 is input to the adders 1 1 4 and 1 1 5. Since the flicker suppressor 1 suppresses the flicker force by outputting a current having a phase opposite to that of the load current, the outputs of the adders 1 1 4 and 1 1 5 are the sign inverters 1 1 6 and 1 1 1. The value whose sign is inverted by 7 is used as the current command value IC a REF, IC
  • the current command values IC and REF, IC; 3 REF are calculated by the DC voltage controller 1 0 5, the positive phase DFT calculator 1 0 7, and the negative phase DFT calculator 1 0 9.
  • the value does not include harmonic components.
  • the alternating current output from flicker suppression device 1 is current sensor 1 1, 1 2 and 1 3 are detected, and detected values I CU, ICV, and I CW are subjected to the same calculation as in (Equation 1) by ⁇ -converter 1 1 8, and their outputs IC a and IC / 3 are respectively Output to subtracters 1 1 9 and 1 2 0.
  • the subtractor 1 1 9 calculates the deviation between the current command value ICQ! REF and IC a, and the subtractor 1 2 0 calculates the deviation between the current command value IC / 3 REF and IC j3.
  • the output is the current controller 1 2 Entered in 1.
  • the current controller 1 2 1 calculates the AC output voltage correction amount to reduce the current deviation calculated by the subtractors 1 1 9 and 1 20.
  • the output value of the current controller 1 2 1 is the AC voltage detection value V u, V V, Vw) 3)
  • the output value V a REF of the adder 1 2 3 and the output value V / 3REF of the adder 1 2 4 are input to the two-phase / three-phase converter 1 2 5 and the three-phase is calculated by the calculation shown in (Equation 7). Voltage command values V u REF, V v REF, and VwR EF are calculated.
  • the output value of the 2-phase to 3-phase converter 1 2 5 is input to the P WM calculator 1 2 6 and compared with the triangular wave output by the carrier wave calculator 1 2 7.
  • the PWM calculator 1 2 6 calculates the I GBT gate signal of the power converter 4 from the magnitude comparison result and outputs it to the power converter 4.
  • the power converter 4 follows the gate signal output from the PWM calculator 1 2 6. 6
  • the flicker suppression device outputs a current in the opposite phase to the positive phase reactive current and the negative phase component of the load current. Fluctuations can be reduced, and flicker force can be suppressed.
  • the load current is detected by the current sensors 20, 2 1, 2 2, but as shown in FIG. 5, the grid currents I SU, ISV, I SW and the flicker suppression device output current ICU The same effect can be obtained by estimating the load currents ILU, ILV, I LW from ICV, ICW.
  • the same effect can be obtained by detecting the system current and outputting the current in the opposite phase to the normal phase reactive current and the opposite phase component.
  • the flicker suppressing device of the present invention outputs the positive phase reactive current, the negative phase component, and the negative phase current of the load current, the fluctuation of the system current can be reduced and the flicker force can be suppressed.
  • the current command value does not include harmonics
  • harmonic components flowing out from the flicker suppressing device 1 can be reduced, and increase in harmonics due to current control delay can be avoided. it can.
  • phase correction filter is provided at the output of the DFT calculator.
  • the response delay to fluctuations in the amplitude of the fundamental current / reverse phase load current can be improved.
  • the positive-phase DFT calculator 1 0 7 and the negative-phase DFT calculator 1 0 9 have the advantage of extracting the positive-phase fundamental component and the negative-phase fundamental component and removing the harmonic component, respectively (Equation 2 ), (3) Since the calculation includes integration as shown in (Equation 3), if the amplitude of the fundamental wave fluctuates quickly, a decrease in gain and a phase delay occur in the amplitude calculation result.
  • the flicker suppressing apparatus of the present embodiment reduces the gain reduction and phase delay of the amplitude calculation result when the fundamental wave amplitude varies in the load current.
  • the flicker suppressing apparatus 1 calculates the positive-phase reactive current amplitude value I L 1 I M by the reverse-phase DFT calculator 1 07.
  • the output is input to the phase correction filter 1 3 0 and the output is input to the notch filter 1 0 8.
  • the transfer function G (s) of the phase correction filter 1 30 is an advance / delay filter represented by (Equation 8).
  • T and T 2 are time constants and s is a Laplace operator. By using this filter, gain characteristics and phase characteristics can be improved.
  • phase correction filter 1 3 1, 1 3 2 are phase correction filters. The same advance and delay Phil evening as Le Yu 1 3 0.
  • the load current can be amplified as shown in (Equation 9).
  • Fig. 4 shows the transfer characteristics of the current amplitude I (t) to the output IL 1 I MF IL of the phase correction filter.
  • the horizontal axis is the amplitude fluctuation frequency f
  • the vertical axis is the gain in Fig. 3 (a) and the phase in Fig. 3 (b).
  • the frequency characteristics when the phase correction filter is not used are indicated by broken lines, and the frequency characteristics when the phase correction filter is used are indicated by solid lines.
  • the fundamental wave amplitude increases as the gain becomes 0 d ⁇ and the phase approaches O deg
  • the phase correction filter is a primary advance / delay filter, but the same applies to a filter combining a plurality of primary phase advance / delay filters and a phase correction filter combining incomplete differentiation as shown in Fig. 4. The effect of can be obtained.
  • the flicker suppressing device of the present invention outputs the positive phase reactive current, the negative phase component, and the negative phase current of the load current, the fluctuation amount of the system current can be reduced and the flicker force can be suppressed.
  • the current command value does not include harmonics
  • harmonic components flowing out from the flicker suppressing device 1 can be reduced, and increase in harmonics due to current control delay can be avoided. it can.
  • the transfer characteristic of the DFT calculator can be improved, it is possible to improve the flicker suppression function when the fundamental wave amplitude vibrates quickly.
  • Example 1 The difference between this example and Example 1 is that a secondary DFT is provided in addition to the fundamental wave,
  • It is configured to calculate the amplitude and phase of the second harmonic component and add it to the current command value.
  • the current control delay is small. Therefore, it is possible to output compensation currents for specific harmonics as well as fundamental waves. For example, if the current control response is 1 0 0 0 radZ s, the cut-off frequency is 1600 Hz, so if the system frequency is 60 Hz, the current control characteristics will reach about the second harmonic. Can be secured.
  • the specific harmonic is a second harmonic.
  • phase ⁇ calculated by the phase detector 1 0 1 is doubled by the multiplier 1 5 4, and the output is input to the sine wave tables 1 5 5 and 1 5 6.
  • the sine wave table 1 5 5 calculates cos 2 ⁇ from the phase 2 0 output from the multiplier 1 5 4, and the sine wave table 1 5 6 calculates sin 2 0 and outputs it to the second-order D F T calculator 1 5 1.
  • the second-order DFT calculator 1 5 1 uses the output value of the 1/3 converter 1 0 6 as input, and calculates the real-axis component amplitude and imaginary-axis component amplitude of the second-order harmonic contained in the load current. From these values, calculate the current command values IC a REF 2 and IC j3 REF 2 in the opposite phase to the second harmonic component contained in the load current.
  • the output of the fundamental wave D F T calculator 1 5 0 and the output of the second-order D F T calculator 1 5 1 are input to the adders 1 5 2 and 1 5 3, respectively.
  • the fritz force suppressing device of the present embodiment extracts the fundamental wave component and the secondary component of the load current, and calculates the current command value of the reverse phase. Since the delay in current control is small with respect to lower-order harmonic current commands such as the second order, the harmonic increase phenomenon can be avoided.
  • Figure 7 shows the calculation block of the secondary D F T calculator 1 51.
  • Load currents IL o ;, IL ⁇ and sine waves cos 2 0, sin 2 0 are input to the positive phase DFT calculator 1 5 1 1.
  • the positive phase DFT calculator 1 5 1 1 performs the calculation shown in (Formula 10) and calculates the outputs IL 1 RE 2 and IL 1 IM 2.
  • IL1 RE2 4f s
  • I L 1 R E 2 and I L 1 I M 2 are second-order positive-phase real axis components and positive-phase imaginary axis components based on the sine waves cos 2 ⁇ and sin 2 0, as in (Equation 2).
  • the negative-phase DFT calculator 1 5 1 2 performs the calculation shown in (Formula 1 1) and calculates the outputs I L 2 R E 2 and I L 2 I M 2.
  • I L 2 RE 2 and I L 2 I M 2 are the fifth-order anti-phase real axis component and anti-phase imaginary axis component when the sine waves cos 2 0 and sin 2 0 are used as a reference, as in (Equation 3).
  • the output of the positive phase DFT calculator 1 5 1 1 is input to the notch filters 1 5 1 3 and 1514, the frequency component of 4 times the system frequency is removed, and input to the inverse d-Q converter 1 5 1 7 .
  • the inverse d-q converter 1 5 1 7 performs the calculation shown in (Equation 1 2) from the output of the notch filters 1 5 1 3 and 1 5 1 4 and the sine waves cos 2 0 and sin 2 0 and outputs the output IL lo ; 2, IL l iS 2 is output.
  • "IL1 a 2" cos2 ⁇ -sin2 ⁇ '
  • the output of the inverse d-Q converter 1 5 1 7 is output to the adders 1 5 1 9 and 1 5 2 0.
  • negative-phase D F T calculator 1 5 1 2 is input to notch filters 1 5 1 5 and 1516, and the frequency component four times the system frequency is removed and input to dq converter 1 5 1 8.
  • the d-q converter 1 5 1 8 performs the calculation shown in (Equation 1 3) from the output of the notch filters 1 5 1 5 and 1 5 1 6 and the sine waves cos 5 0 and sin 5 0, and the output IL 2 a; 2, IL 2 i3 2 is output.
  • d 1 q Output of converter 1 5 1 8 IL 2 Q; 2, output of IL 2) 3 2 is output to adders 1 5 1 9 and 1 5 2 0.
  • Adder 1 5 1 9, 1 5 2 0 is inverse d-Q converter 1 5 1 7 and d-Q converter
  • the value whose sign is inverted by the sign inverters 1 5 2 1 and 1 5 2 2 has the opposite phase to the second harmonic component of the load, and is the output of the second order DFT calculator 1 5 1.
  • a current command value having a phase opposite to that of the second harmonic component included in the load current can be calculated.
  • the second order is selected as the specific harmonic, but the harmonic order may be selected as a lower order harmonic such as the third order.
  • a DFT calculator that compensates for multiple low-order harmonics such as the second and third orders may also be provided.
  • the flicker suppressing device of the present invention outputs the positive phase reactive current, the negative phase component and the negative phase current of the load current, the fluctuation of the system current can be reduced and the flicker can be suppressed.
  • the current command value does not include harmonics
  • harmonic components flowing out from the flicker suppressing device 1 can be reduced, and increase in harmonics due to current control delay can be avoided. it can.
  • the compensation current can be calculated for the specific harmonic contained in the load current, flicker caused by low-order harmonics can be reduced.
  • Embodiment 1 The difference between this embodiment and Embodiment 1 is that a limiter 160 is provided for the current command value.
  • the DC capacitor When the reverse-phase current is output, the DC capacitor generates a pulsation that is twice the system frequency. If the fluctuation of the DC capacitor voltage is large, it will cause heat generation of the capacitor and outflow of harmonics from the power converter. Therefore, the DC capacitor voltage should be kept within the specified range (for example, 10% of the rated voltage). Is desirable.
  • the present invention limits the reverse-phase current with a circular limiter, maintains the phase relationship, and limits only the compensation amount.
  • the limiter 160 limits the current command value so that the grid currents I CU, I C V, and I CW output from the power converter 1 are controlled within the rated output current I MAX.
  • the priority is given to the output I AVR of the DC voltage controller 1 0 5, the output value IL 1 IM 2 of the notch filter 1 0 8, and the current command value IL 2 RE 2, which becomes the negative phase current command value, Perform the limit calculation in the order of IL 2 IM 2.
  • the structure of the limiter 160 is shown in FIG.
  • Output value of DC voltage controller 1 0 5 I AVR is limited to 1 I MAX or more and I MAX or less by limiter 1 6 0 1 and reverse as new positive phase active current command value I AVR 2 d-q converter 1 1 Output to 2.
  • the square value of I MAX is calculated by the multiplier 1 6 0 2
  • the square value of IA VR 2 is calculated by the multiplier 1 6 0 3
  • the difference between them is calculated by the subtractor 1 6 0 4
  • the square root calculator 1 6 0 Outputs to 5.
  • the output value of the square root calculator 1 6 0 5 is the upper limit value of the positive phase reactive current limiter 1 6 0 6, and the value whose sign is inverted by the multiplier 1 6 0 7 is the lower limit value of the limit 1 6 0 6 .
  • the output value of limiter 1 6 0 6 becomes the new normal phase reactive current command value I L 1 I M 3 and is output to the reverse d-ci converter 1 1 2.
  • the amplitude of the positive-phase current is less than I MAX.
  • Positive phase active current command value I AVR 2 and positive phase reactive current command value IL 1 IM 3 are input to amplitude calculator 1 6 0 8 and positive phase current command value amplitude I 1 ABS is output.
  • the power converter I MAX and the amplitude I 1 ABS are input to the subtractor 1 6 0 9, and the difference is output to the minimum value calculator 1 6 1 1. Since the output value of the subtractor 1 6 0 9 is the value obtained by subtracting the positive phase current command value from the rated current of the power converter 1, if the amplitude value of the negative phase current command value is smaller than the difference, the power converter 1 The output current command value does not exceed the rated I MAX.
  • the negative phase current real part component I L 2 R E 2 and the imaginary part component I L 2 I M 2 are input to the amplitude calculator 1 6 1 0 to calculate the negative phase current command value amplitude I 2 ABS.
  • the amplitude value is input to the minimum value calculator 1 6 1 1.
  • the maximum negative phase current value I 2 MAX determined from the allowable vibration width of the DC capacitor voltage is also input to the minimum value calculator 1 6 1 1.
  • Minimum value calculator 1 6 1 1 takes the value obtained by subtracting the positive-phase current command value amplitude from the rated current and the negative-phase current allowable value I 2 MAX and negative-phase current command value amplitude I 2 ABS as the inputs. The smaller one is output to the divider 1 6 1 2.
  • Divider 1 6 1 2 divides minimum value calculator 1 6 1 1 by the negative-phase current command value amplitude and outputs the quotient to multipliers 1 6 1 3 and 1 6 1 4.
  • Multipliers 1 6 1 3 and 1 6 1 4 multiply the negative phase current command value I L 2 R E, I L 2 I M 2 by the output value of divider 1 6 1 3 to obtain a new negative phase current I L 2 R E 3,
  • the negative-phase current command value amplitude matches the smallest of the remaining current output, the maximum negative-phase current value, and the negative-phase current at the rated current, so that the negative-phase current can be limited. Since the negative phase current command value can be limited, the fluctuation range of the DC capacitor voltage can be limited.
  • the calculation when the positive phase current is prioritized is shown, but the current command value may be limited with the negative phase current prioritized.
  • Fig. 13 shows the limiter 160 when priority is given to reverse-phase current.
  • the negative phase current command values I L 2 R E 2 and I L 2 I M 2 are input to the amplitude calculator 1 6 1 0, and the amplitude I 2 ABS is output to the minimum value calculator 1 6 1 1 and the divider 1612.
  • the minimum value calculator 1 6 1 1 is also supplied with the allowable negative phase current value I 2 MAX, and the smaller value is output to the divider 1 6 1 2 and the subtractor 1 6 0 9.
  • Divider 1 6 1 2 divides the output value of the minimum value calculator by I 2 ABS and outputs the result to multipliers 1 6 1 3 and 1 6 1 4.
  • Multipliers 1 6 1 3 and 1 6 1 4 are input with anti-phase real part component IL 2 RE 2 and anti-phase imaginary part component IL 2 IM 2 respectively, and new anti-phase current command values IL2RE3 and IL 2 IM 3 And is output to the d-q converter 1 1 3.
  • the amplitude of the negative-phase current command value becomes I 2 MAX or less, so that the oscillation width of the DC capacitor voltage can be suppressed.
  • the difference between the rated current I MAX of the power converter 1 and the amplitude of the negative-phase current command value is calculated by the subtractor 1 6 0 9 and the limiter that limits the positive-phase current command value IA VR 1 6 0 1
  • the lower limit value is the upper limit value and the half-inverted sign of the subtractor 1 6 0 9.
  • the output value of the subtractor 1 6 0 9 is input to the multiplier 1 6 0 7 and the square value is calculated.
  • the positive-phase active component IAVR2 limited by the limiter 1 6 0 1 is also input to the multiplier 1 6 0 3, the difference is calculated by the subtractor 1 6 0 4, and the square root calculation is the limiter 1 6 0 6
  • the upper limit value and the sign reversed This is the lower limit of 1 6 0 6.
  • the flicker suppressing device of the present invention outputs the positive phase reactive current, the negative phase component and the negative phase current of the load current, the fluctuation of the system current can be reduced and the flicker can be suppressed.
  • the current command value does not include harmonics
  • harmonic components flowing out from the power converter device 1 can be reduced, and increase in harmonics due to current control delay can be avoided.
  • the reverse phase current can be suppressed to a predetermined value or less, the fluctuation range of the DC capacitor voltage can be suppressed.
  • the frequency component that is an integral multiple of the system frequency can be removed from the current command value.
  • the harmonics flowing out from the flicker suppressing device can be reduced while suppressing the above.
  • the present invention can be applied to a power converter connected to an AC system, and more particularly to a power converter that suppresses voltage fluctuations (fretting force) caused by load fluctuations.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Inverter Devices (AREA)
  • Rectifiers (AREA)

Abstract

Selon la présente invention, dans un dispositif de suppression de scintillement destiné à supprimer des scintillements grâce à la détection d'un courant de charge et la sortie du courant en opposition de phase, le courant de charge peut contenir des harmoniques élevés. Dans ce cas, la phase des composants harmoniques élevés devant être sortis par le dispositif de suppression de scintillement peut être retardée par la temporisation de la régulation du courant, et le courant harmonique élevé de la charge peut ne pas s'annuler alors que les harmoniques élevés peuvent au contraire être augmentés. L'expansion de la série de Fourier permet alors de calculer l'amplitude d'une composante ondulatoire fondamentale du courant de charge en vue de calculer la valeur de commande du courant du dispositif de suppression de scintillement sur la base de l'amplitude calculée, de telle sorte que le courant sortant du dispositif de suppression de scintillement peut se réguler selon la valeur de commande du courant. Le calcul de la valeur de commande du courant à partir du coefficient de la série de Fourier permet d'éliminer de la valeur de commande du courant la composante de fréquence plusieurs fois supérieure à la fréquence du système, de manière à ce que le dispositif de suppression de scintillement puisse réduire le flux d'harmoniques élevés qui en sort tout en supprimant les scintillements.
PCT/JP2007/058377 2006-04-13 2007-04-11 Dispositif de conversion d'energie et procede de regulation correspondant WO2007119855A1 (fr)

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CN2007800133675A CN101421684B (zh) 2006-04-13 2007-04-11 功率变换装置及其控制方法
JP2008511024A JP5051127B2 (ja) 2006-04-13 2007-04-11 電力変換装置およびその制御方法

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JP2009176236A (ja) * 2008-01-28 2009-08-06 Hitachi Ltd 電力変換装置
JP2011516024A (ja) * 2008-03-24 2011-05-19 アメリカン パワー コンバージョン コーポレイション Ｕｐｓ、周波数変換器およびラインコンディショナ
JP2012185720A (ja) * 2011-03-07 2012-09-27 Toshiba Mitsubishi-Electric Industrial System Corp 自励式無効電力補償装置の制御装置
US8467210B2 (en) 2007-11-07 2013-06-18 Siemens Aktiengesellschaft Method for controlling a VAr compensator
JP2016082662A (ja) * 2014-10-15 2016-05-16 ダイキン工業株式会社 電力変換器制御装置

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CN102437576B (zh) * 2011-12-20 2014-04-16 安徽佑赛科技有限公司 一种有源电力滤波器的控制器及其控制方法
JP6555389B2 (ja) * 2017-05-30 2019-08-07 ダイキン工業株式会社 電源品質管理システムならびに空気調和装置
WO2019017373A1 (fr) * 2017-07-18 2019-01-24 ダイキン工業株式会社 Système de filtre actif et dispositif de climatisation
CN109617423B (zh) * 2018-10-25 2019-12-31 武汉船舶通信研究所（中国船舶重工集团公司第七二二研究所） 大功率极低频电源及其次谐波抑制装置

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JPH10336897A (ja) * 1997-06-03 1998-12-18 Mitsubishi Heavy Ind Ltd パワーアクティブフィルタ制御装置
JPH11122821A (ja) * 1997-10-09 1999-04-30 Shinko Electric Co Ltd 同期発電機の単独運転検出装置及び同期発電機の単独運転検出方法
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US8467210B2 (en) 2007-11-07 2013-06-18 Siemens Aktiengesellschaft Method for controlling a VAr compensator
JP2009176236A (ja) * 2008-01-28 2009-08-06 Hitachi Ltd 電力変換装置
JP2011516024A (ja) * 2008-03-24 2011-05-19 アメリカン パワー コンバージョン コーポレイション Ｕｐｓ、周波数変換器およびラインコンディショナ
JP2012185720A (ja) * 2011-03-07 2012-09-27 Toshiba Mitsubishi-Electric Industrial System Corp 自励式無効電力補償装置の制御装置
JP2016082662A (ja) * 2014-10-15 2016-05-16 ダイキン工業株式会社 電力変換器制御装置

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