USRE38547E1 - Frequency-modulated converter with a series-parallel resonance - Google Patents

Frequency-modulated converter with a series-parallel resonance Download PDF

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USRE38547E1
USRE38547E1 US09/084,625 US8462598A USRE38547E US RE38547 E1 USRE38547 E1 US RE38547E1 US 8462598 A US8462598 A US 8462598A US RE38547 E USRE38547 E US RE38547E
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capacitor
transistor
inductor
series
transformer
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US09/084,625
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English (en)
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Julius Hartai
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Lumicae Patent AS
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Lumicae Patent AS
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    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/42—Conversion of DC power input into AC power output without possibility of reversal
    • H02M7/44—Conversion of DC power input into AC power output without possibility of reversal by static converters
    • H02M7/48—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5383—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a self-oscillating arrangement
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters
    • H05B41/282—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters with semiconductor devices
    • H05B41/2821—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters with semiconductor devices by means of a single-switch converter or a parallel push-pull converter in the final stage
    • H05B41/2824—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters with semiconductor devices by means of a single-switch converter or a parallel push-pull converter in the final stage using control circuits for the switching element
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention concerns a frequency modulated converter with a series-parallel resonance, particularly for driving any ohmic or inductive load, including gas discharge tubes, wherein a commutating voltage switch in the form of a transistor is provided in is connected in series between the negative electrode of a direct current voltage source and a first terminal of an inductor, wherein a pulse generator circuit is provided between the voltage source and a control electrode of the transistor and wherein a second terminal of the inductor is connected to the primary winding of a transformer.
  • the invention further comprises a first capacitor and a rectifier diode provided in a first and second parallel branch respectively between the charge emitting and the charge receiving electrode of the transistor, and a second capacitor is provided across the electrodes of the voltage source and additionally provides a smoothing capacitance for the voltage source, the second capacitor being connected in series with the inductor via the diode.
  • a more effective way of converting power at increasingly higher frequencies is based on the so-called “zero current switching” wherein a sinusoidal voltage which may be generated by a LC-resonant tank connected either in parallel or series, is used. Such converters are called “resonant converters”.
  • the advantage of using a sine voltage is that losses in the power semiconductors are dramatically reduced, as the switching generally takes place at the zero crossing.
  • the disadvantage of resonant converters is that at a given power level the peak current is many times greater than that of a pulse width modulated converter.
  • a frequency modulated control in the form of an integrated circuit which may be used in the range beyond 1 MHz, with the designation LD 405, obtainable from Gennum Corporation, Burlington, Ontario, Canada.
  • the use of this control circuit in a frequency modulated converter is described in a LD 405 application note from Gennum Corporation with the title “Using LD 405 in a 125 W resonant mode power supply”.
  • LD 405 in a 125 W resonant mode power supply.
  • the circuit comprises an inductance L, a capacitance C, a resistance R and a load R L .
  • a commutating switch S for instance in the form of a transistor, its purpose is to supply direct current from a source V to a series resonant tank LC.
  • the resistance of the load R L drains current from the tank.
  • the switch S opens and the power conversion from the source S to the load R L is interrupted. After a given period of Lime time the switch S again closes and the process is repeated.
  • the commutation frequency may be changed such that the average power dissipated in the load R L is changed.
  • a resonant converter of this type works with two commutating switches each of which handles a respective half-cycle of the resonant period.
  • the switches are based on MOS field effect transistors which themselves are driven by a respective MOSFET stage.
  • the output stage of the shown embodiment is based on Schottky rectifier diodes.
  • U.S. Pat. No. 4,613,769 discloses a transistor oscillator circuit wherein a capacitor is connected in parallel over the terminals of a secondary winding in a transformer, the latter operating in parallel sine wave resonance with the capacitor as a first wave-shaping means at a given frequency.
  • a second wave-shaping means consists of another capacitor connected in parallel over the collector and emitter electrodes of the transistor oscillator and operates in series resonance with an inductor, at twice the given frequency.
  • the object of the present invention is to provide a resonant circuit without the above-mentioned and other disadvantages.
  • This is achieved according to the invention in that the first capacitor and the inductor together form a series resonance circuit, the relationship between the inductor voltage U L and the capacitance of the first capacitor determining the series resonance frequency of a first half-cycle, that the second capacitor and the diode together form a parallel resonance circuit, the inductor voltage U L and the capacitance of a second capacitor determining the parallel resonance frequency of a second half-cycle, that the transistor is in a high ohmic state during both the series and parallel resonance mode, that the diode acting as an impedance selector between the capacitors to maintain the correct current flow in the transformer and the load is conducting in the parallel resonance mode, charging the capacitor above the voltage source level, before the transistor is switched into a low ohmic state and completes that parallel resonance mode and then initiates another series-parallel resonance when switched into the high ohmic state, each half-cycle of the
  • the present invention also consists of a frequency-modulated converter with series-parallel resonance, particularly for driving any ohmic or inductive load (R G ), including gas discharge tubes, wherein a commutating voltage switch (Q) in the form of a transistor is connected in series between the negative electrode of a direct voltage source and a first terminal of an inductor (L,P), wherein a pulse generator circuit is provided between the voltage source and a control electrode of the transistor (Q) with a transformer (T), and further comprises: a first capacitor (C 1 ) and a rectifier diode (D 2 ) provided in a first and second parallel branch respectively between the charge emitting and the charge receiving electrode of the transistor (Q); a second capacitor (C 3 ) provided across the electrodes of the voltage source, the second capacitor (C 3 ) being connected in series with the inductor (L) via the diode (D 2 ); the first capacitor (C 1 ) and the inductor (L) together forming a series resonance circuit, the
  • FIG. 1 shows as already mentioned, the basic circuit of a parallel resonator according to prior art.
  • FIG. 2 shows the basic circuit of a modulated converter with series-parallel resonance according to the present invention and used with a cold cathode gas discharge tube.
  • FIG. 3 shows a section of a variant of the converter of FIG. 2 used with a hot cathode gas discharge tube.
  • FIGS. 4 a-c show respectively the voltage curve measured across the inductor of the converter at normal load on the output, at short-circuited output as well as a cycle of the resonant voltages under different conduction states and load conditions.
  • FIG. 5 shows a practical embodiment of the frequency modulated converter according to the invention and applied to driving a hoe hot cathode gas discharge tube.
  • FIG. 6 shows a practical embodiment of the transformer in as provided in the circuit of FIG. 2 and FIG. 6a shows the transformer T in greater detail.
  • a first resonant capacitor C 1 is provided in parallel across the charge emitting and the charge receiving electrode of a transistor Q which operates as a commutating switch.
  • the load R G is provided in series with an inductor L which is connected to the transistor Q and the capacitor C 1 respectively.
  • a second resonant capacitor C 3 is provided across the electrode of the voltage source and connected to the inductor L over a diode D 2 , the diode D 2 being connected in a further parallel branch between the charge emitting and the charge receiving electrode of the transistor Q.
  • the primary winding P of a transformer T is connected to the inductor L, such that the transformer T, the inductor L and the capacitors C 1 , C 3 provide a RCL resonator operating in series parallel to the transistor Q voltage source and with a quality factor which as well-known is determined by the relationship between the inductor voltage U L or the capacitor voltages U C1 and U C3 and the supply voltage U.
  • the load R G is connected between the terminals of a first secondary winding S 1 of the transformer T and hence as mentioned is connected in series with the inductor L.
  • the dimensioning of the resonator may advantageously be done on basis of the apparent power requirement, such that the resonator or the resonant tank is dimensioned for an apparent power which is 30% larger than that which is required at the chosen operating frequency of the resonator.
  • the transistor Q is controlled at a determined frequency which need not vary in dependence of the load.
  • the transistor Q is controlled by an approximate square pulse.
  • the inductor L is made with a coil and, e.g. of ferrite with an air gap.
  • the counterinduction of the inductor L causes the capacitors C 1 or C 3 to be charged.
  • the capacitor C 3 has however a capacitance which is far greater than the capacitance of the capacitor C 1 and will also be charged with the opposite polarity.
  • the transformer T is now fed with current of the same polarity as the current received over the transistor Q.
  • phase 1 the transistor Q is conducting and the current flows in the direction I A through the transformer T.
  • phase 2 the transistor has ceased conducting but due to the fact that the inductor L works as “tank”, current still is flowing in the direction I A (FIG. 2) through the transformer T, while the capacitor C 1 at the same time is charged due to the counterinduction of the inductor L.
  • phase 3 the counterinduction from the inductor L has ceased and the capacitor C 1 is discharged such that the current flows to the capacitor C 3 and through the transformer T in the direction I B (FIG. 2 ), while the inductor L is “filled”.
  • phase 4 the inductor L then is “emptied” via the diode D 2 and the capacitor C 3 as well as the transformer T until the transistor Q again becomes conducting.
  • the transistor Q may be switched each time the diode D 2 conducts and hence also at “zero” current and voltage.
  • the negative counterinduction voltage U L from the inductor L adds to the supply voltage U and is applied over the primary winding P of the transformer T, while the capacitor C 3 is discharged by both U and U L .
  • the converter according to the invention attains a very high efficiency.
  • the switching losses are completely eliminated, as the transistor switches on in the negative phase of the resonance when the diode D 2 conducts, and when the transistor Q is disconnected, the voltage supply is taken over by the capacitor C 1 .
  • the transistor Q hence works only with the voltage which is necessary to maintain the behavior of the induction curve of the inductor L.
  • the impedance of the transformer T decreases to zero and the phase shift between the inductor L and the transformer T ceases. All energy is then used for maintaining the resonance and the energy consumption of the converter is reduced to “zero”. That is to say that the converter is safe against short-circuiting in any respect.
  • a second secondary winding S 2 is used in the transformer and connected with a rectifier bridge in order to return a part of the energy to respectively the positive and negative electrode of the voltage source. In this way there is always a certain minimum impedance in the transformer T. The resonator will then operate within the given frequency range and the energy will circulate between the source of the supply voltage and the secondary winding S 2 via the rectifier bridge as shown in FIG. 2 .
  • the free-running losses may be minimized and it is possible to provide a detector (not shown) which warns about possible faults of the load R G , for instance a faulty gas discharge tube, in order to disconnect the pulse generator circuit which is connected to the control electrode of the transistor Q. Hence the transistor Q ceases refilling the resonator.
  • a hot cathode gas discharge tube is used as a load on the secondary side of the transformer T, this can simply be done as shown in FIG. 3 by connecting the terminals of the secondary winding S 1 with at least one capacitor C 6 over the electrodes K 1 , K 2 in the gas discharge tube.
  • gas discharge tubes with hot cathodes must be started by means of a preheating of the electrodes in order to achieve sufficient ionization of the gas in the tube and that a discharge may take place.
  • This is achieved by the secondary winding S 1 and the capacitor C 6 being adapted to the resonance frequency of the transformer T with the cathodes K 1 , K 2 in heated condition.
  • Such an adaption may be determined empirically or by the heat resistance of the cathode being measured and added to the impedance. As long as the cathodes K 1 , K 2 are not sufficiently heated, the impedance is too low and the greater part of the current from the secondary winding S 1 is used for heating the cathodes.
  • the capacitor C 6 no longer operates as a resonant capacitor, but nevertheless contributes with a certain glow voltage which keeps the electrodes heated due to the impedance of the former being low compared with the frequency. This is moreover an advantage if dimming is used by reducing the supply voltage.
  • the converter according to the invention may also be used with a pulsating direct current without smoothing for direct driving of gas discharge tubes with a power factor cos ⁇ up to 0.95 and without use of phase compensation, as the new European norms require.
  • the capacitor C 1 is for instance dimensioned to 0.005 ⁇ F and the capacitor C 3 to 0.22 ⁇ F, but at 100 kHz the capacitor C 1 is selected with 0.003 ⁇ F and the capacitor C 3 with 0.15 ⁇ F.
  • the wavelength consideration of the transport of cathode material between the electrodes moreover indicates that an operating frequency of 30-35 kHz is an optimum with the present invention of gas discharge tubes.
  • FIGS. 4a and 4b show the behavior of the inductor voltage U L measured across the terminals of-the inductor L.
  • the voltage in the FIGS. 4a and 4b is referred to an average peak-to-peak value U L .
  • the total period of the voltage is pulse t 1 , while the transistor conducts in t 2 .
  • the load on the secondary winding S 1 is normal, the voltage is low (here 0.6 U L ) and t 1 short (here 0.15 t 1 ) due to the energy drain.
  • the secondary winding is short-circuited.
  • the voltage increases (here to 1.3 U L ) and the same does t 2 because the energy drain has ceased.
  • FIG. 4c shows a period of the resonant voltage under different load conditions.
  • the normal sinusoidal behaviour is denoted by F 1 .
  • the curve F 2 is present when the transformer T “steals” energy and the curve F 3 when the secondary winding S 1 is short-circuited.
  • the free-running diode conducts all the time in the negative half-cycle, i.e. in the period T D .
  • the transistor Q conducts in the period T Q1 . If the energy is drained over the transformer T, the transistor Q operates in the period T Q2 , as the system is self-controlling.
  • the transistor Q conducts in the period T Q3 , energy is no longer drained to the transformer T and in the negative half-cycle the inductor L delivers the greater part of the energy back to the capacitor C 3 .
  • FIGS. 2 and 3 show somewhat more fundamental embodiments of the converter according to the invention.
  • a rectifier bridge B 2 delivers DC voltage from an AC source, and this voltage is smoothed in the capacitors C 2 and C 3 .
  • the diode D 12 supplies a pulse generator circuit which in the embodiment shown comprises a bistable multivibrator in the form of a Schmitt trigger circuit with inverted outputs made up by six gates A 1 -A 6 .
  • the voltage of the pulse generator circuit is regulated by the zener diode Z 1 and smoothed by the capacitor C 4 .
  • the pulse generator circuit provides an astable multivibrator circuit over the resistor R 6 and the diode D 1 as well the variable resistance R V (R 5 ) such that there is provided a tuning of the basic frequency and pulse width to the desired value via the resistors R V and R 6 and the capacitor C 5 .
  • the output of the gate A 1 delivers an approximate square pulse and controls the inputs of the four gates A 3 -A 6 in approximate parallel.
  • the respective outputs of the same gates are also connected in parallel and to the control electrode of the transistor Q which is used as a switch. If a common bipolar transistor is used, the control input is of course the basis of the transistor, but if a MOS field effect transistor is used instead, the control electrode is of course identical to the gate electrode.
  • the converter according to the invention may be realized with an integral free-running diode, such that the rectifier diode D 2 and the second parallel branch in FIG. 2 are dropped.
  • the resonance frequency may be finely tuned over the variable resistance R V .
  • the secondary winding S 1 of the transformer T delivers voltage and current to the provided load as discussed in more detail in connection with FIG. 2 .
  • a third secondary winding S 3 in the transformer T is used for increasing the ionization voltage of the load if the latter is a gas discharge tube in order to provide safer ignition at extremely low temperatures, as its first terminal is connected with the electrodes of the gas discharge tube and its second terminal is led to the ground as shown.
  • the transformer T is in a practical embodiment realized as E-core transformers, as shown in detail in FIG. 6 .
  • the cores and windings may be made for instance in the form of ferrite strips with a dielectric film and the windings deposited thereon.
  • the E-core transformers used in non-conventional applications e.g. for a frequency of 30-100 kHz, still allow a very compact construction.
  • the inductor L is integrated with the primary winding p of the transformer T.
  • the second winding S 2 which is connected to the rectifier bridge B 1 , is dimensioned such that a direct voltage is obtained over the rectifier diodes D 7 -D 10 in the bridge B 1 , the voltage being lower than the voltage across C 2 and C 3 in normal operation.
  • the resistors R 10 and R 11 then constitute a voltage divider against the capacitor C 20 , which is given a value which determines the desired period of time before the pulse generator circuit and via the diode D 5 disconnects the astable multivibrator. If the signal on A 1 is low, the outputs of the gates A 3 -A 6 also go to low. The duration of the disconnection is determined by the capacitor C 20 via the resistors R 7 and R 8 .
  • the resonant capacitor C 1 which in the embodiment in FIG. 5 is realized as a parallel circuit of nine capacitors C 1a -C 1i (not shown), only operates as a resonant capacitor during a half-period of the frequency.
  • the second half-period of the resonant frequency used for refilling the resonant tank, i.e. the inductor L 1 is provided via the capacitor C 3 by the discharge of C 1 .
  • fly-back oscillators tend to give asymmetrical frequency behaviour by energy being drawn from the resonant tank already in the first flyback such that the next half-cycle receives a low energy content.
  • the resonant capacitor C 1 in FIG. 2 hence must receive a greater charge during the second half-period. This is achieved by the already existing charge in the capacitor C 3 being connected in series with the inductor L via diode D 2 and at a voltage level identical to that of the first half-cycle.
  • the transformer will be fed the same amount of energy in both half-cycles.
  • a symmetrization of the energy in each half-cycle of the resonance Together with the use of an air gap in the transformer T, this makes the resonator provide an approximately perfect sinusoidal voltage without the transformer's primary winding P being biased by a DC component.
  • the transistor Q By correctly chosen values of the inductance of the inductor L and the impedance of the transformer T as well as correct capacitance values for the capacitors C 1 and C 3 and suitable supply voltage U, it is possible to achieve a very high efficiency as the switching losses are completely eliminated and the transistor Q only works with a fraction of the current of the circuit due to the phase shift between current and voltage in the inductive components.
  • the transistor Q can in reality be regarded as a voltage switch which sets the resonance circuit to zero in relation to the positive and negative cycles of the resonance. Hence the transistor eliminates the resonator's tendency to relaxation and maintains the given frequency, while current mainly is taken up by the inductor L when the transistor Q is not conducting.
  • the air gap used in the transformer T may also be achieved by individual adaption of the air gap used in the transformer T to the characteristic of the provided load R G .
  • the air gap as best seen in FIG. 6a may hence be used actively for controlling the energy drain of the inductor L and the capacitor C 1 .
  • a correct dimensioning of the transformer T and the air gap used may by full short-circuiting of S 1 bring the resonator to full resonance within the frequency range determined by the transistor Q.
  • the pulse generator appropriately also may be realized in another manner than as an astable multivibrator, as the latter for instance may be replaced by a digital frequency synthesizer.
  • the frequency will only be controllable within 10-15%.
  • a digital frequency synthesizer may drive the converter according to the present invention over a frequency range which stretches from the AF domain and to 100 MHz and beyond, while the generated frequency easily may be controlled over an octave band or more. Then the converter may also be used in HF and VHF applications where high, stable and symmetrisized resonance voltages are required.
  • the loss is limited to losses in the transformer, the pulse generator circuit, the energy dissipation in the resonant inductor and in the rectifier bridge at the input.
  • the total losses may thus be kept to 5% or less, such that in the practical embodiment of the converter according to the invention achieves an efficiency in the order of 97%.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Inverter Devices (AREA)
  • Dc-Dc Converters (AREA)
  • Amplifiers (AREA)
  • Circuit Arrangements For Discharge Lamps (AREA)
  • Radar Systems Or Details Thereof (AREA)
  • Stabilization Of Oscillater, Synchronisation, Frequency Synthesizers (AREA)
  • General Induction Heating (AREA)
  • Discharge-Lamp Control Circuits And Pulse- Feed Circuits (AREA)
  • Magnetic Resonance Imaging Apparatus (AREA)
  • Electrophonic Musical Instruments (AREA)
  • Measurement Of Velocity Or Position Using Acoustic Or Ultrasonic Waves (AREA)
  • Control Of Stepping Motors (AREA)
  • Semiconductor Lasers (AREA)
  • Burglar Alarm Systems (AREA)
  • Electronic Switches (AREA)
  • Ac-Ac Conversion (AREA)
US09/084,625 1991-08-27 1992-08-25 Frequency-modulated converter with a series-parallel resonance Expired - Fee Related USRE38547E1 (en)

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NO913368A NO913368D0 (no) 1991-08-27 1991-08-27 Frekvensmodulert driver med parallell-resonans.
NO913368 1991-08-27
PCT/NO1992/000133 WO1993004570A1 (en) 1991-08-27 1992-08-25 Frequency-modulated converter with a series-parallel resonance
US09/084,625 USRE38547E1 (en) 1991-08-27 1992-08-25 Frequency-modulated converter with a series-parallel resonance
US08/199,212 US5561349A (en) 1991-08-27 1992-08-25 Frequency-modulated converter with a series-parallel resonance

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US20040053584A1 (en) * 2002-09-18 2004-03-18 Mickle Marlin H. Recharging method and apparatus
US20040259604A1 (en) * 2003-05-20 2004-12-23 Mickle Marlin H. Recharging method and associated apparatus
US20050192062A1 (en) * 2002-09-18 2005-09-01 Mickle Marlin H. Recharging method and apparatus

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CA2267366C (en) * 1997-07-22 2006-01-10 Eugen Statnic Process for generating voltage pulse sequences and circuit assembly therefor
US7592753B2 (en) * 1999-06-21 2009-09-22 Access Business Group International Llc Inductively-powered gas discharge lamp circuit
US6570370B2 (en) * 2001-08-21 2003-05-27 Raven Technology, Llc Apparatus for automatic tuning and control of series resonant circuits
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US7440780B2 (en) * 2002-09-18 2008-10-21 University Of Pittsburgh - Of The Commonwealth System Of Higher Education Recharging method and apparatus
US20090098915A1 (en) * 2002-09-18 2009-04-16 University Of Pittsburgh-Of The Commonwealth System Of Higher Education Recharging method and apparatus
US7567824B2 (en) 2002-09-18 2009-07-28 University Of Pittsburgh-Of The Commonwealth System Of Higher Education Recharging method and apparatus
US8090414B2 (en) 2002-09-18 2012-01-03 University of Pittsburgh—of the Commonwealth System of Higher Education Recharging method and apparatus
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KR100297201B1 (ko) 2001-10-24
NO941551L (no) 1994-04-27
SG45432A1 (en) 1998-01-16
MY108100A (en) 1996-08-15
ES2110520T3 (es) 1998-02-16
DK0601091T3 (da) 1998-05-25
CA2116347A1 (en) 1993-03-04
NO941551D0 (enExample) 1994-04-27
JP3339636B2 (ja) 2002-10-28
CN1041787C (zh) 1999-01-20
NO307440B1 (no) 2000-04-03
US5561349A (en) 1996-10-01
JPH06510393A (ja) 1994-11-17
HU9400572D0 (en) 1994-05-30
AU668103B2 (en) 1996-04-26
EP0601091A1 (en) 1994-06-15
FI940877A7 (fi) 1994-04-08
MX9204943A (es) 1993-04-01
HU218120B (hu) 2000-06-28
AU2546192A (en) 1993-03-16
HUT67419A (en) 1995-04-28
ATE161382T1 (de) 1998-01-15
EP0601091B1 (en) 1997-12-17
DE69223633T2 (de) 1998-05-20
CN1073556A (zh) 1993-06-23
CA2116347C (en) 2003-02-18
NO913368D0 (no) 1991-08-27
RU2154886C2 (ru) 2000-08-20
FI940877A0 (fi) 1994-02-25
WO1993004570A1 (en) 1993-03-04
DE69223633D1 (de) 1998-01-29

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