EP0700599B1 - Entree de circuit cmos a seuil dynamique compense en v cc? - Google Patents

Entree de circuit cmos a seuil dynamique compense en v cc? Download PDF

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Publication number
EP0700599B1
EP0700599B1 EP95910702A EP95910702A EP0700599B1 EP 0700599 B1 EP0700599 B1 EP 0700599B1 EP 95910702 A EP95910702 A EP 95910702A EP 95910702 A EP95910702 A EP 95910702A EP 0700599 B1 EP0700599 B1 EP 0700599B1
Authority
EP
European Patent Office
Prior art keywords
current path
inverter
input
output terminal
switching threshold
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Expired - Lifetime
Application number
EP95910702A
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German (de)
English (en)
Other versions
EP0700599A1 (fr
Inventor
Brian Martin
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Koninklijke Philips NV
Original Assignee
Koninklijke Philips Electronics NV
Philips Electronics NV
Philips Norden AB
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Application filed by Koninklijke Philips Electronics NV, Philips Electronics NV, Philips Norden AB filed Critical Koninklijke Philips Electronics NV
Publication of EP0700599A1 publication Critical patent/EP0700599A1/fr
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Publication of EP0700599B1 publication Critical patent/EP0700599B1/fr
Anticipated expiration legal-status Critical
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K19/00—Logic circuits, i.e. having at least two inputs acting on one output; Inverting circuits
    • H03K19/003—Modifications for increasing the reliability for protection
    • H03K19/00369—Modifications for compensating variations of temperature, supply voltage or other physical parameters
    • H03K19/00384—Modifications for compensating variations of temperature, supply voltage or other physical parameters in field effect transistor circuits
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K19/00—Logic circuits, i.e. having at least two inputs acting on one output; Inverting circuits
    • H03K19/02—Logic circuits, i.e. having at least two inputs acting on one output; Inverting circuits using specified components
    • H03K19/08—Logic circuits, i.e. having at least two inputs acting on one output; Inverting circuits using specified components using semiconductor devices
    • H03K19/094—Logic circuits, i.e. having at least two inputs acting on one output; Inverting circuits using specified components using semiconductor devices using field-effect transistors

Definitions

  • the invention relates to an electronic circuit comprising a logic gate with a first switching threshold and having at least one input terminal for receiving at least one an input signal; an output terminal for providing an output signal; first and second supply voltage nodes for connection to respective voltage supplies; first current path means between the first supply voltage node and the output terminal for charging the output terminal in response to the input signal; second current path means between the output terminal and the second supply voltage node for discharging the output terminal in response to the input signal; additional current path means in parallel to at least a specific one of the first or second current path means; first control means responsive to the input signal for turning on the additional current path means when the respective current path is turned on; and second control means responsive to the input signal for turning off the additional current path means and comprising a buffer means with a second switching threshold different from the first switching threshold.
  • Such a circuit is known from IBM Technical Disclosure bulletin, Vol 31, No. 5, pp331-2. This known circuit reduces the parasitic ringing of the output during an output transition.
  • Another such a circuit of similar structure is known from PATENTS ABSTRACTS OF JAPAN, Vol 14, No. 248 (E-933) [4191] for JP-A-2 070 120. This another known circuit controls the rise and full waveforms of the output signal.
  • the invention relates in particular, but not exclusively, to an electronic circuit with field-effect transistors (FETs) of complementary conductivity type, i.e., with an N-channel FET (NFET) and a P-channel FET (PFET), also commonly referred to as a CMOS circuit, and that typically performs an inverting operation such as NOT, NOR or NAND.
  • FETs field-effect transistors
  • NFET N-channel FET
  • PFET P-channel FET
  • FIG. 1 A conventional CMOS inverter 10 is shown in Fig. 1.
  • Inverter 10 includes a PFET 12 and an NFET 14.
  • a gate 12g of PFET 12 is coupled to a gate 14g of NFET 14.
  • a source electrode 12s of PFET 12 is coupled to Vcc and a source electrode 14s of NFET 14 is coupled to ground.
  • a drain electrode 12d of PFET 12 is coupled to a drain electrode 14d of NFET 14.
  • the switching threshold of inverter 10 depends on the relative size of FETs 12 and 14 and the supply voltage Vcc. The higher the ratio of PFET to NFET, the higher the switching threshold.
  • the switching threshold is Vcc/2 and that Vcc is allowed to vary from 2.7 to 3.6 volts, the switching threshold could then vary from 1.35 to 1.8 volts. This is undesirable, for example, if the input of inverter 10 is a clock pulse specified to ramp from 2.7 to 0 volts with a 2.5 ns slew rate and the design requires the propagation time of the clock signal to be independent of Vcc.
  • the circuit is characterized in that the additional current path means comprises a first field effect transistor with a main path, and an input electrode coupled with an output of the buffer means; and a second field effect transistor with a main path, and an input electrode responsive to the input terminal; and in that the main paths of the first and the second field effect transistors are coupled in series between the first supply terminal and the output terminal.
  • the invention is based on the insight that the dependence of the first switching threshold of the logic gate on the supply voltage level can be compensated for through the dependence of operation of the current boosting means on the supply voltage level.
  • Fig. 2 depicts an inverter in accordance with a first embodiment of the present invention.
  • Inverter 20 is operable in the range from a voltage supply Vcc to ground and has an input terminal 22 and an output terminal 24.
  • Inverter 20 includes a first PFET 26 and an NFET 28.
  • PFET 26 has a gate 26g coupled to input terminal 22, a source 26s coupled to Vcc, and a drain 26d.
  • NFET 28 has a gate 28g coupled to gate 26g and to input terminal 22, a drain 28d coupled to drain 26d and a source 28s coupled to ground.
  • Inverter 20 also includes a second PFET 30 and a third PFET 32.
  • Second PFET 30 includes a source 30s coupled to Vcc, a drain 30d and a gate 30g.
  • Third PFET 32 includes a source 32s coupled to drain 30d, a drain 32d coupled to output terminal 24 and a gate 32g coupled to input terminal 22.
  • Inverter 20 also includes a further inverter 36 having its input coupled to input terminal 22 and its output coupled to gate 30g.
  • FETs 26 and 28 should preferably be sized so that when forming a standard inverter (consisting of FETs 26 and 28) the high-to-low switching threshold (the voltage at which standard inverter 26/28 switches) is approximately forty (40) percent of Vcc.
  • the switching threshold of further inverter 36 should preferably be set to approximately sixty (60) percent of Vcc. This relationship allows PFETs 30 and 32 and further inverter 36 to raise the effective switching threshold as explained with the following example.
  • Vcc is in the range from 2.7 to 3.6 volts. If Vcc equals 2.7 volts, the switching thresholds of standard inverter 26/28, and further inverter 36 are approximately 1.1 and 1.6 volts, respectively. Prior to a high-to-low transition, further inverter 36 holds gate 30g low so that PFET 30 is turned on. PFETs 26 and 32 are off and NFET 28 is on, holding output terminal 24 to ground. At this point no current is flowing through the circuit. As the input voltage ramps down, PFETs 26 and 32 begin to turn on and NFET 28 begins to turn off.
  • the effective switching threshold of inverter 20 is thus controlled by both by PFETs 26 and 32 and is equal to approximately 1.4 volts.
  • PFET 30 When the input voltage reaches 1.6 volts, further inverter 36 begins to switch, turning PFET 30 off.
  • PFET 30 Because of a propagation delay through further inverter 36, PFET 30 does not turn off instantly. Therefore, if the input edge rate is fast enough so that the input voltage passes through 1.4 volts before PFET 30 has started to turn off, the voltage at output terminal 24 will begin to switch at 1.4 volts.
  • the effective switching threshold of inverter 20 would be approximately 1.4 volts.
  • Vcc 3.6 volts
  • the high-to-low switching thresholds of standard inverter 26/28, and further inverter 36 would be 1.4 and 2.2 volts, respectively. If the same input signal is applied to inverter 20, further inverter 36 would begin to switch sooner relative to the voltage at output terminal 24 since the difference between the switching thresholds increases as Vcc increases. Therefore, the contribution from PFET 32 to the effective switching threshold of inverter 20 would be significantly less because PFET 30 would turn off sooner. As a result, the effective switching threshold of inverter 20 would be approximately 1.5 volts.
  • this embodiment of the invention creates a dynamic switching threshold of inverter 20 which varies by only 0.1 volts over a Vcc range of 0.9 volts.
  • Fig. 2 does not affect the switching threshold during a low-to-high transition. As the input voltage rises, the voltage passes through the switching threshold of standard inverter 26/28, before further inverter 36 turns PFET 30 on. Therefore, PFETs 30 and 32 have no function in determining the dynamic switching threshold of inverter 20.
  • Fig. 3 illustrates an inverter 120 constructed in accordance with a second embodiment of the invention wherein the effective switching threshold is compensated for during a fast low-to-high input signal transition affected.
  • Inverter 120 is operable in the range from a voltage supply Vcc to ground and has an input terminal 122 and an output terminal 124.
  • Inverter 120 includes a PFET 126 and a first NFET 128.
  • PFET 126 has a gate 126g coupled to input terminal 122, a source 126s coupled to Vcc and a drain 126d.
  • NFET 128 has a gate 128g coupled to gate 126g, a source 126s coupled to ground and a drain 128d coupled to drain 126d.
  • Inverter 120 also includes a second NFET 130 and a third NFET 132.
  • Second NFET 132 has a drain 132d coupled to output terminal 124, a source 132s and a gate 132g.
  • Third NFET 130 has a drain 130d coupled to source 132s, a source 130s coupled to ground and a gate 130g.
  • a further inverter 136 having its input coupled to input terminal 122 and its output coupled to gate 132g is also included in inverter 120.
  • FETs 126 and 128 should preferably be sized so that when forming a standard inverter (consisting of FETs 126 and 128) the low-to-high switching threshold is approximately forty (40) percent of Vcc.
  • the low-to-high threshold of further inverter 136 should preferably be set to approximately sixty (60) percent of Vcc.
  • NFETs 128 and 132 begin to turn on and PFET 126 begins to turn off.
  • the effective switching threshold of inverter 120 is now a combination of the effects of NFETs 128 and 132 and is approximately 1.4 volts.
  • further inverter 136 begins to switch, turning NFET 130 off.
  • NFET 130 is off, NFET 132 no longer contributes to the effective switching threshold so that the threshold returns to 1.1 volts.
  • NFET 130 does not turn off instantly.
  • the effective switching threshold of inverter 120 would be approximately 1.4 volts.
  • the switching thresholds of standard inverter 126/128 and further inverter 136 would be 2.2 and 1.4 volts, respectively. If the same input signal is applied to inverter 120, further inverter 136 would begin to switch sooner relative to the voltage at output terminal 124 since the difference between the switching thresholds increases as Vcc increases. Therefore, the contribution from NFET 132 to the effective low-to-high switching threshold of inverter 120 would be significantly less because NFET 130 would turn off sooner. As a result, the effective low-to-high switching threshold of inverter 120 would be approximately 1.5 volts. Thus, this embodiment of the invention creates a dynamic switching threshold for inverter 120 which varies by only 0.1 volts over a Vcc range of 0.9 volts.
  • Fig. 3 does not affect a high-to-low input signal transition. As the input voltage decreases, the voltage passes through the switching threshold of standard inverter 126/128 before further inverter 136 turns on NFET 130. Therefore, NFETs 130 and 132 do not affect the switching threshold of inverter 20.

Claims (2)

  1. Circuit électronique comprenant :
    une porte logique avec un premier seuil de commutation et comportant :
    au moins une borne d'entrée (22; 122) destinée à recevoir au moins un signal d'entrée ;
    une borne de sortie (24; 124) destinée à fournir un signal de sortie ;
    un premier (VCC) et un deuxième points logiques de tension d'alimentation en vue de la connexion aux alimentations de tension respectives;
    un premier moyen à trajet de courant (Q1) entre le premier point logique de tension d'alimentation (VCC) et la borne de sortie, ayant pour fonction de charger la borne de sortie en réaction au signal d'entrée;
    un deuxième moyen à trajet de courant (Q2) entre la borne de sortie et le deuxième point logique de tension d'alimentation, ayant pour fonction de décharger la borne de sortie en réaction au signal d'entrée;
    un moyen à trajet de courant supplémentaire (Q3, Q4 ; Q5, Q6) en parallèle à au moins un moyen à trajet de courant spécifique des premier (Q1) et deuxième (Q2) moyens à trajet de courant;
    un moyen de commande réagissant au signal d'entrée afin de bloquer le moyen à trajet de courant supplémentaire et comprenant un moyen tampon (36; 136) avec un deuxième seuil de commutation différent du premier seuil de commutation;
    caractérisé en ce que le moyen à trajet de courant supplémentaire comprend un premier transistor à effet de champ (Q4; Q6) avec un trajet principal et une électrode d'entrée couplée à une sortie du moyen tampon (36); ainsi qu'un deuxième transistor à effet de champ (Q3; Q5) avec un trajet principal et une électrode d'entrée connectée à la borne d'entrée; et en ce que les trajets principaux du premier et du deuxième transistors à effet de champ sont couplés en série entre l'une desdites bornes d'alimentation et la borne de sortie.
  2. Circuit suivant la revendication 1, dans lequel le premier moyen à trajet de courant (Q1) comprend des transistors à effet de champ d'un premier type de conductivité, dans lequel le deuxième moyen à trajet de courant (Q2) comprend des transistors à effet de champ d'un deuxième type de conductivité opposé au premier, et dans lequel le moyen à trajet de courant supplémentaire comprend des transistors à effet de champ d'un type de conductivité identique au moyen à trajet de courant spécifique des premier (Q1) et deuxième (Q2) moyens à trajet de courant.
EP95910702A 1994-03-25 1995-03-17 Entree de circuit cmos a seuil dynamique compense en v cc? Expired - Lifetime EP0700599B1 (fr)

Applications Claiming Priority (3)

Application Number Priority Date Filing Date Title
US21848194A 1994-03-25 1994-03-25
US218481 1994-03-25
PCT/IB1995/000183 WO1995026590A1 (fr) 1994-03-25 1995-03-17 Entree de circuit cmos a seuil dynamique compense en v¿cc?

Publications (2)

Publication Number Publication Date
EP0700599A1 EP0700599A1 (fr) 1996-03-13
EP0700599B1 true EP0700599B1 (fr) 1998-07-01

Family

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Family Applications (1)

Application Number Title Priority Date Filing Date
EP95910702A Expired - Lifetime EP0700599B1 (fr) 1994-03-25 1995-03-17 Entree de circuit cmos a seuil dynamique compense en v cc?

Country Status (6)

Country Link
US (1) US5495187A (fr)
EP (1) EP0700599B1 (fr)
JP (1) JP3586467B2 (fr)
KR (1) KR100365605B1 (fr)
DE (1) DE69503205T2 (fr)
WO (1) WO1995026590A1 (fr)

Families Citing this family (9)

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Publication number Priority date Publication date Assignee Title
JP2896305B2 (ja) * 1993-05-15 1999-05-31 株式会社東芝 半導体集積回路装置
US5532617A (en) * 1994-03-25 1996-07-02 Philips Electronics North America Corporation CMOS input with temperature and VCC compensated threshold
US5668488A (en) * 1994-11-17 1997-09-16 Advanced Micro Devices, Inc. Input buffer for a high density programmable logic device
US5739713A (en) * 1995-12-19 1998-04-14 Advanced Micro Devices, Inc. Deconvolution input buffer compensating for capacitance of a switch matrix of a high density programmable logic device
US5744982A (en) * 1996-04-22 1998-04-28 Taiwan Semiconductor Manufacturing Company, Ltd. Input buffer circuit
US6147513A (en) * 1998-05-01 2000-11-14 Winbond Electronics Corporation Method and circuit for logic input buffer
AU2003207915A1 (en) * 2002-03-06 2003-09-16 Igor Anatolievich Abrosimov Line driver with reduced power consumption
US7203243B2 (en) * 2003-03-10 2007-04-10 Acuid Corporation (Guernsey) Limited Line driver with reduced power consumption
WO2015045207A1 (fr) 2013-09-27 2015-04-02 パナソニック株式会社 Circuit intégré à semi-conducteurs et dispositif à circuit intégré à semi-conducteurs

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JPH0270120A (ja) * 1988-09-05 1990-03-09 Nec Corp 出力回路

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Also Published As

Publication number Publication date
JP3586467B2 (ja) 2004-11-10
WO1995026590A1 (fr) 1995-10-05
KR100365605B1 (ko) 2003-03-15
JPH08511399A (ja) 1996-11-26
EP0700599A1 (fr) 1996-03-13
KR960702698A (ko) 1996-04-27
DE69503205T2 (de) 1999-02-11
DE69503205D1 (de) 1998-08-06
US5495187A (en) 1996-02-27

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