EP0388492B1 - Onduleur capable de contrôler la fréquence de service - Google Patents

Onduleur capable de contrôler la fréquence de service Download PDF

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Publication number
EP0388492B1
EP0388492B1 EP19890105131 EP89105131A EP0388492B1 EP 0388492 B1 EP0388492 B1 EP 0388492B1 EP 19890105131 EP19890105131 EP 19890105131 EP 89105131 A EP89105131 A EP 89105131A EP 0388492 B1 EP0388492 B1 EP 0388492B1
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EP
European Patent Office
Prior art keywords
transistor
voltage
coupled
comparing
capacitor
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EP19890105131
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German (de)
English (en)
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EP0388492A1 (fr
Inventor
Keiichi Shimizu
Kenichi Inui
Nanjou Aoike
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Toshiba Electric Equipment Corp
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Toshiba Electric Equipment Corp
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Priority to EP19890105131 priority Critical patent/EP0388492B1/fr
Priority to DE1989612764 priority patent/DE68912764T2/de
Publication of EP0388492A1 publication Critical patent/EP0388492A1/fr
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Publication of EP0388492B1 publication Critical patent/EP0388492B1/fr
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    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters

Definitions

  • This invention relates to an inverter, and more particularly, to an inverter for operating a gas discharge lamp, such as a fluorescent lamp, in a high frequency manner.
  • the most common example for an inverter for converting a DC power to an AC power is a voltage resonance type inverter provided with a parallel voltage resonance circuit and a switching element.
  • the switching element interrupts an input DC voltage at a high frequency, which is higher than an acoustic frequency, e.g., 20 to 100 kHz, and applies the thus formed AC voltage to the voltage resonance circuit.
  • the AC voltage induced into the voltage resonance circuit is supplied to a load.
  • any particular measure is not used for controlling an operating frequency, i.e., a frequency of the on/off operation of the switching element. Therefore, the operating frequency is normally fixed in the self-excited inverter, while it depends on a load to which an AC voltage is to be supplied.
  • These inverters are disclosed in Japanese Patent Publication No. 57-45040, Japanese Patent Disclosure (Kokai) No. 61-2299, and Japanese Utility Model Disclosure (Kokai) No. 62-69396.
  • the voltage applied to the switching element depends on an operating state of the load receiving the AC voltage.
  • the voltage applied to the switching element is apt to be an over-voltage.
  • the used switching element must have a high breakdown voltage. This implies that the switching element for the inverter is expensive and hence the cost to manufacture the resultant inverter is increased.
  • an object of the present invention is to provide an inverter which surely prevents a switching element from being applied with an overvoltage at low cost and lessens variation in output voltage with respect to a power voltage.
  • an inverter capable of controlling an operating frequency comprising means for supplying a DC voltage, switching means for switching the voltage from the DC voltage supply means, parallel voltage resonance circuit including an inductor coupled in series with the switching means, the inductor and the switching means both being connected between both ends of the DC voltage supply means, and a resonance capacitor, voltage detecting means for detecting a voltage applied to the switching means, and control means for controlling a switching frequency of the switching means, on the basis of the result of comparison of a voltage value detected by the voltage detecting means and a predetermined reference voltage.
  • Fig. 1 shows a circuit arrangement of an apparatus for operating a discharge lamp to which a separately excited inverter according to an embodiment of the present invention is applied.
  • DC power source 102 makes up a power source circuit 101, which may be a pure DC power source such as a battery, or a rectifier circuit for rectifying an AC power source, which is of smoothing, partially smoothing, or nonsmoothing type.
  • Reference numeral 201 designates a voltage resonance circuit.
  • the primary winding 2021 of output transformer 202 of the leakage type is connected at one end to the positive terminal of DC power source 102 in the DC power source circuit 101, and at the other end to the collector of transistor 302 constituting switching element 301.
  • Resonance capacitor 204 is coupled in parallel with the primary winding 2021 of output transformer 202.
  • Load 12 such as a discharge lamp including a fluorescent lamp is connected across the secondary winding 2022 of output transformer 202.
  • Diode 14 is connected to the collector and emitter (negative terminal of DC power source 102) of transistor 302, in a back-to-back manner.
  • Voltage detector 401 is further connected between the collector and emitter of transistor 302.
  • Detector 401 includes a series circuit made up of diode 402 and capacitor 404.
  • Diode 402 is coupled with the collector-emitter path of transistor 302 in a forward direction.
  • This capacitor 404 is coupled in parallel with a series circuit including resistors 406 and 408. These resistors 406, 408 divide the voltage across capacitor 404, to form a voltage for application to control circuit 501 to be described later.
  • Control circuit 501 drives transistor 302 by a frequency depending on the collector-emitter voltage of transistor 302.
  • control circuit 501 is made up of reference voltage source 502, error amplifier 504, and voltage controlled oscillator (VCO) 506.
  • Amplifier 504 compares the output voltage of voltage detector 401 as appears at a node resistors 406 and 408, and the output voltage of reference voltage source 502, and produces a difference voltage as an error voltage.
  • VCO 506 oscillates at a frequency depending on the output signal of error amplifier 504.
  • VCO 506 may be an integrated circuit, such as NJM 555 manufactured by Shin-Nihon Musen Co., in Japan.
  • Fig. 2 shows a circuit arrangement including VCO 506 consisting of the above IC and its peripheral circuitry.
  • V+ designates a power source voltage for the IC
  • R A , R B , and C are resistors and capacitor with proper values.
  • DC power source 102 is turned on, VCO 506 starts oscillation, and the oscillator output signal is applied to the base of transistor 302.
  • Transistor 302 is alternately turned on and off at the oscillating frequency (f1) of VCO 506.
  • the output signal of transistor 302 drives parallel (voltage) resonance circuit 201 substantially consisting of primary winding 2021 of output transformer 202 and resonance capacitor 204, so that a high frequency output voltage is induced in the secondary winding 2022 of output transformer 202. In this way, the inverter is started up.
  • oscillating frequency f1 of VCO 506 is controlled so that voltage Vp/n is equal to reference voltage Vref as the output voltage of reference voltage source 502.
  • the voltage Vp/n is obtained by dividing the terminal voltage (Vp) of capacitor 404 by resistors 406 and 408, and "n" is a voltage dividing ratio.
  • Oscillating frequency f1 of VCO 506 is selected to be much higher than the resonance frequency (f0) of voltage resonance circuit 201, for example.
  • Figs. 10A through 10C showing variations of an input voltage to error amplifier 504, an input voltage to VCO 506, and an oscillating frequency of VCO 506. It is assumed that voltage Vp/n is lower than reference voltage Vref at time t1, as shown in Fig. 10A. As the voltage V CE across the collector-path of transistor 302 becomes high, voltage Vp/n gradually rises, and reaches reference voltage Vref at time t2, and exceeds during period from time t2 and t3. On the other hand, the input voltage to VCO 506 gradually decreases from time t1 and is zeroed at time t3, as shown in Fig. 10B. An oscillating frequency f1 of VCO 506 gradually increases from time t1 to t3 as its input voltage decreases. The increasing oscillating frequency f1 causes the collector-emitter voltage of transistor 302 to drop, and it is kept constant.
  • the inverter is applied for an apparatus for operating a discharge lamp, and it must be operated stably.
  • the output transformer of the leakage type is used.
  • the normal transformer, not leakage type transformer may be used.
  • FIG. 3 An embodiment of an apparatus for a discharge lamp to which a self-excited inverter is applied, will be described referring to Fig. 3.
  • like reference symbols are used for designate like or equivalent portions in Fig. 1, for simplicity.
  • voltage resonance circuit 202 is so arranged that one end of the primary winding 2061 of output transformer 206 is connected to the positive terminal of DC power source 102 of power source circuit 101, and the other end is connected to the collector of the transistor 302 of switching element 301.
  • Resonance capacitor 204 is coupled in parallel with the primary winding 2061 of output transformer 206.
  • Secondary winding 2062 of output transformer 206 is connected at one end to load 12 and at the other end to one end of the primary winding 5181 of feedback current transformer (CT) 518.
  • CT feedback current transformer
  • Control circuit 502 contains transistor 508 as an error detector.
  • the base of the transistor 508 is connected to a node between resistors 406 and 408 of voltage detector 401.
  • the emitter and collector of transistor 508 are respectively coupled with the positive and negative terminals of DC power source 102 of power source circuit 101.
  • Zener diode 514 as a reference potential source is inserted with its cathode connecting to the emitter of transistor 508.
  • the collector of transistor 508 is coupled with the gate of field effect transistor (FET) 516.
  • the source of FET 516 is connected to the negative terminal of DC power source 102.
  • Control circuit 502 further includes feedback current transformer (CT) 518.
  • CT feedback current transformer
  • One end of secondary winding 5182 is connected to the base of transistor 302 of switching element 301, while the other end, to the first ends of frequency control capacitors 520 and 522.
  • the capacitor 520 is coupled at the second end with the drain of FET 516, and capacitor 522 is coupled at the second end with the emitter of transistor 302.
  • Diode 14 is connected across the collector-emitter path of transistor 302.
  • Primary winding 5181 of feedback current transformer (CT) 518 is formed between the secondary winding 2062 of output transformer 206, and load 12.
  • CT feedback current transformer
  • Transistor 302 is in a slight conductive state by a base current fed from a start-up circuit (not shown). A slight current in turn flows through the primary winding 2061 of output transformer 206, so that a load current flows through the secondary winding 2062. This load current is detected by CT 518, and is fed back to the base of transistor 302. Transistor 302 is swiftly turned on through a route including the base of transistor 302, the emitter, capacitor 522 (FET 516 - capacitor 520), CT 518, and the base. The capacitors 520 and 522 are charged by the base current, which turns on transistor 302. Therefore, when transistor 302 is turned on, the base current gradually decreases.
  • transistor 302 rapidly turns off.
  • voltage resonance circuit 201 (including the primary winding 2061 of output transformer 206 and capacitor 204) resonates to induce an AC voltage in the secondary winding 2062 of output transformer 206.
  • the load current is inverted in polarity, and then its polarity is returned to the original polarity.
  • the voltage at the base of transistor 302 is positive in polarity, and the transistor 302 is turned on again through the positive feedback loop.
  • the inverter continuously oscillates the aid of the positive feedback loop and the voltage resonance.
  • the capacitor 404 of voltage detector 401 is charged at the voltage Vp based on the peak value of the collector voltage of transistor 302.
  • the voltage Vp is divided by resistors 406 and 408 at dividing ratio 1/n, and is applied to the base of transistor 508.
  • the emitter of transistor 508 has been biased at reference voltage Vref as the voltage across Zener diode 514. If voltage Vp is greater than "n" times a voltage difference between reference voltage Vref and voltage V BE across the base-emitter path of transistor 508 ( Vp > (Vref - V BE ) ⁇ n ), transistor 508 is turned off and FET 516 is in turn turned off.
  • capacitor 520 is disconnected an substantially only capacitor 522 is connected to the base of transistor 302. If voltage Vp is smaller than ( (Vref - V BE ) ⁇ n ), transistor 508 is active or in a conductive state. Under this condition, FET 516 functions as a variable impedance, and its impedance varies depending on the collector voltage of transistor 508.
  • the off period of transistor 302 as switching element 301 is determined by the resonance of resonance circuit 201, and is fixed.
  • the on period of transistor 302 is determined by the base current of transistor 302, which flows into capacitors 520 and 522.
  • the base current of transistor 302 is determined depending on capacitors 520 and 522, and impedance of FET 516, which are connected in series with the base of the transistor 302, through the primary winding 5181 of CT 518. Accordingly, an oscillating frequency (f1) of the inverter is variable by varying the impedance of FET 516.
  • the separately excited and self-excited inverters according to the first and second embodiments are each able to absorb a surge voltage applied to the power source voltage.
  • the surge voltage comes in, and it is applied to transistor 302
  • the surge is by-passed through the series circuit including diode 402 and capacitor 404.
  • a steep surge is by-passed by the series circuit of diode 402 and capacitor 404, so that the peak value of the collector voltage of transistor 302 is limited.
  • a gentle surge is also by-passed by the series circuit of diode 402 and capacitor 404, and if it is not completely by-passed, the remaining surge is absorbed through the above stabilizing operation. Therefore, the voltage applied across the collector-emitter path of transistor 302 can be limited.
  • the inverters are capable of preventing transistor 302 from being applied with an overvoltage, and protecting it against degradation and breakdown.
  • the ordinary inverter of this type is provided with an output transformer whose primary winding provides an inductance component in the voltage resonator.
  • the transformer provides an insulation between the primary and secondary sides.
  • the feedback loop may be formed by only the primary side of the output transformer. Therefore, there is no need for using an insulating means for the feedback loop.
  • Fig. 4 shows a circuit arrangement of a modification of the self-excited inverter of the second embodiment.
  • like reference symbols are used for designating like portions in Figs. 1 and 3.
  • a series circuit of diode 402 and capacitor 404 is connected across the collector-emitter path of transistor 302 of switching element 301.
  • Diode 402 is forwardly arranged with respect to transistor 302.
  • Capacitor 404 is connected in parallel with a series circuit of resistors 406 and 408 and varistor 410 such as a ceramic varistor.
  • a connection point of resistors 406 and 408 is connected to the base of transistor 508 in control circuit 502.
  • inverter using ceramic varistor 410 when a surge is superposed on the power source voltage, it serves as a pure varistor to absorb an over-voltage applied to transistor 302. In a stationary mode, it cooperates with capacitor 404 to serve as the peak detector. At this time, capacitor 404 is charged up to a peak value of the collector voltage of transistor 302.
  • the other operation of the present embodiment is similar to that of the second embodiment, and hence no further description will be given.
  • Fig. 5 shows a circuit arrangement of a fourth embodiment of the present invention, in which the output transformer is a normal transformer, not the leakage type transformer.
  • like reference symbols designate like or equivalent portions in Figs. 1 and 3, and the construction and a basic operation of the fourth embodiment will not be described, for simplicity.
  • control circuit 503 includes transistor 508 whose base is connected to a node between resistors 406 and 408.
  • the emitter and controller of transistor 508 are connected through resistors 510 and 512 to the positive and negative terminals of DC power source 102, respectively.
  • Zener diode 514 as a reference voltage source is inserted, with its cathode connecting to the emitter of transistor 508.
  • the collector of transistor 508 is connected to the gate of FET 516, and the source of FET 516 is connected to the negative terminal of DC power source 102.
  • Control circuit 503 also contains feedback winding 524 of the current feedback type, which is magnetically coupled with the primary winding of the output transformer 206.
  • One end of that winding is connected to the base of transistor 302, and to the first ends of capacitors 520 and 522 for frequency control.
  • the second end of capacitor 520 is coupled with the drain of FET 516, and the second end of capacitor 522, with the emitter of transistor 302.
  • a series circuit substantially consisting of die 526 and resistor 528 is coupled between the base and emitter of transistor 302, with the cathode connecting to the base.
  • Diode 14 is connected between the collector and emitter of the same.
  • ballast 16 is inserted in a series circuit including the secondary winding 2062 of output transformer 206 and load 12.
  • DC power source 102 is turned on, and transistor 302 is turned on.
  • the primary winding 2061 of output transformer 206 is slightly driven, so that a load current flows through secondary winding 2062.
  • the base current flows from the transistor 302 into capacitors 520 and 522 to charge these capacitors.
  • capacitors 520 and 522 are discharged and the current flows into transformer 524, through resistor 528 and diode 526.
  • the resonance operation of voltage resonance circuit 202 causes an AC voltage to induce in the secondary winding 2062 of output transformer 206.
  • the polarity of the load current is inversed, and then is reversed.
  • transistor 302 is turned on again by the positive voltage applied to the base. In this way, the inverter oscillates.
  • Fig. 6 shows a fifth embodiment of the present invention, in which a ballast with a feedback winding is used in the load side of the circuit.
  • like reference symbols designate like or equivalent portions in Figs. 1, 3 through 5, and the construction and a basic operation of the fifth embodiment will not be described, for simplicity.
  • transformer 524 in control circuit 503 shown in Fig. 5 is replaced by the ballast with a feedback winding.
  • One end of the secondary winding 5302 of ballast 530 with a feedback winding is connected to the base of transistor 302, and the other end to one end of capacitor 522 for frequency control.
  • the primary winding 5301 is inserted in a series circuit of the secondary winding 2062 of output transformer 206 and load 12. The remaining circuit arrangement is substantially the same as that of the fourth embodiment, and hence no further description will be given.
  • the inverter operates like the second embodiment.
  • a positive feedback loop is routed from the base of transistor 302, through the emitter, capacitor 522 (FET 516 - capacitor 520), and ballast 530, to the base.
  • current flows from transistor 302 to capacitors 520 and 522 to charge these capacitors.
  • capacitors 520 and 522 are discharged and current flows through resistor 528 and diode 526, and into the secondary winding 5302 of ballast 530.
  • voltage resonance circuit 202 resonates to induce an AC voltage in the secondary winding 2062 of transformer 206.
  • the load current is inverted in polarity and its polarity is returned to the original one.
  • the voltage applied to the base of transistor 302 becomes positive in polarity again, and the transistor 302 is turned on. In this way, the inverter of this embodiment oscillates.
  • resistor may be used for the capacitor.
  • control circuit 505 uses transistor 508 for an error detector.
  • the base of the transistor 508 is connected to a node between resistors 406 and 408.
  • the emitter of transistor 508 is connected through resistor 510 to the positive terminal of DC power source 102, the collector of transistor 508, to the base of transistor 534 via resistor 532.
  • a Zener diode 514 as a reference voltage source is placed with the cathode connecting to the emitter of transistor 508.
  • Transistor 508 is connected at the collector to the negative terminal of DC power source 102.
  • Diode 536, and resistors 538 and 540 are connected in series between the base and emitter of that transistor, as shown.
  • Diode 542 is forwardly connected between the connection point of resistors 538 and 540 and the base of transistor 302.
  • Control circuit 505 further includes ballast 530 with a feedback winding.
  • Ballast 530 is connected at one end of its secondary winding 5302 with the base of transistor 302, and at the other end of its secondary winding 5302 with one end of capacitor 544.
  • the other end of capacitor 544 is connected to the collector of transistor 534.
  • Diode 14 is connected between the collector and emitter of transistor 302.
  • a series circuit of the secondary winding 2062 of output transformer 206 and load 12 contains the primary winding 5301 of ballast 530.
  • the remaining circuit arrangement of the present embodiment is substantially the same as that of the fifth embodiment, and further description of it will be omitted.
  • capacitor 544 is charged by the base current, and at the termination of charging the capacitor, the base current of transistor 302 decreases to zero and the transistor is turned off.
  • the discharge current flows out of capacitor 544, and goes through the collector and emitter of transistor 534, and resistor 540 and diode 542, and reaches the secondary winding 5302 of ballast 530.
  • the resonating operation of voltage resonance circuit 202 induces an AC voltage in the secondary winding 2062 of output transformer 206.
  • the induction of the AC voltage inverts the polarity of the load current, and returns it to the original one. In turn, the voltage applied to the base becomes positive in polarity, and it turns on transistor 302 again.
  • the frequency control is based on a discharge time constant of capacitor 544.
  • a potential applied to the resistors 406 and 408 of voltage detector 401 varies with the collector-emitter voltage of transistor 302.
  • a conduction state of transistor 508 changes, and the collector current of transistor 508 changes.
  • a current flowing into diode 536 and resistor 538 also changes. Consequently, a base potential of transistor 534 also changes.
  • an equivalent resistance of transistor 534 is large, an amount of discharge from capacitor 544 is lessened during the off period of transistor 302, and an amount of base current flowing during the on period of transistor 302 decreases. Conversely, if it is small, the base current increases.
  • the on period of transistor 302 is determined by an amount (period) of the base current of transistor 302, and the base current is determined by transistor 534 and resistor 540, as described above. Accordingly, an oscillating frequency of the inverter may be varied by varying the impedance of transistor 534.
  • Fig. 8 shows a seventh embodiment of the present invention, in which the ballast with a feedback winding in the fifth embodiment is replaced by a feedback current transformer of the saturable type.
  • like reference symbols designate like or equivalent portions in Figs. 1, 3 through 7, and the construction and a common operation of the fourth embodiment will not be described, for simplicity.
  • Control circuit 506 in Fig 8 uses a feedback current transformer (CT) of the saturable type in place of the ballast 530 in the control circuit of the fifth embodiment.
  • Saturable type CT 546 for causing transistor 302 to oscillate in a self-excited mode is so arranged that the secondary winding 5462 of CT 546 is connected at one end to the base of transistor 302, and at the other end to one end of capacitor 522 for frequency control.
  • Diode 548 is coupled across the secondary winding 5462 of CT 546, with its cathode connecting to the base of transistor 302.
  • output transformer 202 of the leakage type is used in place of output transformer 206.
  • Capacitor 204 is connected across the primary winding 2021.
  • a series circuit of the secondary winding 2022 of output transformer 202 of the leakage type, load (discharge lamp) 12, and start-up capacitor 550 contains the primary winding 5461 of saturable type CT 546.
  • Start-up capacitor 550 resonates mainly with a leakage inductance of output transformer 202, before discharge lamp (load) 12 lights on. A high voltage is generated by the resonance, and lights on the discharge lamp. After the lamp lights on, an equivalent resistance is inserted in the resonance circuit of capacitor 550 and transformer 202, and therefore, the resonating operation is stopped. Further, before the lamp lights on, a current enough to preheat the filament is fed, and it is limited to a proper value after the lamp lights on.
  • Start-up resistor 552 is connected to the base of transistor 302 and the positive terminal of DC power source 102.
  • Frequency control capacitor 522 is couples in parallel with diode 554 whose polarity is arranged with respect to the capacitor, as shown.
  • the remaining circuit arrangement is substantially the same as that of the fifth embodiment, and hence no further description will be given.
  • FIG. 9 An eighth embodiment of the present invention will be described with reference to Fig. 9.
  • the present embodiment is equivalent to a case that the ballast with a feedback winding in the Fig. 7 embodiment is substituted by a feedback current transformer of the saturable type.
  • like reference symbols designate like or equivalent portions in Figs. 1, 3 through 8, and the construction and a basic operation of the fourth embodiment will not be described, for simplicity.
  • a feedback current transformer (CT) of the saturable type is used in place of the ballast in the Fig. 7 control circuit.
  • Saturable type CT 546 is arranged such that the secondary winding 5462 is connected at one end to the base of transistor 302, and at the other end to one end of capacitor 544 for frequency control.
  • Diode 548 is connected across the secondary winding 5462 of CT 546, with its cathode connecting to the base of transistor 302.
  • Output transformer 202 is of the leakage type, with its primary winding 2021 coupled across capacitor 204.
  • a series circuit of the secondary winding 2022 of output transformer and load 12 contains the primary winding 5461 of CT 546.

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  • Circuit Arrangements For Discharge Lamps (AREA)
  • Inverter Devices (AREA)

Claims (19)

  1. Un onduleur capable de commander une fréquence de fonctionnement, comprenant :
       des moyens d'alimentation pour fournir une tension continue;
       un circuit à résonance de tension parallèle, comprenant une inductance et un condensateur résonnant, cette inductance et ce condensateur résonnant étant connectés à une extrémité des moyens d'alimentation à tension continue; et
       des moyens de commutation pour commuter la tension provenant des moyens d'alimentation à tension continue, ces moyens de commutation et ce circuit à résonance de tension parallèle étant connectés mutuellement en série et étant connectés à une autre extrémité des moyens d'alimentation à tension continue;
       caractérisé en ce qu'il comprend en outre
       des moyens de détection de tension (40₁) destinés à détecter une tension qui est appliquée aux moyens de commutation (30₁); et
       des moyens de commande (50₁) destinés à commander une fréquence de commutation des moyens de commutation (30₁), sur la base du résultat de la comparaison d'une valeur de tension détectée par les moyens de détection de tension (40₁) et d'une tension de référence prédéterminée.
  2. L'onduleur selon la revendication 1, caractérisé en ce que les moyens de commande (50₁) comprennent des moyens de comparaison destinés à comparer la tension détectée par les moyens de détection de tension (40₁), et des moyens de commande de fréquence pour commander un signal de commutation devant être appliqué aux moyens de commutation (40₁), conformément au résultat de la comparaison qui est effectuée par les moyens de comparaison.
  3. L'onduleur selon la revendication 2, caractérisé en ce que les moyens de détection de tension (40₁) comprennent un circuit série qui comporte une diode (402) et un condensateur (404), ce circuit série étant connecté en parallèle avec les moyens de commutation (30₁), et une tension présente aux bornes de ce condensateur chargé est détectée.
  4. L'onduleur selon la revendication 3, caractérisé en ce que les moyens de commutation (30₁) comprennent un transistor de commutation (302).
  5. L'onduleur selon la revendication 4, caractérisé en ce que les moyens de détection de tension (40₁) comprennent en outre des résistances (406, 408) connectées aux bornes du condensateur de détection (404), pour diviser la tension précitée, et les moyens de comparaison comparent la tension divisée et la tension de référence.
  6. L'onduleur selon la revendication 5, caractérisé en ce que les moyens de comparaison comportent un amplificateur d'erreur (54), et les moyens de commande de fréquence comportent un oscillateur commandé par tension (506).
  7. L'onduleur selon la revendication 5, caractérisé en ce que les moyens de comparaison (504) comprennent un transistor de comparaison (508) dont une base est connectée à un noeud entre les résistances de division de tension (406, 408), un émetteur et un collecteur sont respectivement connectés aux deux extrémités des moyens d'alimentation à tension continue (10₁), et une diode Zener (514) connectée en inverse entre l'émetteur et le collecteur du transistor de comparaison (508), cette diode Zener (514) fournissant la tension de référence, et les moyens de commande de fréquence comprennent un transformateur d'intensité de rétroaction (518) dont l'enroulement secondaire (518₂) est connecté, à une première extrémité, à la base du transistor de commutation (302), et des moyens de variation de fréquence connectés entre le collecteur du transistor de comparaison (508) et la seconde extrémité du transistor de comparaison (508).
  8. L'onduleur selon la revendication 7, caractérisé en ce que les moyens de variation de fréquence comprennent un premier condensateur de commande de fréquence (520), un élément à impédance variable d'un transistor à effet de champ (516) dont une grille est connectée au collecteur du transistor de comparaison (508), un drain est connecté à une première extrémité du premier condensateur de commande de fréquence (520), et une source est connectée à la seconde extrémité des moyens d'alimentation à tension continue (10₁), et un second condensateur de commande de fréquence (522), connecté à la seconde extrémité du premier condensateur de commande de fréquence (520) et à la source du transistor à effet de champ (516).
  9. L'onduleur selon la revendication 5, caractérisé en ce que les moyens de comparaison comprennent un transistor de comparaison (508) dont une base est connectée à un noeud entre les résistances de division de tension (406, 408), un émetteur et un collecteur sont respectivement connectés aux deux extrémités des moyens d'alimentation à tension continue (10₁), et une diode Zener (514) connectée en inverse entre l'émetteur et le collecteur du transistor de comparaison (508), cette diode Zener (514) fournissant la tension de référence, et les moyens de commande de fréquence comprennent un enroulement de rétroaction (524) qui est en couplage magnétique, à une première extrémité, avec la base du transistor de commutation (302), des moyens de variation de fréquence connectés entre le collecteur du transistor de comparaison (508) et la seconde extrémité de l'enroulement de rétroaction (524), et un circuit série de rétroaction connecté entre la base et l'émetteur du transistor de commutation (302), ce circuit série comprenant une résistance et une diode (526) dont la polarité est orientée de façon inverse par rapport au transistor de commutation (302).
  10. L'onduleur selon la revendication 9, caractérisé en ce que les moyens de variation de fréquence comprennent un premier condensateur de commande de fréquence (520), un élément à impédance variable d'un transistor à effet de champ (516) dont une grille est connectée au collecteur du transistor de comparaison (508), un drain est connecté à une première extrémité du premier condensateur de commande de fréquence (520), et une source est connectée à la seconde extrémité des moyens d'alimentation à tension continue (10₁), et un second condensateur de commande de fréquence (522) connecté à la seconde extrémité du premier condensateur de commande de fréquence (520) et à la source du transistor à effet de champ (302).
  11. L'onduleur selon la revendication 5, caractérisé en ce que les moyens de comparaison comprennent un transistor de comparaison (508) dont une base est connectée à un noeud entre les résistances de division de tension (406, 408), un émetteur et un collecteur sont respectivement connectés aux deux extrémités des moyens d'alimentation à tension continue (10₁) et une diode Zener (514) connectée en inverse entre l'émetteur et le collecteur du transistor de comparaison (508), cette diode Zener (514) produisant la tension de référence, et les moyens de commande de fréquence comprennent un enroulement secondaire (530₂) d'un transistor de régulation (530) ayant un enroulement de rétroaction, l'enroulement secondaire précité (530₂) étant connecté, à une première extrémité, à la base du transistor de commutation (302), des moyens de variation de fréquence connectés entre le collecteur du transistor de comparaison (508) et la seconde extrémité de l'enroulement secondaire (530₂) du transformateur de régulation (530), et un circuit série de rétroaction connecté entre la base et l'émetteur du transistor de commutation (302), ce circuit série comprenant une résistance (528) et une diode (526) dont la polarité est orientée de façon inversée par rapport au transistor de commutation (302).
  12. L'onduleur selon la revendication 9, caractérisé en ce que les moyens de variation de fréquence comprennent un premier condensateur de commande de fréquence (520), un élément à impédance variable d'un transistor à effet de champ (516) dont une grille est connectée au collecteur du transistor de comparaison (508), un drain est connecté à une première extrémité du premier condensateur de commande de fréquence (520), et une source est connectée à la seconde extrémité des moyens d'alimentation à tension continue (10₁), et un second condensateur de commande de fréquence (522) connecté entre la seconde extrémité du premier condensateur de commande de fréquence (520) et la source du transistor à effet de champ (516).
  13. L'onduleur selon la revendication 5, caractérisé en ce que les moyens de comparaison comprennent un transistor de comparaison dont une base est connectée à un noeud se trouvant entre les résistances de division de tension (406, 408), un émetteur et un collecteur sont respectivement connectés à une première extrémité des moyens d'alimentation à tension continue (10₁), et un point de connexion prédéterminé, et une diode Zener (514) connectée en inverse entre l'émetteur et le collecteur du transistor de comparaison (508), cette diode Zener (514) produisant la tension de référence, et les moyens de commande de fréquence comprennent un enroulement secondaire (530₂) d'un transformateur de régulation (530) ayant un enroulement de rétroaction, cet enroulement secondaire (530₂) étant connecté, à une première extrémité, à la base du transistor de commutation (302), et des moyens de variation de fréquence connectés entre une seconde extrémité des moyens d'alimentation à tension continue (10₁), et le point de connexion précité et l'enroulement secondaire (530₂) du transformateur de régulation (530).
  14. L'onduleur selon la revendication 13, caractérisé en ce que les moyens de variation de fréquence comprennent un transistor de commande (534) dont la base est connectée au point de connexion précité, et le collecteur est connecté aux moyens d'alimentation à tension continue (10₁), une résistance de commande (538) et une diode (536) étant connectées entre la base et l'émetteur de ce transistor de commande (534), une diode de rétroaction (542) connectée entre la résistance de commande (538) et la première extrémité de l'enroulement secondaire (530₂) du transformateur de régulation (530), et un condensateur de commande de fréquence connecté entre la seconde extrémité de l'enroulement secondaire (530₂) du transformateur de régulation (530) et la seconde extrémité des moyens d'alimentation à tension continue (10₁).
  15. L'onduleur selon la revendication 5, caractérisé en ce que les moyens de comparaison comprennent un transistor de comparaison (508) dont une base est connectée à un noeud entre les résistances de division de tension (406, 408), un émetteur et un collecteur sont respectivement connectés aux deux extrémités des moyens d'alimentation à tension continue (10₁), et une diode Zener (514) connectée en inverse entre l'émetteur et le collecteur du transistor de comparaison (508), cette diode Zener (514) produisant la tension de référence, et les moyens de commande de fréquence comprennent un enroulement secondaire (546₂) d'un transformateur d'intensité de rétroaction saturable (546) qui est connecté, à une première extrémité, à la base du transistor de commutation (302), une diode (548) connectée en parallèle sur l'enroulement secondaire (546₂) du transformateur d'intensité de rétroaction saturable (546), des moyens de variation de fréquence connectés entre le collecteur du transistor de comparaison (508) et la seconde extrémité de l'enroulement secondaire (546₂) du transformateur (546), et un circuit série de rétroaction connecté entre la base et l'émetteur du transistor de commutation (302), ce circuit série comprenant une résistance (528) et une diode (526) qui a une polarité orientée de façon inversée par rapport au transistor de commutation (302).
  16. L'onduleur selon la revendication 15, caractérisé en ce que les moyens de variation de fréquence comprennent un premier condensateur de commande de fréquence (520), un élément à impédance variable d'un transistor à effet de champ (516) dont une grille est connectée au collecteur du transistor de comparaison (508), un drain est connecté à une première extrémité du premier condensateur de commande de fréquence (520), et une source est connectée à la seconde extrémité des moyens d'alimentation à tension continue (10₁), et un second condensateur de commande de fréquence (522) connecté à la seconde extrémité du premier condensateur de commande de fréquence (520) et à la source du transistor à effet de champ (516).
  17. L'onduleur selon la revendication 5, caractérisé en ce que les moyens de comparaison comprennent un transistor de comparaison (508) dont une base est connectée à un noeud entre les résistances de division de tension (406, 408), un émetteur et un collecteur sont respectivement connectés à une première extrémité des moyens d'alimentation à tension continue (10₁) et à un point de connexion prédéterminé, et une diode Zener (514) connectée en inverse entre l'émetteur et le collecteur du transistor de comparaison (508), cette diode Zener (514) produisant la tension de référence, et les moyens de commande de fréquence comprennent un enroulement secondaire (546₂) d'un transformateur d'intensité de rétroaction saturable (546) dont une première extrémité est connectée à la base du transistor de commutation (302), une diode (548) connectée en parallèle sur l'enroulement secondaire (546₂) du transformateur d'intensité de rétroaction saturable (546), et des moyens de variation de fréquence connectés entre une seconde extrémité des moyens d'alimentation à tension continue (10₁), et le point de connexion précité et l'enroulement secondaire (546₂) du transformateur (546).
  18. L'onduleur selon la revendication 17, caractérisé en ce que les moyens de variation de fréquence comprennent un transistor de commande (534) dont la base est connectée au point de connexion précité, et le collecteur est connecté aux moyens d'alimentation à tension continue (10₁), une résistance de commande (538) et une diode (536) étant connectées entre la base et l'émetteur du transistor de commande (534), une diode de rétroaction (542) connectée entre la résistance de commande (538) et la première extrémité de l'enroulement secondaire (546₂) du transformateur (546), et un condensateur de commande de fréquence (544) connecté entre la seconde extrémité de l'enroulement secondaire (546₂) du transformateur (546) et la seconde extrémité des moyens d'alimentation à tension continue (10₁).
  19. L'onduleur selon la revendication 5, caractérisé en ce qu'il comprend en outre une varistance en céramique (410) pour absorber une surtension transitoire, connectée en parallèle avec le condensateur de détection (404).
EP19890105131 1989-03-22 1989-03-22 Onduleur capable de contrôler la fréquence de service Expired - Lifetime EP0388492B1 (fr)

Priority Applications (2)

Application Number Priority Date Filing Date Title
EP19890105131 EP0388492B1 (fr) 1989-03-22 1989-03-22 Onduleur capable de contrôler la fréquence de service
DE1989612764 DE68912764T2 (de) 1989-03-22 1989-03-22 Arbeitsfrequenzbestimmender Wechselrichter.

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
EP19890105131 EP0388492B1 (fr) 1989-03-22 1989-03-22 Onduleur capable de contrôler la fréquence de service

Publications (2)

Publication Number Publication Date
EP0388492A1 EP0388492A1 (fr) 1990-09-26
EP0388492B1 true EP0388492B1 (fr) 1994-01-26

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EP19890105131 Expired - Lifetime EP0388492B1 (fr) 1989-03-22 1989-03-22 Onduleur capable de contrôler la fréquence de service

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EP (1) EP0388492B1 (fr)
DE (1) DE68912764T2 (fr)

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2780177B2 (ja) * 1988-09-13 1998-07-30 東芝ライテック株式会社 放電灯点灯装置

Family Cites Families (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
FR2503954A1 (fr) * 1981-04-09 1982-10-15 Sefli Procede de decoupage essentiellement sinusoidal d'une tension continue avec regulation et dispositif pour sa mise en oeuvre
DE3303374A1 (de) * 1983-02-02 1984-08-02 Rheintechnik Weiland & Kaspar Kg, 6680 Neunkirchen Stromversorgungsschaltung fuer leuchtstoffroehren
DE3315481A1 (de) * 1983-04-28 1984-10-31 Innovatron Krauss & Co., Feldbrunnen-St. Niklaus, Solothurn Leuchtvorrichtung mit einem oszillator, einer leistungsstufe und einer gasentladungslampe sowie verfahren zum betreiben einer gasentladungslampe

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Publication number Publication date
DE68912764T2 (de) 1994-05-11
EP0388492A1 (fr) 1990-09-26
DE68912764D1 (de) 1994-03-10

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