WO2022237431A1 - 谐振变换器、电源、终端和升压方法 - Google Patents

谐振变换器、电源、终端和升压方法 Download PDF

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Publication number
WO2022237431A1
WO2022237431A1 PCT/CN2022/086351 CN2022086351W WO2022237431A1 WO 2022237431 A1 WO2022237431 A1 WO 2022237431A1 CN 2022086351 W CN2022086351 W CN 2022086351W WO 2022237431 A1 WO2022237431 A1 WO 2022237431A1
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WO
WIPO (PCT)
Prior art keywords
voltage
power supply
secondary winding
output
charging
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PCT/CN2022/086351
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English (en)
French (fr)
Inventor
郭红光
张晨松
李建国
田晨
张加亮
张锦
Original Assignee
Oppo广东移动通信有限公司
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Publication of WO2022237431A1 publication Critical patent/WO2022237431A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J7/00—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries
    • H02J7/02—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries for charging batteries from ac mains by converters
    • H02J7/04—Regulation of charging current or voltage
    • H02J7/06—Regulation of charging current or voltage using discharge tubes or semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
    • H02M7/068—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode mounted on a transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
    • H02M7/08—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode arranged for operation in parallel
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/23—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only arranged for operation in parallel
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J2207/00—Indexing scheme relating to details of circuit arrangements for charging or depolarising batteries or for supplying loads from batteries
    • H02J2207/20—Charging or discharging characterised by the power electronics converter
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present application relates to the technical field of charging, in particular to a resonant converter, a power supply, a terminal and a voltage boosting method.
  • the LLC resonant converter has been widely used in medium and high-power power supply products due to its soft switching technology, low switching loss, and high power density design.
  • the main components of the LLC resonant converter include winding transformers and energy storage inductors, and the volume of the winding transformer is large, resulting in the large circuit volume of the entire system of the power supply product using the LLC resonant converter.
  • reducing the volume and increasing the power density is the trend of power supply development. Based on this, the volume of the power supply product can be reduced by reducing the volume of the winding transformer, and the voltage doubler circuit is used to ensure the reduction of the transformer volume while avoiding the output power. reduce.
  • an embodiment of the present application provides a resonant converter, the resonant converter includes a winding transformer and a step-up circuit; the winding transformer includes at least one set of secondary windings, and each set of secondary windings includes a first secondary winding and a second secondary winding. Two secondary windings; the first secondary winding is connected in series with the second secondary winding;
  • the winding transformer is used to transform the voltage provided by the power supply, and output the transformed voltage from the first secondary winding in the positive half cycle of the voltage provided by the power supply, and convert the transformed voltage to the voltage provided by the power supply
  • the negative half cycle is output from the second secondary winding
  • the boost circuit is used to boost the voltage output by the first secondary winding during the positive half period of the voltage provided by the power supply, and boost the voltage output by the second secondary winding during the negative half period of the voltage provided by the power supply.
  • the boost circuit corresponding to each group of secondary windings includes a first charging and discharging circuit and a second charging and discharging circuit; the first end of the first charging and discharging circuit is connected to the first end of the first secondary winding connected, the first end of the second charging and discharging circuit is connected to the second end of the second secondary winding, and the second end of the first charging and discharging circuit and the second end of the second charging and discharging circuit are both grounded;
  • the first charging and discharging circuit is used for charging according to the voltage output by the first secondary winding in the negative half cycle of the voltage provided by the power supply, and discharging in the positive half cycle to boost the voltage output by the first secondary winding;
  • the second charging and discharging circuit is used for charging according to the voltage output by the second secondary winding in the positive half cycle of the voltage provided by the power supply, and discharging in the negative half cycle to boost the voltage output by the second secondary winding.
  • the above-mentioned first charging and discharging circuit includes a first capacitor, a first rectifier and a second rectifier; the first end of the first capacitor is connected to the first end of the first secondary winding, and the first capacitor The second end of the first rectifier is connected to the first end of the first rectifier, and the second end of the first rectifier is the output end of the resonant converter; the first end of the second rectifier is grounded, and the second end of the second rectifier is connected to the The second end of the first capacitor is connected to the first end of the first rectifier tube;
  • the first capacitor is charged according to the voltage output by the first secondary winding during the negative half period of the voltage provided by the power supply, and discharged during the positive half period of the voltage provided by the power supply;
  • the first rectifier is turned on during the positive half cycle of the voltage provided by the power supply, and turned off during the negative half cycle of the voltage provided by the power supply;
  • the second rectifier tube is turned off during the positive half period of the voltage provided by the power supply, and turned on during the negative half period of the voltage provided by the power supply.
  • the first rectifier and the second rectifier are both diodes; or, the first rectifier and the second rectifier are both switch tubes.
  • the second charging and discharging circuit includes a second capacitor, a third rectifier and a fourth rectifier; the first end of the second capacitor is connected to the second end of the second secondary winding, and the second capacitor The second end of the fourth rectifier is connected to the first end of the fourth rectifier, and the second end of the fourth rectifier is connected to the output end of the resonant converter; the first end of the third rectifier is grounded, and the second end of the third rectifier is connected to the The second end of the second capacitor is connected to the first end of the fourth rectifier tube;
  • the second capacitor is charged according to the voltage output by the second secondary winding during the positive half period of the voltage provided by the power supply, and discharged during the negative half period of the voltage provided by the power supply;
  • the third rectifier is turned on in the positive half cycle of the voltage provided by the power supply, and turned off in the negative half cycle of the voltage provided by the power supply;
  • the fourth rectifier tube is turned off during the positive half period of the voltage provided by the power supply, and turned on during the negative half period of the voltage provided by the power supply.
  • the third rectifier and the fourth rectifier are both diodes; or, both the third rectifier and the fourth rectifier are switch tubes.
  • the boost circuit further includes a third charging and discharging circuit; the first terminal of the third charging and discharging circuit is connected to the second terminal of the first charging and discharging circuit, and the second terminal of the third charging and discharging circuit is grounded ;
  • the third charging and discharging circuit is used to boost the output voltage of the first secondary winding by superimposing with the first charging and discharging circuit in the positive half cycle, and superimposing the second secondary winding with the second charging and discharging circuit in the negative half cycle of the power supply.
  • the output voltage of the side winding is boosted.
  • the third charging and discharging circuit includes a third capacitor; the first terminal of the third capacitor is connected to the second terminal of the first charging and discharging circuit, and the second terminal of the third capacitor is grounded.
  • an embodiment of the present application provides a power supply, which includes the resonant converter in any embodiment of the first aspect above.
  • embodiments of the present application provide a terminal, where the terminal includes the power supply in any embodiment of the second aspect above.
  • an embodiment of the present application provides a voltage boosting method, the method including:
  • the first voltage is boosted during the positive half period of the voltage provided by the power supply, and the second voltage is boosted during the negative half period of the voltage provided by the power supply.
  • An embodiment of the present application provides a resonant converter, a power supply, a terminal and a boosting method
  • the resonant converter includes a winding transformer and a boosting circuit
  • the winding transformer includes at least one set of secondary windings, and each set of secondary windings includes a first The secondary winding and the second secondary winding
  • the first secondary winding is connected in series with the second secondary winding
  • the winding transformer transforms the voltage provided by the power supply, and the transformed voltage is in the positive half cycle of the voltage provided by the power supply
  • the voltage is boosted, and the voltage output by the second secondary winding is boosted during the negative half cycle of the voltage provided by the power supply.
  • the step-up circuit By setting the step-up circuit to boost the output voltage of the first secondary winding and the second secondary winding of the transformer to ensure that the voltage amplitude output from the output terminal to the load remains unchanged, and then increase the winding between the primary winding and the secondary winding of the transformer
  • the turn ratio avoids the reduction of output power, so as to reduce the volume of the winding transformer by reducing the cross-sectional area of the magnetic core, thereby reducing the volume of the power supply product; at the same time, each half cycle, the first secondary winding of the winding transformer and the second
  • the voltage output by the secondary winding can be boosted by the booster circuit, so that the first secondary winding and the second secondary winding have secondary windings to supply power to the load in every half cycle, so that the output of full-wave rectification can be achieved.
  • the effect is to greatly reduce the ripple of the output voltage of the power supply. Therefore, the resonant converter provided by the embodiment of the present application can reduce the volume of the power supply product and at the same time
  • Fig. 1 is a schematic structural diagram of a resonant converter in an embodiment
  • FIG. 2 is a schematic structural diagram of a resonant converter in another embodiment
  • FIG. 3 is a schematic structural diagram of a resonant converter in another embodiment
  • FIG. 4 is a schematic structural diagram of a resonant converter in another embodiment
  • FIG. 5 is a schematic structural diagram of a resonant converter in another embodiment
  • FIG. 6 is a schematic structural diagram of a resonant converter in another embodiment
  • Fig. 7 is a schematic structural diagram of a resonant converter in another embodiment
  • Fig. 8 is a schematic structural diagram of a resonant converter in another embodiment
  • Fig. 9 is a schematic structural diagram of a resonant converter in another embodiment.
  • Fig. 10 is a schematic structural diagram of a resonant converter in another embodiment
  • Fig. 11 is a schematic structural diagram of a resonant converter in another embodiment
  • Fig. 12 is a schematic structural diagram of a resonant converter in another embodiment
  • Fig. 13 is a schematic structural diagram of a resonant converter in another embodiment
  • Fig. 14 is a schematic structural diagram of a resonant converter in another embodiment
  • Fig. 15 is a schematic flow chart of a boosting method in an embodiment.
  • Boost circuit 20: Boost circuit; 201: The first charging and discharging circuit;
  • 2011 The first rectification circuit
  • 2012 The second rectification circuit
  • connection and “connection” mentioned in this application all include direct and indirect connection (connection) unless otherwise specified.
  • orientation or positional relationship indicated by the orientation words such as “upper” and “lower” is based on the orientation or positional relationship shown in the drawings, and is only for the convenience of describing the application and simplifying the Describe, but do not indicate or imply that the device or element referred to must have a particular orientation, be constructed in a particular orientation, and operate in a particular orientation, and thus should not be construed as limiting the application.
  • a first feature being "on” or “under” a second feature may mean that the first and second features are in direct contact, or that the first and second features are indirect through an intermediary. touch.
  • the difference in names is not used as a way to distinguish elements, but the difference in function of elements is used as a principle of distinction.
  • the embodiment of the present application provides a resonant converter
  • the resonant converter includes a winding transformer 10, a booster circuit 20
  • the winding transformer 10 includes at least one set of secondary windings, and each set of secondary windings includes a second A secondary winding 101 and a second secondary winding 102; the first secondary winding 101 and the second secondary winding 102 are connected in series;
  • the winding transformer 10 is used to transform the voltage provided by the power supply and transform the transformed voltage
  • the positive half cycle of the voltage provided by the power supply is output from the first secondary winding 101, and the transformed voltage is output from the second secondary winding 102 in the negative half cycle of the voltage provided by the power supply;
  • the booster circuit 20 is used for The voltage output by the first secondary winding 101 is boosted during the positive half period of the voltage provided by the power supply, and the voltage output by the second secondary winding 102 is boosted during the negative half period of the voltage provided by the power supply.
  • the winding transformer may include multiple sets of secondary windings, each set of secondary windings includes two secondary windings in series, and each set of secondary windings corresponds to a boost circuit.
  • FIG. 1 shows a group of secondary windings.
  • winding transformer 10 in Fig. 1 comprises primary winding 103 and first secondary winding 101 and second secondary winding 102 (first secondary winding 101 and second secondary winding 102 are a group secondary winding).
  • Both the first terminal S1 of the first secondary winding 101 and the second terminal S4 of the second secondary winding 102 are connected to the booster circuit 20, the second terminal S2 of the first secondary winding 101 and the second terminal S2 of the second secondary winding 102
  • the first terminal S3 is connected in series and grounded.
  • the winding transformer 10 is used to transform the voltage provided by the power supply.
  • the voltage provided by the power supply enters from the primary winding 103 of the winding transformer 10 and outputs from the secondary winding of the winding transformer 10 .
  • the winding transformer 10 obtains the output voltage of the secondary winding according to the input voltage of the primary winding and the turn ratio between the primary winding and the secondary winding.
  • the turn ratio between the primary winding 103 and the secondary winding is 10:1, assuming that the input voltage of the primary winding 103 is 10V, the output voltage of the secondary winding is 1V. It should be emphasized that the number of coil turns of the primary winding and the secondary winding in Fig. 1 is only for illustration and is not used to limit the actual number of coil turns.
  • the primary side is connected to the power supply, that is, the power supply side
  • the secondary side is connected to the load, that is, the load side
  • the primary side input and secondary side output of the transformer are both alternating current
  • the waveform of alternating current can be a sine , triangle wave, square wave, etc. Therefore, the output voltage of the secondary winding of the winding transformer 10 in the embodiment of the present application is divided into a positive half cycle and a negative half cycle.
  • the relationship between the winding mode of the primary coil (primary side) and the secondary coil (secondary side) of the transformer determines the same-named end and the different-named end of the transformer.
  • the starting point of the two windings is the end of the same name, but if the direction of the two windings is opposite, the starting point of one winding and the ending point of the other winding are the same name end.
  • the end with the same name refers to the end with the same potential polarity in two (or more than two) windings at any time under the action of the same alternating magnetic flux, such as the primary winding 103 and the first secondary winding in Figure 1
  • the end marked with a dot in the winding 101 and the second secondary winding 102 is the end with the same name.
  • the end that is not the end with the same name is the end with the same name, that is, the end that is not marked with a dot among the primary winding 103 , the first secondary winding 101 and the second secondary winding 102 in FIG. 1 .
  • the relationship between the positive half cycle, the negative half cycle and the terminal of the same name is illustrated: when the primary side winding 103 inputs the positive half cycle of the voltage (P1 is positive, P2 is negative), that is, the first The first end S1 of the secondary winding 101 is greater than the second end S2 of the first secondary winding 101, the first end S3 of the second secondary winding is greater than the second end S4 of the second secondary winding 102; and S2 and S3 are connected in series After grounding, so the voltage of S2 is equal to the voltage of S3 is equal to 0, assuming that the input voltage of the primary winding 103 is 10V, and the turn ratio is 10:1, then the output voltage of the first secondary winding 101 and the second secondary winding 102 is 1V, that is, the voltage of S1 is +1V, the voltage of S2 is equal to the voltage of S3 is equal to 0V, and the voltage of S4 is -1V.
  • the step-up circuit 20 is connected with the winding transformer 10, and the connection method can be referred to as shown in FIG. 1 .
  • the connection method can be referred to as shown in FIG. 1 .
  • the common terminal in the middle of the series connection is grounded, and the two ends of the edges other than the common terminal are respectively connected to the booster circuit 20 . Therefore, the voltage output by the secondary winding of the winding transformer 10 is output from the output terminal Vo after passing through the boost circuit 20 .
  • the boost circuit 20 is used to boost the voltage output by the first secondary winding 101 during the positive half cycle of the voltage provided by the power supply, and boost the voltage output by the second secondary winding 102 during the negative half cycle of the voltage provided by the power supply.
  • the voltage is boosted. That is, the boost circuit can boost and rectify the lower AC voltage output by the first secondary winding 101 and the second secondary winding 102 into a higher DC voltage, and then output it from the output terminal Vo.
  • the method of boosting the voltage of the booster circuit can be to use the charging and discharging circuit to store energy on the voltage output by the first secondary winding 101 and the second secondary winding 102, and then superimpose the voltage discharged by the charging and discharging circuit with the first secondary
  • the output voltage of the winding 101 and the second secondary winding 102 itself is boosted to the output voltage of the output terminal Vo, so as to achieve the purpose of voltage boosting.
  • the embodiment of the present application can reduce the magnetic induction intensity by keeping the number of turns of the primary winding unchanged, reducing the number of turns of the secondary winding, and increasing the turn ratio between the primary winding and the secondary winding, thereby reducing loss and avoiding The output power drops.
  • one way to reduce the number of turns of the secondary winding is to set the first secondary winding 101 and the second secondary winding 102 as a plurality of parallel windings, then optionally, the first secondary winding 101 and The second secondary winding 102 can include a plurality of parallel secondary windings, as shown in Figure 2, a plurality of parallel secondary windings constitute the first secondary winding 101, and a plurality of parallel secondary windings constitute the second secondary winding
  • the embodiment of the present application can also adopt the design method of planar transformer in the power supply product, reduce the volume of the transformer as a whole from the manufacturing process, further greatly reduce the volume of the power supply, and realize high-quality output of the power supply.
  • the booster circuit 20 boosts the voltage output by the first secondary winding 101 during the positive half cycle of the voltage provided by the power supply, and boosts the voltage output by the second secondary winding 101 during the negative half cycle of the voltage provided by the power supply.
  • the voltage output by 102 is boosted, so that every half cycle, the voltage output by the secondary winding of the winding transformer 10 can be boosted by the boost circuit 20, that is, the output voltage from the output terminal Vo is discharged to the load every half cycle
  • the secondary winding has the secondary winding to supply power to the load in every half cycle, so that the effect of full-wave rectification output can be achieved, and the ripple of the output voltage of the power supply can be greatly reduced.
  • the turn ratio of the primary winding 103 and the secondary winding increases (the number of turns of the primary winding remains unchanged, and the number of turns of the secondary winding decreases, and, in the embodiment of the present application, the subsequent direct use of the turn ratio
  • the term refers to the turn ratio of the primary winding and the secondary winding), so the output voltage of the secondary winding will become smaller, so for the positive half cycle of the input voltage of the primary winding 103, corresponding to from the first secondary winding
  • the voltage output from the first terminal S1 and the second terminal S2 of the winding 101 itself will be smaller than the output voltage corresponding to the original turn ratio, and the booster circuit 20 will output from the first terminal S1 and the second terminal S1 of the first secondary winding 101
  • the voltage output by S2 is superimposed on the voltage output from the first secondary winding 101 by using the charging and discharging circuit to store and release energy, so as to boost the voltage output by the first secondary winding 101, so that the voltage provided by the power supply is In the positive half cycle,
  • the booster circuit 20 performs the voltage output from the first terminal S3 and the second terminal S4 of the second secondary winding 102 and the voltage output from the second secondary winding 102 by using the charging and discharging circuit.
  • the voltage superposition of energy storage and release can boost the voltage output by the second secondary winding 102 , so that the output voltage from the output terminal Vo remains unchanged during the negative half cycle of the voltage provided by the power supply.
  • the booster circuit boosts the voltage output by the first secondary winding during the positive half cycle of the voltage provided by the power supply, and outputs the second secondary winding during the negative half cycle of the voltage provided by the power supply
  • the voltage is boosted, so that no matter in the positive half cycle or the negative half cycle, the secondary winding outputs the same voltage to the output terminal to discharge the load, achieving the effect of full-wave rectified output, which greatly reduces the ripple of the output voltage of the power supply. Wave.
  • the resonant converter provided by the embodiment of the present application includes a winding transformer and a boost circuit;
  • the winding transformer includes at least one set of secondary windings, and each set of secondary windings includes a first secondary winding and a second secondary winding ;
  • the first secondary winding is connected in series with the second secondary winding;
  • the winding transformer transforms the voltage provided by the power supply, and outputs the transformed voltage from the first secondary winding in the positive half cycle of the voltage provided by the power supply, and the The transformed voltage is output from the second secondary winding in the negative half cycle of the voltage provided by the power supply, and the boost circuit boosts the voltage output by the first secondary winding in the positive half cycle of the voltage provided by the power supply.
  • the negative half cycle of the voltage of the second secondary winding boosts the voltage output by the second secondary winding.
  • the turn ratio avoids the reduction of output power, so as to reduce the volume of the winding transformer by reducing the cross-sectional area of the magnetic core, thereby reducing the volume of the power supply product; at the same time, each half cycle, the first secondary winding of the winding transformer and the second
  • the voltage output by the secondary winding can be boosted by the booster circuit, so that the first secondary winding and the second secondary winding have a secondary winding to supply power to the load in every half cycle, so that the full-wave rectified output can be achieved
  • the effect greatly reducing the ripple of the output voltage of the power supply. Therefore, the resonant converter provided by the embodiment of the present application can reduce the
  • each set of secondary windings corresponds to a boost circuit
  • the boost circuit 20 Including a first charging and discharging circuit 201 and a second charging and discharging circuit 202; the first end of the first charging and discharging circuit 201 is connected to the first end of the first secondary winding 101, and the first end of the second charging and discharging circuit 202 is connected to the first The second ends of the two secondary windings 102 are connected, and the second ends of the first charge and discharge circuit 201 and the second end of the second charge and discharge circuit 202 are both grounded; the first charge and discharge circuit 201 is used for the voltage provided by the power supply.
  • the second charge and discharge circuit 202 is used to charge in the positive half cycle of the voltage provided by the power supply and discharge in the negative half cycle
  • the periodic discharge is used to boost the voltage output by the second secondary winding 102 .
  • the boost circuit 20 includes a first charging and discharging circuit 201 and a second charging and discharging circuit 202 .
  • the charging and discharging circuit can realize charging and discharging, that is, the first charging and discharging circuit 201 charges according to the voltage output by the first secondary winding in the negative half cycle of the voltage provided by the power supply, and discharges in the positive half cycle; the second charging and discharging circuit 202 In the positive half cycle of the voltage provided by the power supply, it is charged according to the voltage output by the second secondary winding, and discharged in the negative half cycle.
  • the first charging and discharging circuit 201 since the first charging and discharging circuit 201 has completed charging in the last negative half cycle using the voltage output by the first secondary winding 101, then the first charging and discharging circuit will be charged in the current positive half cycle. Discharge to the output terminal Vo to discharge the load. At this time, the voltage output by the first secondary winding 101 itself in the current positive half cycle will also be output to the output terminal Vo to discharge the load. In this way, for the output terminal Vo, the voltage output by the first secondary winding 101 itself and the voltage discharged by the first charge-discharge circuit are superimposed together to discharge the load, that is, the output voltage of the first secondary winding 101 itself is relatively high. The small voltage is superimposed on the voltage discharged by the first charging and discharging circuit to boost the voltage output by the first secondary winding 101 .
  • the output voltage of the first secondary winding 101 is 2V1, and the fixed voltage required by the output terminal Vo is also 2V1.
  • the voltage is reduced to V1.
  • the output voltage discharged by the first charging and discharging circuit 201 is also V1, so in the current positive half cycle, the output voltage V1 of the first secondary winding 101 and After the voltage V1 discharged by the first charging and discharging circuit 201 is superimposed, the voltage output from the output terminal Vo is still 2V1.
  • the second charging and discharging circuit 202 since the second charging and discharging circuit 202 has completed charging with the voltage output by the second secondary winding 102 in the previous positive half cycle, the second charging and discharging circuit will be charged in the current negative half cycle. Periodically discharge to the output terminal Vo to discharge the load. At this time, the voltage output by the second secondary winding 102 itself in the current negative half cycle will also be output to the output terminal Vo to discharge the load. In this way, for the output terminal Vo, the voltage output by the second secondary winding 102 itself and the voltage discharged by the second charging and discharging circuit are superimposed together to discharge the load, that is, the output voltage of the second secondary winding 102 itself is relatively high. The small voltage is superimposed on the voltage discharged by the second charging and discharging circuit to boost the voltage output by the second secondary winding 102 .
  • the output voltage of the second secondary winding 102 is 2V1, and the fixed voltage required by the output terminal Vo is also 2V1.
  • the voltage is reduced to V1.
  • the output voltage discharged by the second charge-discharge circuit 202 is also V1, so in the current negative half cycle, the output voltage V1 of the second secondary winding 102 and After the voltage V1 discharged by the second charging and discharging circuit 202 is superimposed, the voltage output from the output terminal Vo is still 2V1.
  • the first end of the first charging and discharging circuit is connected to the first end of the first secondary winding, and the second charging and discharging circuit
  • the first end of the circuit is connected to the second end of the second secondary winding, the second end of the first charging and discharging circuit and the second end of the second charging and discharging circuit are both grounded; Charge in the negative half cycle and discharge in the positive half cycle to boost the output voltage of the first secondary winding; the second charging and discharging circuit charges in the positive half cycle of the voltage provided by the power supply and discharges in the negative half cycle to boost the output voltage of the first secondary winding.
  • the voltage output by the secondary winding is boosted.
  • the two charging and discharging circuits charge the voltages output by different secondary windings respectively, and then superimpose discharge in the discharge period of the corresponding secondary windings to realize the boost of the output voltage of the corresponding secondary windings.
  • the idea is to compensate for the defect that the output voltage of the secondary winding is smaller after increasing the turn ratio (the number of turns of the primary winding remains unchanged and the number of turns of the secondary winding is reduced).
  • no matter in the positive half cycle or in the negative half cycle there is a secondary winding and the corresponding charge and discharge voltage to discharge the load, so that the amplitude of the output voltage changes little, thereby reducing the ripple of the output voltage.
  • the specific circuit implementation manner of the charging and discharging circuit may be in any manner, which is not limited in the embodiment of the present application, as long as the purpose of boosting and rectifying can be achieved.
  • the following embodiments of the present application respectively provide a specific circuit implementation for the first charging and discharging circuit and the second charging and discharging circuit.
  • the first charging and discharging circuit 201 includes a first capacitor C1, a first rectifier tube 2011 and a second rectifier tube 2012; the first end of the first capacitor C1 is connected to the first secondary winding 101 is connected to the first end, the second end of the first capacitor C1 is connected to the first end of the first rectifier tube 2011, and the second end of the first rectifier tube 2011 is the output terminal Vo of the resonant converter; the second rectifier tube 2012 The first end of the second rectifier tube 2012 is connected to the second end of the first capacitor C1 and the first end of the first rectifier tube 2011; wherein, the first capacitor C1 is at the negative half of the voltage provided by the power supply The cycle is charged according to the voltage output by the first secondary winding, and discharged in the positive half cycle of the voltage provided by the power supply; the first rectifier tube 2011 is turned on in the positive half cycle of the voltage provided by the power supply, and in the negative half cycle of the voltage provided by the power supply Turn off; the second rectifier tube
  • the first charging and discharging circuit 201 includes the first capacitor C1 , the first rectifying tube 2011 and the second rectifying tube 2012 as an example for illustration.
  • the first capacitor C1 can be charged and discharged, and the first rectifier tube 2011 and the second rectifier tube 2012 perform rectification and filtering, that is, the alternating current is rectified and filtered into direct current.
  • the first charging and discharging circuit 201 is charged with the voltage output by the first secondary winding 101 in the negative half cycle, and is superimposed and discharged on the basis of the voltage output by the first secondary winding 101 in the positive half cycle. Therefore, the first capacitor in the first charging and discharging circuit 201 is charged with the voltage output from the first secondary winding 101 in the negative half cycle, and superimposed and discharged on the basis of the voltage output from the first secondary winding 101 in the positive half cycle.
  • the first rectifier tube 2011 is turned on in the positive half cycle and turned off in the negative half cycle; the second rectifier tube 2012 is opposite to the first rectifier tube 2011, turned off in the positive half cycle and turned on in the negative half cycle.
  • the first capacitor of the first charging and discharging circuit 201 may be a bootstrap capacitor, or may be multiple first capacitors connected in parallel, and the number of first capacitors is not limited in this embodiment of the present application. By connecting the first capacitors in parallel, the capacitance value of each first capacitor can be reduced, reducing the difficulty of implementing the first charging and discharging circuit.
  • both the first rectifier tube 2011 and the second rectifier tube 2012 are diodes; or, both the first rectifier tube 2011 and the second rectifier tube 2012 are switch tubes.
  • the switch tube can be a MOS tube or GaN, SiC and other devices, thereby reducing the switching loss and conduction loss of the switch tube, and improving reliability.
  • the embodiment of the present application does not limit the specifics of the switch tube, as long as the rectification switching function is realized That's it.
  • FIG. 5 is a schematic diagram showing that both the first rectifier tube 2011 and the second rectifier tube 2012 are diodes.
  • the first end of the first rectifier tube 2011 and the first end of the second rectifier tube 2012 refer to Both are the anode of the diode; the second end of the first rectifier tube 2011 and the second end of the second rectifier tube 2012 both refer to the cathode of the diode.
  • the first rectifier 2011 is a diode D1
  • the second rectifier 2012 is a diode D2
  • the anode of the diode D1 is connected to the second end of the first capacitor C1
  • the cathode of the diode D1 is connected to the output terminal Vo of the resonant converter
  • the diode D2 The anode of the diode D2 is connected to the ground, and the cathode of the diode D2 is connected to the second terminal of the first capacitor C1 and the anode of the diode D1.
  • the diode D1 is turned on in the positive half cycle of the voltage provided by the power supply, and turned off in the negative half cycle of the voltage provided by the power supply; the diode D2 is turned off in the positive half cycle of the voltage provided by the power supply, and in the negative half cycle of the voltage provided by the power supply conduction.
  • FIG. 6 is a schematic diagram of both the first rectifier tube 2011 and the second rectifier tube 2012 being MOS tubes.
  • the first end of the first rectifier tube 2011 and the first end of the second rectifier tube 2012 Both refer to the source of the switch tube; the second end of the first rectifier tube 2011 and the second end of the second rectifier tube 2012 both refer to the drain of the switch tube.
  • the first rectifier 2011 is a MOS transistor Q1
  • the second rectifier 2012 is a MOS transistor Q2
  • the source of the MOS transistor Q1 is connected to the second end of the first capacitor C1
  • the drain of the MOS transistor Q1 is connected to the resonant converter.
  • the output terminal Vo; the source of the MOS transistor Q2 is grounded, and the drain of the MOS transistor Q2 is connected to the second end of the first capacitor C1 and the source of the MOS transistor Q1.
  • the MOS transistor Q1 is turned on in the positive half cycle of the voltage provided by the power supply, and turned off in the negative half cycle of the voltage provided by the power supply; the MOS transistor Q2 is turned off in the positive half cycle of the voltage provided by the power supply, and in the negative half cycle of the voltage provided by the power supply. half-cycle conduction.
  • the gate (control electrode) of the MOS transistor is connected to drive a controller (not shown in the figure), such as a half-bridge driver, which receives a control signal through the control electrode, and is turned on or off according to the control signal.
  • a controller such as a half-bridge driver, which receives a control signal through the control electrode, and is turned on or off according to the control signal.
  • the control of the MOS transistor in the embodiment of the present application is not limited.
  • the first charging and discharging circuit is realized by the first capacitor, the first rectifier tube and the second rectifier tube.
  • the discharge is superimposed on the voltage output by the first secondary winding to boost the voltage output by the first secondary winding, thereby ensuring the stability of the output voltage at the output end.
  • the voltage output by the first secondary winding and the voltage discharged by the first capacitor will be rectified and filtered by the rectifier tube, and the alternating current will be converted into direct current, so as to discharge the load by outputting the voltage from the output terminal.
  • the second charging and discharging circuit 202 includes a second capacitor C2, a third rectifier tube 2013, and a fourth rectifier tube 2014; the first end of the second capacitor C2 and the second secondary winding 102, the second end of the second capacitor C2 is connected to the first end of the fourth rectifier tube 2014, and the second end of the fourth rectifier tube 2014 is connected to the output terminal Vo of the resonant converter; the third rectifier tube 2013 The first end of the rectifier tube 2013 is grounded, the second end of the third rectifier tube 2013 is connected to the second end of the second capacitor C2 and the first end of the fourth rectifier tube 2014; wherein, the second capacitor C2 is at the positive half of the voltage provided by the power supply The cycle is charged according to the voltage output by the second secondary winding, and discharged in the negative half cycle of the voltage provided by the power supply; the third rectifier tube 2013 is turned on in the positive half cycle of the voltage provided by the power supply, and in the negative half cycle of the voltage provided by the power supply Turn off
  • the second charge and discharge circuit 202 includes the second capacitor C2 , the third rectifier 2013 and the fourth rectifier 2014 as an example for illustration.
  • the second capacitor C2 can be charged and discharged, and the third rectifier 2013 and the fourth rectifier 2014 perform rectification and filtering, that is, the alternating current is rectified and filtered into direct current.
  • the second charging and discharging circuit 202 is charged with the voltage output by the second secondary winding 102 in the positive half cycle, and is superimposed and discharged on the basis of the voltage output by the second secondary winding 102 in the negative half cycle. Therefore, the second capacitor C2 in the second charging and discharging circuit 202 is also charged by the voltage output from the second secondary winding 102 in the positive half cycle, and superimposed and discharged on the basis of the voltage output by the second secondary winding 102 in the negative half cycle.
  • the third rectifier tube 2013 is turned on in the positive half cycle, and turned off in the negative half cycle; the fourth rectifier tube 2014 is opposite to the third rectifier tube 2013, turned off in the positive half cycle, and turned on in the negative half cycle.
  • the second capacitor in practical application, it may also be a bootstrap capacitor, or multiple first capacitors connected in parallel, and the number of the second capacitors in this embodiment of the present application is not limited. By connecting the second capacitors in parallel, the capacitance value of each second capacitor can be reduced, reducing the difficulty of implementing the second charging and discharging circuit.
  • both the third rectifier tube 2013 and the fourth rectifier tube 2014 are diodes; or, both the third rectifier tube 2013 and the fourth rectifier tube 2014 are switch tubes.
  • the switch tube can be a MOS tube or GaN, SiC and other devices, thereby reducing the switching loss and conduction loss of the switch tube, and improving reliability.
  • the embodiment of the present application does not limit the specifics of the switch tube, as long as the rectification switching function is realized That's it.
  • FIG. 8 is a schematic diagram showing that the third rectifier 2013 and the fourth rectifier 2014 are diodes.
  • the first end of the third rectifier 2013 and the first end of the fourth rectifier 2014 refer to Both are the anode of the diode; the second end of the third rectifier 2013 and the second end of the fourth rectifier 2014 both refer to the cathode of the diode.
  • the third rectifier 2013 is a diode D3 and the fourth rectifier 2014 is a diode D4, the anode of the diode D3 is grounded, the cathode of the diode D3 is connected to the second end of the second capacitor C2 and the anode of the diode D4; the anode of the diode D4 is connected to The second terminal of the second capacitor C2 and the cathode of the diode D4 are connected to the output terminal Vo of the resonant converter.
  • the diode D3 is turned on in the positive half cycle of the voltage provided by the power supply, and turned off in the negative half cycle of the voltage provided by the power supply; the diode D4 is turned off in the positive half cycle of the voltage provided by the power supply, and in the negative half cycle of the voltage provided by the power supply conduction.
  • FIG. 9 is a schematic diagram of both the third rectifying tube 2013 and the fourth rectifying tube 2014 being MOS tubes.
  • the first end of the third rectifying tube 2013 and the first end of the fourth rectifying tube 2014 Both refer to the source of the switch tube; the second end of the third rectifier 2013 and the second end of the fourth rectifier 2014 both refer to the drain of the switch.
  • the third rectifier 2013 is a MOS transistor Q3, and the fourth rectifier 2014 is a MOS transistor Q4, the source of the MOS transistor Q3 is grounded, the drain of the MOS transistor Q3 is connected to the second end of the second capacitor C2, and the MOS transistor Q4 The source of the MOS transistor Q4 is connected to the output terminal Vo of the resonant converter; and the MOS transistor Q3 is turned on in the positive half cycle of the voltage provided by the power supply, and turned off in the negative half cycle of the voltage provided by the power supply; the MOS transistor Q4 Q4 is turned off during the positive half cycle of the voltage provided by the power supply and turned on during the negative half cycle of the voltage provided by the power supply.
  • the gate (control electrode) of the MOS transistor is also connected to drive a controller (not shown in the figure), such as a half-bridge driver, which receives a control signal through the control electrode, and turns on or off according to the control signal. broken.
  • a controller such as a half-bridge driver, which receives a control signal through the control electrode, and turns on or off according to the control signal. broken.
  • the control of the MOS transistor in the embodiment of the present application is not limited.
  • the diodes and switches in the first charging and discharging circuit 201 and the second charging and discharging circuit 202 can be turned on or off according to the output voltage of the secondary winding. Moreover, by adopting the structure of the embodiment of the present application, the reverse withstand voltage of the rectifier can be reduced, thereby improving the reliability of the circuit.
  • the second charging and discharging circuit is implemented with the second capacitor, the third rectifier and the fourth rectifier.
  • the second capacitor is charged with the voltage output by the second secondary winding in the positive half cycle, and in the The discharge is superimposed on the voltage output by the second secondary winding, so as to boost the voltage output by the second secondary winding and ensure the stability of the output voltage at the output terminal.
  • the voltage output by the second secondary winding and the voltage discharged by the second capacitor will be rectified and filtered by the rectifier tube to convert the alternating current into direct current, so as to output the voltage from the output end to discharge the load.
  • the embodiment of the present application In the resonant converter provided, there are secondary windings and charging and discharging circuits to superimpose and discharge the load, which makes up for the phenomenon that the output voltage of the output terminal will decrease due to the decrease of the output voltage of the secondary side after the increase of the turn ratio, and ensures the positive and negative During the two half-cycle switching process, the stability of the output voltage at the output terminal reduces the ripple of the output power supply; and increasing the turn ratio is to reduce the area of the magnetic core and thereby reduce the volume of the transformer, thereby reducing the volume of the power supply product , so the volume of the power supply product is reduced while reducing the output ripple of the power supply.
  • the boost circuit may further include a third charging and discharging circuit 203; the first terminal of the third charging and discharging circuit 203 is connected to the second terminal of the first charging and discharging circuit 201 , the second end of the third charging and discharging circuit 203 is grounded; the third charging and discharging circuit 203 is used to superimpose the output voltage of the first secondary winding with the first charging and discharging circuit in the positive half cycle, and in the negative half cycle of the power supply superimposed with the second charge and discharge circuit to boost the output voltage of the second secondary winding.
  • the third charging and discharging circuit 203 includes a third capacitor C3; the first terminal of the third capacitor C3 is connected to the second terminal of the first charging and discharging circuit 201, and the second terminal of the third capacitor C3 is grounded.
  • the third capacitor C3 in the third charging and discharging circuit 203 may be a plurality of capacitors connected in parallel, or may be one capacitor, which is not limited in this embodiment of the present application.
  • a resonant conversion is combined with the third capacitor C3 (third charge and discharge circuit 203) and the internal circuit structure of the first charge and discharge circuit 201 and the second charge and discharge circuit 202. device.
  • the first end of the third charging and discharging circuit 203 is connected to the second end of the first charging and discharging circuit 201, and the second end of the first charging and discharging circuit 201 is the position where the voltage is output to the output terminal Vo after boosting and rectifying.
  • the second end of the discharge circuit is grounded.
  • the first secondary winding 101 and the first charging and discharging circuit 201 when they superimpose the output voltage to the output terminal, they will charge the third charging and discharging circuit at the same time, and because of the superimposed charging, the third charging and discharging circuit will be Fast charging, after charging to the peak value, on the basis of the first secondary winding 101 and the first charging and discharging circuit 201, the output voltage of the output terminal Vo is further superimposed to discharge the load.
  • the second terminal of the first charging and discharging circuit 201 is the position where the voltage is output to the output terminal Vo after boosting and rectifying, which is substantially the same as the position of the output voltage of the second terminal of the second charging and discharging circuit 202. belong to the same point, so the first end of the third charging and discharging circuit 203 is equivalent to being connected to the second end of the second charging and discharging circuit 202, therefore, in the negative half cycle, the second secondary winding 102 and the second charging and discharging circuit
  • the second secondary winding 102 and the second charging and discharging circuit When 202 superimposes the output voltage to the output terminal, it will also charge the third charging and discharging circuit at the same time, and because of the superimposed charging, the third charging and discharging circuit will be fully charged quickly.
  • the output voltage to the output terminal Vo will be further superimposed to discharge the load.
  • the third charging and discharging circuit 203 uses the output voltage of the first secondary winding 101 to charge in the positive half cycle, and then continues to discharge in the positive half cycle after being fully charged, so as to achieve superimposition with the first charging and discharging circuit on the first secondary winding.
  • the output voltage of 101 is boosted.
  • the output voltage of the second secondary winding 102 is used to charge and then continue to discharge in the negative half cycle after being fully charged, so as to realize the superimposition of the second secondary winding 102 with the second charge and discharge circuit.
  • the output voltage of the winding 102 is boosted. Therefore, the stability of the output voltage at the output terminal can be further ensured by setting the third charging and discharging circuit 203 .
  • the winding transformer includes multiple sets of secondary windings, where the multiple sets of secondary windings are arranged in parallel, and each set of secondary windings corresponds to a booster circuit , that is, multiple parallel boost circuits need to be set.
  • FIG. 13 on the basis of the above-mentioned FIG. 12 , that is, still taking the MOS tube as an example, a schematic diagram of a resonant converter in which booster circuits are connected in parallel is provided.
  • FIG. 13 there are m sets of secondary windings (boosting circuit 1 to boosting circuit m), and each set of secondary windings corresponds to a boosting circuit 20 .
  • the booster circuit 1 to the booster circuit m are all the same circuit, so the output voltages from V1O to VnO are the same, so as to achieve the effect of supplying power to different loads.
  • the first secondary winding 101 and the second secondary winding 102 can be composed of multiple secondary windings connected in parallel, so in the scenario where multiple loads are applied, in one embodiment, the Parallel connection of secondary windings (referring to the parallel connection of n first secondary windings for the first secondary winding, and parallel connection of n second secondary windings for the second secondary winding) and parallel connection of multiple sets of secondary windings (corresponding to multiple booster circuits parallel connection), as shown in FIG. 14 , which is a schematic diagram of a resonant converter provided after combining the above-mentioned FIG. 13 and the above-mentioned FIG. 2 .
  • the parallel connection of the secondary windings further increases the primary and secondary turns ratio of the transformer, so increasing the turns ratio of the primary windings and secondary windings can further reduce the magnetic induction intensity, thereby reducing losses and avoiding a drop in output power.
  • the cross-sectional area of the magnetic core can be reduced, thereby reducing the volume of the winding transformer, so that it can also be used in multiple load scenarios to reduce the volume of the winding transformer to reduce the volume of the power supply product.
  • m in Fig. 13 and Fig. 14 and n in Fig. 2 both represent multiple, and all are positive integers, and are represented by n and m in order to distinguish them. In practical applications, n can be equal to m , may also be unequal, which is not limited in this embodiment of the present application.
  • the embodiment of the present application also provides a power supply, which includes any resonant converter provided in the previous embodiments.
  • a terminal is also provided, and the terminal includes any power source formed by a resonant converter.
  • the resonant converter in the above embodiments includes a winding transformer and a booster circuit; the output voltage of the secondary winding is boosted during both the positive half cycle and the negative half cycle of the voltage provided by the power supply.
  • the step-up circuit By setting the step-up circuit to boost the output voltage of the secondary winding of the transformer, it can ensure that the output voltage to the load remains unchanged, and then increase the turn ratio between the primary winding and the secondary winding of the winding transformer to avoid the reduction of the output power.
  • By reducing the cross-sectional area of the magnetic core the volume of the winding transformer is reduced, thereby reducing the volume of the power supply product.
  • a terminal refers to any electronic device that requires an external power supply or a built-in power supply, such as various personal computers, notebook computers, mobile phones (smart mobile terminals), tablet computers, and portable wearable devices, which are not limited in this embodiment. If it is an external power supply, the power supply may be a power adapter, a mobile power supply (power bank, travel charger), etc., which is not limited in this embodiment.
  • terminals in addition to terminals, it can also be devices that require power, for example, electric vehicles, drones, e-books, electronic cigarettes, smart electronic devices (including watches, bracelets, smart glasses, sweeping robots, etc.), small electronic products ( Including wireless earphones, Bluetooth speakers, electric toothbrushes, rechargeable wireless mice, etc.), or (5G) communication module power supplies, etc., which are not limited in this embodiment of the application.
  • electric vehicles for example, electric vehicles, drones, e-books, electronic cigarettes, smart electronic devices (including watches, bracelets, smart glasses, sweeping robots, etc.), small electronic products ( Including wireless earphones, Bluetooth speakers, electric toothbrushes, rechargeable wireless mice, etc.), or (5G) communication module power supplies, etc., which are not limited in this embodiment of the application.
  • the resonant converter provided in the example can not only reduce the volume of the transformer but also the volume of the output capacitor, reduce the output ripple of the power supply, improve the quality of the output voltage of the power supply, and further increase the power density of the power supply.
  • the resonant converter provided by the embodiment of this application is not only suitable for LLC architecture power supply, but also for the phase-shifted full bridge, DCX and many other circuits that require full-wave rectification on the secondary side, and the output through voltage doubler rectification, the embodiment of this application can be used Reduce the volume of the transformer and output capacitor while reducing the output ripple of the power supply.
  • the embodiment of the present application also provides an embodiment of a voltage boosting method, as shown in FIG. 15 , this embodiment involves running a computer program to realize the output of the resonant converter to the load.
  • the specific process of voltage boosting includes:
  • a program instruction for instructing voltage transformation can be preset, and after the computer device receives the trigger of the program instruction, it will perform the corresponding operation, that is, according to the preset configuration, the voltage provided by the power supply will be transformed and output The transformed voltage, and the output transformed voltage includes the first voltage and the second voltage.
  • the voltage values of the first voltage and the second voltage may be equal or different, which is not limited in this embodiment of the present application.
  • the difference between the first voltage and the second voltage may be output from different ports, for example, taking a transformer including two series-connected secondary windings as an example, the first voltage may be output from the first secondary winding of the transformer, The second voltage can be output from the second secondary winding of the transformer; conversely, the first voltage can be output from the second secondary winding of the transformer, and the second voltage can be output from the first secondary winding of the transformer.
  • the example is not limited to this.
  • the computer device can continue to execute preset program instructions to boost the voltage, and when boosting, the first voltage is boosted in the positive half cycle of the voltage provided by the power supply , boosting the second voltage during the negative half cycle of the voltage provided by the power supply.
  • the voltage output by the first secondary winding is boosted in the positive half cycle
  • the voltage output by the second secondary winding is boosted in the negative half cycle.
  • it can be realized by setting a charging and discharging device. Specifically, it can be used to charge the charging and discharging device with the first voltage during the positive half cycle of the voltage provided by the power supply.
  • the charging and discharging device In the negative half cycle of the voltage provided by the power supply, the charging and discharging device is also charged with the second voltage, so that in the positive half cycle, the charging and discharging device will also discharge, and the discharged voltage will also be output to the load to supply power to the load, while at the same time The positive half-cycle first voltage itself will also be output to the load to supply power to the load.
  • both the positive half cycle and the negative half cycle of the voltage provided by the power supply boost the transformed voltage, so that for the load, the magnitude of the accepted voltage change is relatively small in the positive half cycle and the negative half cycle. Small, the overall is more stable, thereby reducing the ripple effect of the output voltage to the load.
  • these computer program instructions can also be loaded on a computer or other programmable data processing device, so that a series of operation steps are performed on the computer or other programmable device to produce a computer-implemented process, so that on the computer or other programmable device Executing the computer program instructions realizes the above functions.

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Abstract

本申请涉及一种谐振变换器、电源、终端和升压方法，该谐振变换器包括绕组变压器、升压电路；绕组变压器包括至少一组副边绕组，每组副边绕组包括第一副边绕组和第二副边绕组；第一副边绕组与第二副边绕组串联；绕组变压器将电源提供的电压进行变压，并将变压后的电压在电源提供的电压的正半周期从第一副边绕组输出，将变压后的电压在电源提供的电压的负半周期从第二副边绕组输出；升压电路在电源提供的电压的正半周期对第一副边绕组输出的电压进行升压，在电源提供的电压的负半周期对第二副边绕组输出的电压进行升压。该谐振变换器可以通过减小磁芯横截面积减小绕组变压器的体积，从而减小电源产品体积；同时可以大大降低电源输出电压的纹波。

Description

谐振变换器、电源、终端和升压方法 技术领域
本申请涉及充电技术领域，特别是涉及一种谐振变换器、电源、终端和升压方法。
背景技术
随着开关电源的发展，软开关技术得到了广泛的发展和应用。例如，LLC谐振变换器由于其采用软开关技术，具有开关损耗小、适用于高功率密度设计等优点，已经被广泛应用于中大功率电源产品上。
LLC谐振变换器的主要器件包括绕组变压器和储能电感，而绕组变压器的体积较大，导致采用LLC谐振变换器的电源产品整个系统电路体积都偏大。目前，减小体积提高功率密度是电源发展的趋势，基于此，可以通过减小绕组变压器的体积来减小电源产品体积，在采用倍压电路的方式来保证变压器体积的减小同时避免输出功率降低。
发明内容
基于此，有必要针对上述技术问题，提供一种谐振变换器、电源、终端和升压方法，能够减小电源产品的体积的同时减小电源输出纹波。
第一方面，本申请实施例提供一种谐振变换器，该谐振变换器包括绕组变压器、升压电路；绕组变压器包括至少一组副边绕组，每组副边绕组包括第一副边绕组和第二副边绕组；第一副边绕组与第二副边绕组串联；
绕组变压器，用于将电源提供的电压进行变压，并将变压后的电压在电源提供的电压的正半周期从第一副边绕组输出，将变压后的电压在电源提供的电压的负半周期从第二副边绕组输出；
升压电路，用于在电源提供的电压的正半周期对第一副边绕组输出的电压进行升压，在电源提供的电压的负半周期对第二副边绕组输出的电压进行升压。
在其中一个实施例中，上述每组副边绕组对应的升压电路包括第一充放电电路和第二充放电电路；第一充放电电路的第一端与第一副边绕组的第一端连接，第二充放电电路的第一端与第二副边绕组的第二端连接，第一充放电电路的第二端和第二充放电电路的第二端均接地；
第一充放电电路，用于在电源提供的电压的负半周期根据第一副边绕组输出的电压进行充电，并在正半周期放电以对第一副边绕组输出的电压进行升压；
第二充放电电路，用于在电源提供的电压的正半周期根据第二副边绕组输出的电压进行充电，并在负半周期放电以对第二副边绕组输出的电压进行升压。
在其中一个实施例中，上述第一充放电电路包括第一电容、第一整流管和第二整流管；第一电容的第一端与第一副边绕组的第一端连接，第一电容的第二端与第一整流管的第一端连接，第一整流管的第二端为谐振变换器的输出端；第二整流管的第一端接地，第二整流管的第二端与第一电容的第二端、第一整流管的第一端连接；其中，
第一电容在电源提供的电压的负半周期根据第一副边绕组输出的电压进行充电，在电源提供的电压的正半周期放电；
第一整流管在电源提供的电压的正半周期导通，在电源提供的电压的负半周期关断；
第二整流管在电源提供的电压的正半周期关断，在电源提供的电压的负半周期导通。
在其中一个实施例中，上述第一整流管和第二整流管均为二极管；或者，第一整流管和第二整流管均为开关管。
在其中一个实施例中，上述第二充放电电路包括第二电容、第三整流管和第四整流管；第二电容的第一端与第二副边绕组的第二端连接，第二电容的第二端与第四整流管的第一端连接，第四整流管的第二端连接谐振变换器的输出端；第三整流管的第一端接地，第三整流管的第二端与第二电容的第二端、第四整流管的第一端连接；其中，
第二电容在电源提供的电压的正半周期根据第二副边绕组输出的电压进行充电，在电源提供的电压的负半周期放电；
第三整流管在电源提供的电压的正半周期导通，在电源提供的电压的负半周期关断；
第四整流管在电源提供的电压的正半周期关断，在电源提供的电压的负半周期导通。
在其中一个实施例中，上述第三整流管和第四整流管均为二极管；或者，第三整流管和第四整流管均为开关管。
在其中一个实施例中，上述升压电路还包括第三充放电电路；第三充放电电路的第一端连接在第一充放电电路的第二端，第三充放电电路的第二端接地；
第三充放电电路，用于在正半周期与第一充放电电路叠加对第一副边绕组的输出电压进行升压，并在电源的负半周期与第二充放电电路叠加对第二副边绕组的输出电压进行升压。
在其中一个实施例中，上述第三充放电电路包括第三电容；第三电容的第一端连接在第一充放电电路的第二端，第三电容的第二端接地。
第二方面，本申请实施例提供一种电源，该电源包括上述第一方面任一实施例中的谐振变换器。
第三方面，本申请实施例提供一种终端，该终端包括上述第二方面任一实施例中的电源。
第四方面，本申请实施例提供一种升压方法，该方法包括：
将电源提供的电压进行变压，在电源提供的电压的正半周期得到第一电压，在所述提供的电压的负半周期得到第二电压；
在电源提供的电压的正半周期对第一电压进行升压，在电源提供的电压的负半周期对第二电压进行升压。
本申请实施例提供的一种谐振变换器、电源、终端和升压方法，该谐振变换器包括绕组变压器、升压电路；绕组变压器包括至少一组副边绕组，每组副边绕组包括第一副边绕组和第二副边绕组；第一副边绕组与第二副边绕组串联；绕组变压器将电源提供的电压进行变压，并将变压后的电压在电源提供的电压的正半周期从第一副边绕组输出，将变压后的电压在电源提供的电压的负半周期从第二副边绕组输出，升压电路在电源提供的电压的正半周期对第一副边绕组输出的电压进行升压，在电源提供的电压的负半周期对第二副边绕组输出的电压进行升压。通过设置升压电路对变压器第一副边绕组和第二副边绕组输出的电压升压，保证输出端输出到负载电压幅值不变，然后增大绕组变压器原边绕组与副边绕组之间的匝比避免了输出功率降低，以通过减小磁芯横截面积减小绕组变压器的体积，从而减小电源产品体积；同时，每个半周期，绕组变压器的第一副边绕组和第二副边绕组输出的电压都可以经过升压电路升压，实现第一副边绕组和第二副边绕组在每半个周期内均有副边绕组对负载供电，从而可以达到全波整流输出的效果，大大降低电源输出电压的纹波。因此，本申请实施例提供的谐振变换器能够减小电源产品的体积的同时减小电源输出纹波。
附图说明
图1为一个实施例中谐振变换器的结构示意图；
图2为另一个实施例中谐振变换器的结构示意图；
图3为另一个实施例中谐振变换器的结构示意图；
图4为另一个实施例中谐振变换器的结构示意图；
图5为另一个实施例中谐振变换器的结构示意图；
图6为另一个实施例中谐振变换器的结构示意图；
图7为另一个实施例中谐振变换器的结构示意图；
图8为另一个实施例中谐振变换器的结构示意图；
图9为另一个实施例中谐振变换器的结构示意图；
图10为另一个实施例中谐振变换器的结构示意图；
图11为另一个实施例中谐振变换器的结构示意图；
图12为另一个实施例中谐振变换器的结构示意图；
图13为另一个实施例中谐振变换器的结构示意图；
图14为另一个实施例中谐振变换器的结构示意图；
图15为一个实施例中升压方法流程示意图。
附图标记说明：
10：绕组变压器；                     103：原边绕组；
101：第一副边绕组；                  102：第二副边绕组；
20：升压电路；                       201：第一充放电电路；
202：第二充放电电路；                203：第三充放电电路；
2011：第一整流电路；                 2012：第二整流电路；
2013：第三整流电路；                 2014：第四整流电路。
具体实施方式
为了使本申请的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本申请进行进一步详细说明。应当理解，此处描述的具体实施例仅仅用以解释本申请，并不用于限定本申请。
可以理解，本申请中为部件所编序号本身，例如“第一”、“第二”等，仅用于区分所描述的对象，不具有任何顺序或技术含义。而本申请所说“连接”、“联接”，如无特别说明，均包括直接和间接连接(联接)。在本申请的描述中，需要理解的是，方位词例如“上”、“下”等指示的方位或位置关系为基于附图所示的方位或位置关系，仅是为了便于描述本申请和简化描述，而不是指示或暗示所指的装置或元件必须具有特定的方位、以特定的方位构造和操作，因此不能理解为对本申请的限制。在本申请中，除非另有明确的规定和限定，第一特征在第二特征“上”或“下”可以是第一和第二特征直接接触，或第一和第二特征通过中间媒介间接接触。本申请中，并不以名称的差异来作为区分元件的方式，而是以元件在功能上的差异来作为区分原则。
如图1所示，本申请实施例提供了一种谐振变换器，该谐振变换器包括绕组变压器10、升压电路20；绕组变压器10包括至少一组副边绕组，每组副边绕组包括第一副边绕组101和第二副边绕组102；第一副边绕组101和第二副边绕组102串联；绕组变压器10，用于将电源提供的电压进 行变压，并将变压后的电压在电源提供的电压的正半周期从第一副边绕组101输出，以及将变压后的电压在电源提供的电压的负半周期从第二副边绕组102输出；升压电路20，用于在电源提供的电压的正半周期对第一副边绕组101输出的电压进行升压，在电源提供的电压的负半周期对第二副边绕组102输出的电压进行升压。
本申请实施例中，绕组变压器可以包括多组副边绕组，每组副边绕组包括两个串联的副边绕组，每组副边绕组对应升压电路，多组副边绕组的情况将在后续实施例中结合升压电路中具体的结构进行说明，在此为了简洁清楚地进行说明，图1中示意出一组副边绕组的情况。请继续参见上述图1，图1中的绕组变压器10包括原边绕组103和第一副边绕组101和第二副边绕组102(第一副边绕组101和第二副边绕组102为一组副边绕组)。第一副边绕组101的第一端S1和第二副边绕组102的第二端S4均与升压电路20连接，第一副边绕组101的第二端S2和第二副边绕组102的第一端S3串联且接地。工作时，绕组变压器10用于将电源提供的电压进行变压，电源提供的电压从绕组变压器10的原边绕组103中进入，从绕组变压器10的副边绕组输出。实际应用中，绕组变压器10根据原边绕组的输入电压、以及原边绕组与副边绕组之间的匝比，得到副边绕组的输出电压。例如，原边绕组103与副边绕组之间的匝比为10:1，假设原边绕组103的输入电压为10V，则副边绕组的输出电压为1V。需要强调的是，图1中的原边绕组和副边绕组的线圈匝数只是示意，不用于限定实际的线圈匝数。
对于变压器本身来说，其原边接的是电源即电源侧，副边接的是负载即负荷侧，且变压器的原边输入和副边输出都为交流电，例如，交流电的波形可以是正弦波形、三角形波、方波形等。所以本申请实施例中的绕组变压器10的副边绕组输出的电压分为正半周期和负半周期。变压器的初级线圈(原边)与次级线圈(副边)的缠绕方式的关系，决定了变压器的同名端和异名端。其中，若原边和副边两个绕组方向一致时，两个绕组的起绕点是同名端，但若两个绕组方向相反时，其中一个绕组的起饶点和另一个绕组的结束点是同名端。同名端指的是在同一交变磁通的作用下任一时刻两个(或两个以上)绕组中都具有相同电势极性的端头，例如图1的原边绕组103、第一副边绕组101和第二副边绕组102中标示圆点一端为同名端。同理，不是同名端的一端即为异名端，即图1中的原边绕组103、第一副边绕组101和第二副边绕组102中未标示圆点的一端。
结合图1以方波电压为例，对电压的正半周期、负半周期以及同名端的关系举例说明：在原边绕组103输入电压正半周期(P1为正，P2为负)时，即第一副边绕组101的第一端S1大于第一副边绕组101的第二端S2，第二副边绕组的第一端S3大于第二副边绕组102的第二端S4；且S2和S3串联后接地，所以S2的电压等于S3的电压等于0，假设原边绕组103的输入电压为10V，匝比为10:1，则第一副边绕组101和第二副边绕组102的输出电压为1V，即S1的电压为+1V，S2的电压等于S3的电压等于0V，S4的电压为-1V。
其中，升压电路20是与绕组变压器10连接的，连接方式可参见图1所示。具体地，绕组变压器10的第一副边绕组101和第二副边绕组102串联之后，中间串联的公共端接地，除公共端以外的边缘的两端分别与升压电路20连接。所以绕组变压器10的副边绕组输出的电压是经过升压电路20之后再从输出端Vo输出的。
其中，升压电路20，用于在电源提供的电压的正半周期对第一副边绕组101输出的电压进行升压，在电源提供的电压的负半周期对第二副边绕组102输出的电压进行升压。即升压电路可以将 第一副边绕组101和第二副边绕组102输出的较低的交流电压升压并整流为较高的直流电压，再从输出端Vo输出。例如，升压电路升压的方式可以是利用充放电电路对第一副边绕组101和第二副边绕组102输出的电压进行储能，然后通过叠加充放电电路放电的电压与第一副边绕组101及第二副边绕组102本身输出的电压的方式提升到输出端Vo的输出电压，从而达到升压的目的。
可以理解的是，对于负载来说，其所需的电压是固定的，即输出端Vo的输出电压应是固定的，在此前提下，本申请实施例中副边绕组101输出的电压允许较低一些，这样较低的电压经过升压电路20升压后可达到输出端Vo要求的固定输出电压。基于此思路，本申请实施例可以通过保持原边绕组的匝数不变，减少副边绕组的匝数，提高原边绕组和副边绕组的匝比来减小磁感应强度，从而降低损耗，避免输出功率下降。而增大匝比后就可以减小磁芯横截面积，进而减小绕组变压器的体积，实现了通过减小绕组变压器的体积来减小电源产品体积的目的。可选地，减少副边绕组的匝数的一种方式为将第一副边绕组101和第二副边绕组102设置为多个并联的绕组，则可选地，第一副边绕组101和第二副边绕组102均可以包括多个并联的副边绕组，如图2所示，多个并联的副边绕组构成第一副边绕组101，多个并联的副边绕组构成第二副边绕组102的示意图，为了简洁清楚，图2中仅示意出第1个和第n个，中间的副边绕组均以省略号替代。可以理解的是，并联的所有副边绕组均为相同副边绕组。可选地，本申请实施例还可以在电源产品中采用平面变压器的设计方法，从制作工艺上整体缩小变压器体积，进一步大大减小电源的体积，实现电源高质量的输出。
从对负载输出电压来说，升压电路20在电源提供的电压的正半周期对第一副边绕组101输出的电压进行升压，在电源提供的电压的负半周期对第二副边绕组102输出的电压进行升压，这样每个半周期，绕组变压器10的副边绕组输出的电压都可以经过升压电路20升压，即每个半周期都在从输出端Vo输出电压对负载放电，这样实现副边绕组在每半个周期内均有副边绕组对负载供电，从而可以达到全波整流输出的效果，大大降低电源输出电压的纹波。
具体到图1中，则是原边绕组103和副边绕组的匝比增大(原边绕组匝数不变，副边绕组匝数减小，且，本申请实施例中后续直接采用匝比一词，均指代原边绕组和副边绕组的匝比)，所以副边绕组的输出电压会变小，所以对于原边绕组103输入电压的正半周期时，对应在从第一副边绕组101的第一端S1和第二端S2输出的电压本身会比原来匝比对应输出的电压小，而升压电路20通过将从第一副边绕组101的第一端S1和第二端S2输出的电压和利用充放电电路对第一副边绕组101输出的电压进行储能再释放的电压叠加，实现对第一副边绕组101输出的电压进行升压，使得在电源提供的电压的正半周期，从输出端Vo的输出电压不变。
同理，因为匝比的增加，对于原边绕组103输入电压的负半周期时，对应在从第二副边绕组102的第一端S3和第二端S4输出的电压本身也会比原来匝比对应输出的电压小，而升压电路20通过将从第二副边绕组102的第一端S3和第二端S4输出的电压和利用充放电电路对第二副边绕组102输出的电压进行储能再释放的电压叠加，实现对第二副边绕组102输出的电压进行升压，使得在电源提供的电压的负半周期，从输出端Vo的输出电压不变。
因此，本实施例中，通过将第一副边绕组的第一端和第二副边绕组的第二端均与升压电路连接，第一副边绕组的第二端和第二副边绕组的第一端串联且接地；然后升压电路在电源提供的电压的正半周期对第一副边绕组输出的电压进行升压，在电源提供的电压的负半周期对第二副边绕组输出的电压进行升压，使得无论是在正半周期还是负半周期，都存在副边绕组输出相同的电压到输出端对 负载放电，达到全波整流输出的效果，大大降低了电源输出电压的纹波。
本申请实施例提供的谐振变换器，该谐振变换器包括绕组变压器、升压电路；绕组变压器包括括至少一组副边绕组，每组副边绕组包括第一副边绕组和第二副边绕组；第一副边绕组与第二副边绕组串联；绕组变压器将电源提供的电压进行变压，并将变压后的电压在电源提供的电压的正半周期从第一副边绕组输出，将变压后的电压在电源提供的电压的负半周期从第二副边绕组输出，升压电路在电源提供的电压的正半周期对第一副边绕组输出的电压进行升压，在电源提供的电压的负半周期对第二副边绕组输出的电压进行升压。通过设置升压电路对变压器第一副边绕组和第二副边绕组输出的电压升压，保证输出端输出到负载电压幅值不变，然后增大绕组变压器原边绕组与副边绕组之间的匝比避免了输出功率降低，以通过减小磁芯横截面积减小绕组变压器的体积，从而减小电源产品体积；同时，每个半周期，绕组变压器的第一副边绕组和第二副边绕组输出的电压都可以经过升压电路升压，实现第一副边绕组和第二副边绕组在每半个周期内均有一个副边绕组对负载供电，从而可以达到全波整流输出的效果，大大降低电源输出电压的纹波。因此，本申请实施例提供的谐振变换器能够减小电源产品的体积的同时减小电源输出纹波。
在上述实施例的基础上，提供一种的升压电路内部实现结构的实施例，如图3所示，该实施例中，每组副边绕组对应的一个升压电路，该升压电路20包括第一充放电电路201和第二充放电电路202；第一充放电电路201的第一端与第一副边绕组101的第一端连接，第二充放电电路202的第一端与第二副边绕组102的第二端连接，第一充放电电路201的第二端和第二充放电电路202的第二端均接地；第一充放电电路201，用于在电源提供的电压的负半周期充电，并在正半周期放电以对第一副边绕组101输出的电压进行升压；第二充放电电路202，用于在电源提供的电压的正半周期充电，并在负半周期放电以对第二副边绕组102输出的电压进行升压。
升压电路20中包括了第一充放电电路201和第二充放电电路202。充放电电路可以实现充电和放电，即第一充放电电路201在电源提供的电压的负半周期根据第一副边绕组输出的电压进行充电，并在正半周期放电；第二充放电电路202在电源提供的电压的正半周期根据第二副边绕组输出的电压进行充电，并在负半周期放电。具体地，在正半周期，原边绕组103的P1为正，P2为负，利用从第二副边绕组102输出的电压对第二充放电电路201进行充电；在负半周期，原边绕组103的P1为负，P2为正，利用从第一副边绕组101输出的电压对第一充放电电路201进行充电。
所以对于当前的正半周期，由于第一充放电电路201已经在上一个负半周期利用第一副边绕组101输出的电压完成了充电，那么第一充放电电路就会在当前的正半周期放电到输出端Vo以对负载放电，此时，在当前的正半周期第一副边绕组101本身输出的电压也会输出到输出端Vo以对负载放电。这样，对于输出端Vo来说，就是第一副边绕组101本身输出的电压和第一充放电电路放电的电压两者共同叠加，对负载放电，即对第一副边绕组101本身输出的较小的电压叠加第一充放电电路放电的电压，实现对第一副边绕组101输出的电压进行升压。
例如，假设匝比增加前，第一副边绕组101的输出电压为2V1，输出端Vo所需的固定电压也是2V1，而本申请实施例对匝比增加后，第一副边绕组101的输出电压降低为V1。那么利用第一副边绕组101对第一充放电电路201充电后，第一充放电电路201放电的输出电压也是V1，所以在当前的正半周期，第一副边绕组101的输出电压V1和第一充放电电路201放电的电压V1进行叠加后，输出端Vo输出的电压仍为2V1。
同样，对于当前的负半周期，由于第二充放电电路202已经在上一个正半周期利用第二副边绕组102输出的电压完成了充电，那么第二充放电电路就会在当前的负半周期放电到输出端Vo以对负载放电，此时，在当前的负半周期第二副边绕组102本身输出的电压也会输出到输出端Vo以对负载放电。这样，对于输出端Vo来说，就是第二副边绕组102本身输出的电压和第二充放电电路放电的电压两者共同叠加，对负载放电，即对第二副边绕组102本身输出的较小的电压叠加第二充放电电路放电的电压，实现对第二副边绕组102输出的电压进行升压。
例如，假设匝比增加前，第二副边绕组102的输出电压为2V1，输出端Vo所需的固定电压也是2V1，而本申请实施例对匝比增加后，第二副边绕组102的输出电压降低为V1。那么利用第二副边绕组102对第二充放电电路202充电后，第二充放电电路202放电的输出电压也是V1，所以在当前的负半周期，第二副边绕组102的输出电压V1和第二充放电电路202放电的电压V1进行叠加后，输出端Vo输出的电压也仍为2V1。
本申请实施例中，通过在升压电路中设置第一充放电电路和第二充放电电路，第一充放电电路的第一端与第一副边绕组的第一端连接，第二充放电电路的第一端与第二副边绕组的第二端连接，第一充放电电路的第二端和第二充放电电路的第二端均接地；第一充放电电路在电源提供的电压的负半周期充电，并在正半周期放电以对第一副边绕组输出的电压进行升压；第二充放电电路在电源提供的电压的正半周期充电，并在负半周期放电以对第二副边绕组输出的电压进行升压。两个充放电电路分别为利用不同的副边绕组输出的电压充电，然后在其对应副边绕组放电的周期内叠加放电，实现对对应副边绕组输出电压的升压，这样，通过储能的思路补偿增加匝比(原边绕组匝数不变，减少副边绕组匝数)后副边绕组输出电压较小的缺陷。且，无论是在正半周期还是在负半周期，均有一个副边绕组和对应的充放电电压对负载放电，使得输出电压的幅度变化较小，从而减小了输出电压的纹波。
对于充放电电路的具体电路实现方式，可以是任何方式，本申请实施例不作限定，只要能达到升压整流的目的即可。下面本申请实施例分别对第一充放电电路和第二充放电电路提供一种具体地电路实现。
在一个实施例中，如图4所示，第一充放电电路201包括第一电容C1、第一整流管2011和第二整流管2012；第一电容C1的第一端与第一副边绕组101的第一端连接，第一电容C1的第二端与第一整流管2011的第一端连接，第一整流管2011的第二端为谐振变换器的输出端Vo；第二整流管2012的第一端接地，第二整流管2012的第二端与第一电容C1的第二端、第一整流管2011的第一端连接；其中，第一电容C1在电源提供的电压的负半周期根据第一副边绕组输出的电压进行充电，在电源提供的电压的正半周期放电；第一整流管2011在电源提供的电压的正半周期导通，在电源提供的电压的负半周期关断；第二整流管2012在电源提供的电压的正半周期关断，在电源提供的电压的负半周期导通。
本实施例中，以第一充放电电路201包括第一电容C1、第一整流管2011和第二整流管2012为例进行说明。其中，第一电容C1可以充电和放电，第一整流管2011和第二整流管2012进行整流以及滤波，即将交流电整流及滤波为直流电。
根据前面实施例的说明可知，第一充放电电路201是在负半周期利用第一副边绕组101输出的电压充电，到了正半周期在第一副边绕组101输出的电压基础上叠加放电，所以第一充放电电路 201中的第一电容是在负半周期利用第一副边绕组101输出的电压充电，到了正半周期在第一副边绕组101输出的电压基础上叠加放电。而第一整流管2011在正半周期导通，在负半周期关断；第二整流管2012与第一整流管2011相反，在正半周期关断，在负半周期导通。
具体地，请参见图4，在负半周期时，原边绕组103的P1为负，P2为正，则第一副边绕组101的S2端大于S1端，第一副边绕组101输出的电压经第二整流管2012后进入第一电容C1为C1充电；等到了正半周期，原边绕组103的P1为正，P2为负，则第一副边绕组101输出的电压经过C1后，再进入第一整流管2011整流滤波后从输出端Vo输出对负载放电，同时第一电容C1也在放电，其放电的电压也是经过第一整流管2011整流滤波后从输出端Vo输出对负载放电。
可选地，第一充放电电路201的第一电容可以是一个自举电容，也可以是多个并联的第一电容，本申请实施例第一电容的数量不做限定。通过并联第一电容可以减小每个第一电容的电容值，降低第一充放电电路的实现难度。
可选地，第一整流管2011和第二整流管2012均为二极管；或者，第一整流管2011和第二整流管2012均为开关管。可选地，开关管可以是MOS管或者GaN、SiC等器件，从而减少开关管的开关损耗和导通损耗，提高可靠性，本申请实施例对开关管的具体不作限定，只要实现整流开关功能即可。
请参见图5所示，为第一整流管2011和第二整流管2012均为二极管的示意图，该实施例中，第一整流管2011的第一端和第二整流管2012的第一端指的均是二极管的阳极；第一整流管2011的第二端和第二整流管2012的第二端指的均是二极管的阴极。其中，第一整流管2011为二极管D1，第二整流管2012为二极管D2，则二极管D1阳极与第一电容C1的第二端连接，二极管D1的阴极连接谐振变换器的输出端Vo；二极管D2的阳极接地，二极管D2的阴极与第一电容C1的第二端、二极管D1的阳极连接。且二极管D1在电源提供的电压的正半周期导通，在电源提供的电压的负半周期关断；二极管D2在电源提供的电压的正半周期关断，在电源提供的电压的负半周期导通。
请参见图6所示，为第一整流管2011和第二整流管2012均为MOS管的示意图，该实施例中，第一整流管2011的第一端和第二整流管2012的第一端指的均是开关管的源极；第一整流管2011的第二端和第二整流管2012的第二端指的均是开关管的漏极。其中，第一整流管2011为MOS管Q1，第二整流管2012为MOS管Q2，则MOS管Q1源极与第一电容C1的第二端连接，MOS管Q1的漏极连接谐振变换器的输出端Vo；MOS管Q2的源极接地，MOS管Q2的漏极与第一电容C1的第二端、MOS管Q1的源极连接。且MOS管Q1在电源提供的电压的正半周期导通，在电源提供的电压的负半周期关断；MOS管Q2在电源提供的电压的正半周期关断，在电源提供的电压的负半周期导通。其中，MOS管的栅极(控制极)连接以驱动控制器(图中未示意)，例如半桥驱动器等，其通过控制极接收控制信号，根据控制信号导通或关断。本申请实施例MOS管的控制极不作限定。
本实施例中，第一充放电电路以第一电容、第一整流管和第二整流管实现，第一电容是在负半周期利用第一副边绕组输出的电压充电，到了正半周期在第一副边绕组输出的电压基础上叠加放电，实现对第一副边绕组输出的电压的升压，保证了输出端的输出电压的稳定性。且第一副边绕组输出的电压和第一电容放电的电压均会经过整流管进行整流滤波，将交流电转换为直流电，以从输 出端输出电压对负载放电。
在一个实施例中，如图7所示，第二充放电电路202包括第二电容C2、第三整流管2013和第四整流管2014；第二电容C2的第一端与第二副边绕组102的第二端连接，第二电容C2的第二端与第四整流管2014的第一端连接，第四整流管2014的第二端连接谐振变换器的输出端Vo；第三整流管2013的第一端接地，第三整流管2013的第二端与第二电容C2的第二端、第四整流管2014的第一端连接；其中，第二电容C2在电源提供的电压的正半周期根据第二副边绕组输出的电压进行充电，在电源提供的电压的负半周期放电；第三整流管2013在电源提供的电压的正半周期导通，在电源提供的电压的负半周期关断；第四整流管2014在电源提供的电压的正半周期关断，在电源提供的电压的负半周期导通。
本实施例中，以第二充放电电路202包括第二电容C2、第三整流管2013和第四整流管2014为例进行说明。同样，第二电容C2可以充电和放电，第三整流管2013和第四整流管2014进行整流以及滤波，即将交流电整流及滤波为直流电。
结合前面实施例的说明可知，第二充放电电路202是在正半周期利用第二副边绕组102输出的电压充电，到了负半周期在第二副边绕组102输出的电压基础上叠加放电，所以第二充放电电路202中的第二电容C2也是在正半周期利用第二副边绕组102输出的电压充电，到了负半周期在第二副边绕组102输出的电压基础上叠加放电。而第三整流管2013在正半周期导通，在负半周期关断；第四整流管2014与第三整流管2013相反，在正半周期关断，在负半周期导通。
具体地，请参见图7，在正半周期时，原边绕组103的P1为正，P2为负，则第二副边绕组102的S3端大于S4端，第二副边绕组102输出的电压经第三整流管2013后进入第二电容C2为C2充电；等到了负半周期，原边绕组103的P1为负，P2为正，第二副边绕组102的S4端大于S3端，则第二副边绕组102输出的电压经过C2后，再进入第四整流管2014整流滤波后从输出端Vo输出对负载放电，同时第二电容C2也在放电，其放电的电压也是经过第四整流管2014整流滤波后从输出端Vo输出对负载放电。
对于第二电容，在实际应用时，也可以是一个自举电容，或者是多个并联的第一电容，本申请实施例第二电容的数量不做限定。通过并联第二电容可以减小每个第二电容的电容值，降低第二充放电电路的实现难度。
可选地，第三整流管2013和第四整流管2014均为二极管；或者，第三整流管2013和第四整流管2014均为开关管。可选地，开关管可以是MOS管或者GaN、SiC等器件，从而减少开关管的开关损耗和导通损耗，提高可靠性，本申请实施例对开关管的具体不作限定，只要实现整流开关功能即可。
请参见图8所示，为第三整流管2013和第四整流管2014均为二极管的示意图，该实施例中，第三整流管2013的第一端和第四整流管2014的第一端指的均是二极管的阳极；第三整流管2013的第二端和第四整流管2014的第二端指的均是二极管的阴极。其中，第三整流管2013为二极管D3第四整流管2014为二极管D4，则二极管D3阳极接地，二极管D3的阴极与第二电容C2的第二端、二极管D4的阳极连接；二极管D4的阳极连接第二电容C2的第二端，二极管D4的阴极连接谐振变换器的输出端Vo。且二极管D3在电源提供的电压的正半周期导通，在电源提供的电压的负半周期关断；二极管D4在电源提供的电压的正半周期关断，在电源提供的电压的负半周期导 通。
请参见图9所示，为第三整流管2013和第四整流管2014均为MOS管的示意图，该实施例中，第三整流管2013的第一端和第四整流管2014的第一端指的均是开关管的源极；第三整流管2013的第二端和第四整流管2014的第二端指的均是开关管的漏极。其中，第三整流管2013为MOS管Q3，第四整流管2014为MOS管Q4，则MOS管Q3的源极接地，MOS管Q3的漏极与第二电容C2的第二端、MOS管Q4的源极连接；MOS管Q4的漏极连接谐振变换器的输出端Vo；且MOS管Q3在电源提供的电压的正半周期导通，在电源提供的电压的负半周期关断；MOS管Q4在电源提供的电压的正半周期关断，在电源提供的电压的负半周期导通。其中，本实施例中，MOS管的栅极(控制极)也是连接以驱动控制器(图中未示意)，例如半桥驱动器等，其通过控制极接收控制信号，根据控制信号导通或关断。本申请实施例MOS管的控制极不作限定。
上述第一充放电电路201和第二充放电电路202中的二极管和开关管，均可以根据副边绕组的输出电压导通或者关断。并且，采用本申请实施例的结构，可以降低整流管的反向耐压，从而提高电路的可靠性。
本实施例中，第二充放电电路以第二电容、第三整流管和第四整流管实现，第二电容是在正半周期利用第二副边绕组输出的电压充电，到了负半周期在第二副边绕组输出的电压基础上叠加放电，实现对第二副边绕组输出的电压的升压，保证了输出端的输出电压的稳定性。且第二副边绕组输出的电压和第二电容放电的电压均会经过整流管进行整流滤波，将交流电转换为直流电，以从输出端输出电压对负载放电。
如图10和图11所示，是分别以二极管和MOS管为例，结合上述第一充电放电电路201和第二充放电电路202的内部电路结构结合的一种谐振变换器，图10中各电路的实现可参见前述说明，在此不再赘述。从以上第一充电放电电路201和第二充放电电路202的内部电路结构以及各自在正半周期和负半周期的工作情况，可以看出，无论是正半周期还是负半周期，本申请实施例提供的谐振变换器中均有副边绕组和充放电电路对负载叠加放电，弥补了增加了匝比后副边输出电压的减小会导致输出端输出的电压降低的现象，又保证了正负两个半周期切换过程中，输出端输出的电压的稳定性，减小了输出电源的纹波；而增加匝比是为了减小磁芯面积从而减小变压器体积，进而减小电源产品的体积，所以实现了减小电源产品的体积的同时减小电源输出纹波。
在以上实施例的基础上，在一个实施例中，升压电路中还可以包括第三充放电电路203；第三充放电电路203的第一端连接在第一充放电电路201的第二端，第三充放电电路203的第二端接地；第三充放电电路203，用于在正半周期与第一充放电电路叠加对第一副边绕组的输出电压，并在电源的负半周期与第二充放电电路叠加对对第二副边绕组的输出电压进行升压。可选地，第三充放电电路203包括第三电容C3；第三电容C3的第一端连接在第一充放电电路201的第二端，第三电容C3的第二端接地。
其中，第三充放电电路203中的第三电容C3可以是多个并联的电容，也可以是一个电容，本申请实施例对此不作限定。
如图12所示，以MOS管为例，结合第三电容C3(第三充放电电路203)和上述第一充电放电电路201和第二充放电电路202的内部电路结构结合的一种谐振变换器。
第三充放电电路203的第一端连接在第一充放电电路201的第二端，第一充放电电路201的第 二端是电压经过升压整流输出到输出端Vo的位置，第三充放电电路的第二端接地。所以在正半周期，第一副边绕组101和第一充放电电路201在叠加向输出端输出电压时，会同时对第三充放电电路充电，且由于是叠加充电，第三充放电电路会快速充满，充满到峰值后，在第一副边绕组101、第一充放电电路201的基础上进一步叠加对输出端Vo输出电压以对负载放电。
在负半周期相同，第一充放电电路201的第二端是电压经过升压整流输出到输出端Vo的位置，此位置与第二充放电电路202的第二端输出电压的位置点实质上属于同一点，所以第三充放电电路203的第一端也相当于连接在第二充放电电路202的第二端，因此，在负半周期，第二副边绕组102和第二充放电电路202在叠加向输出端输出电压时，也会同时对第三充放电电路充电，且也由于是叠加充电，第三充放电电路会快速充满，充满到峰值后，在第二副边绕组102、第二充放电电路202的基础上也会进一步叠加对输出端Vo输出电压以对负载放电。
因此，第三充放电电路203是在正半周期，利用第一副边绕组101的输出电压充电然后充满后在正半周期内继续放电，实现与第一充放电电路叠加对第一副边绕组101的输出电压进行升压的效果，在负半周期，利用第二副边绕组102的输出电压充电然后充满后在负半周期内继续放电，实现与第二充放电电路叠加对第二副边绕组102的输出电压进行升压的效果。因此，通过设置第三充放电电路203可以进一步保证输出端的输出电压的稳定性。
可选地，针对多个负载的场景，可以以绕组变压器中包括多组副边绕组的情况进行说明，其中该多组副边绕组之间并联设置，又每组副边绕组对应一个升压电路，即需设置多个并联的升压电路。如图13所示，为在上述图12的基础上，即仍以MOS管为例，提供一种升压电路并联的谐振变换器示意图。图13中，包括m组副边绕组(升压电路1到升压电路m)，每组副边绕组对应一升压电路20。其中，升压电路1到升压电路m(中间的升压电压以省略号替代)均为相同电路，所以V1O到VnO输出的电压相同，从而实现对不同负载供电的效果。
前面实施例有提及，第一副边绕组101和第二副边绕组102可以为多个并联的副边绕组构成，那么应用在多个负载的场景中，在一个实施例中，还可以将副边绕组并联(指第一副边绕组为n个第一副边绕组并联，第二副边绕组为n个第二副边绕组并联)和多组副边绕组并联(对应多个升压电路并联)的情况相结合，则如图14所示，为结合上述图13和上述图2后提供的一种谐振变换器的示意图。图14中，副边绕组的并联使得变压器的原副边匝比进一步增大，这样提高原边绕组和副边绕组的匝比可以进一步减小磁感应强度，从而降低损耗，避免输出功率下降。而增大匝比后就可以减小磁芯横截面积，进而减小绕组变压器的体积，使得应用在多个负载的场景中，也可以是实现减少绕组变压器的体积来减小电源产品体积的目的。可以理解的是，图13和图14中的m以及图2中的n均表示的是多个，且均为正整数，为了区分而采用n和m来表示，实际应用时，n可以等于m，也可以不相等，本申请实施例对此不作限定。
针对图13和图14中各升压电路的工作原理、过程以及实现效果与前面各实施例中相同，在此不再赘述。
另外，本申请实施例还提供了一种电源，该电源包括前面实施例中所提供的任一种谐振变换器。在一个实施例中，还提供了一种终端，该终端包括任一种谐振变换器构成的电源。
上述实施例谐振变换器包括绕组变压器和升压电路；在电源提供的电压的正半周期和负半周期均对副边绕组的输出电压进行升压。通过设置升压电路对变压器副边绕组输出的电压升压，保证输 出端输出到负载电压不变，然后增大绕组变压器原边绕组与副边绕组之间的匝比避免了输出功率降低，以通过减小磁芯横截面积减小绕组变压器的体积，从而减小电源产品体积。
终端表示任何需要外接电源或者内置电源的电子设备，例如，各种个人计算机、笔记本电脑、手机(智能移动终端)、平板电脑和便携式可穿戴装置等，本实施例对此不做限定。若是外置电源，该电源可以是电源适配器、移动电源(充电宝、旅充)等，本实施例对此也不做限定。当然，除了终端，还可以是需要电源的设备，例如，电动汽车、无人机、电子书、电子烟、智能电子设备(包括手表、手环、智能眼镜、扫地机器人等)、小型电子产品(包括无线耳机、蓝牙音响、电动牙刷、可充电无线鼠标等)，也可以是(5G)通讯模块电源等等，本申请实施例对此均不作限定。
现阶段高功率充电器时发展趋势，但是减小体积增加功率密度提高便携性是当前考虑的主要问题，对充电器来说电容和磁性器件占整个充电器的体积的80％，采用本申请实施例提供的谐振变换器既可以减小变压器的体积又可以较小输出电容的体积，并减少电源输出纹波，提升电源输出电压的质量，进一步提升电源的功率密度。本申请实施例提供的谐振变换器不仅适用于LLC架构电源，对于移相全桥、DCX等多这种需要副边采用全波整流的电路，通过倍压整流输出，均可以采用本申请实施例减小变压器和输出电容的体积的同时减少电源输出纹波。
另外，在一个实施例中，本申请实施例还提供了一种升压方法的实施例，如图15所示，该实施例涉及的是通过运行计算机程序实现对谐振变换器的输出到负载的电压进行升压的具体过程。则该实施例包括：
S101，将电源提供的电压进行变压，在电源提供的电压的正半周期得到第一电压，在所述提供的电压的负半周期得到第二电压。
S102，在电源提供的电压的正半周期对第一电压进行升压，在电源提供的电压的负半周期对第二电压进行升压。
其中，可以预先设置一用于指示变压的程序指令，在计算机设备接收到该程序指令的触发后，执行相应的操作，即根据预设的配置，对电源提供的电压进行变压，并输出变压后的电压，且输出的变压包括第一电压和第二电压。这里的第一电压和第二电压的电压值可以相等，也可以不相同，本申请实施例对此不作限定。另外，第一电压和第二电压的区别可以是从不同的端口输出，例如，以包括两个串联的副边绕组的变压器为例，第一电压可以是从变压器的第一副边绕组输出，第二电压可以是从变压器的第二副边绕组输出；反之，第一电压可以是从变压器的第二副边绕组输出，第二电压可以是从变压器的第一副边绕组输出，本申请实施例对此也不作限定。
对于输出的第一电压和第二电压，计算机设备可继续执行预设的程序指令，对其进行升压，且升压时，是在电源提供的电压的正半周期对第一电压进行升压，在电源提供的电压的负半周期对第二电压进行升压。例如，仍以包括两个串联的副边绕组的变压器为例，在正半周期对第一副边绕组输出的电压升压，在负半周期则对第二副边绕组输出的电压升压。又例如，升压时，可以通过设置充放电装置来实现，具体地，可以是在电源提供的电压的正半周期，利用第一电压对充放电装置充电，这样到了负半周期，充放电装置就会放电，放电的电压就会输出给负载以对负载供电，而同时第二电压本身也会输出给负载对负载供电，这样，充放电装置的放电电压和第二电压叠加对负载供电，从而实现对第二电压升压效果。同理。在电源提供的电压的负半周期，利用第二电压也对充放电装置充电，这样到了正半周期，充放电装置也会放电，放电的电压同样会输出给负载以对负载供 电，而同时在正半周期第一电压本身也会输出给负载对负载供电，这样，充放电装置的放电电压和第一电压叠加对负载供电，从而实现对第一电压升压效果。因此，在电源提供的电压的正半周期和负半周期均对变压后的电压实现了升压，这样对于负载来说，其在正半周期和负半周期，接受的电压变化的幅度较小，整体较为稳定，从而减少了输出到负载的电压的纹波的效果。
可以理解的是，以上过程通过计算机程序指令实现，这些计算机程序指令提供到通用计算机、专用计算机、嵌入式处理机或其他可编程数据处理设备的处理器中，使得通过该计算机或其他可编程数据处理设备的处理器执行的指令可实现本实施例升压以减小输出电压纹波的功能。当然，这些计算机程序指令也可存储在能引导计算机或其他可编程数据处理设备以特定方式工作的计算机可读存储器中，使得存储在该计算机可读存储器中的指令产生包括指令装置的制造品。或者，这些计算机程序指令也可装载到计算机或其他可编程数据处理设备上，使得在计算机或其他可编程设备上执行一系列操作步骤以产生计算机实现的处理，从而在计算机或其他可编程设备上执行该计算机程序指令实现上述功能。
以上实施例的各技术特征可以进行任意的组合，为使描述简洁，未对上述实施例中的各个技术特征所有可能的组合都进行描述，然而，只要这些技术特征的组合不存在矛盾，都应当认为是本说明书记载的范围。
以上所述实施例仅表达了本申请的几种实施方式，其描述较为具体和详细，但并不能因此而理解为对发明专利范围的限制。应当指出的是，对于本领域的普通技术人员来说，在不脱离本申请构思的前提下，还可以做出若干变形和改进，这些都属于本申请的保护范围。因此，本申请专利的保护范围应以所附权利要求为准。

Claims (11)

  1. 一种谐振变换器，其特征在于，所述谐振变换器包括绕组变压器、升压电路；所述绕组变压器包括至少一组副边绕组，每组所述副边绕组包括第一副边绕组和第二副边绕组；所述第一副边绕组与所述第二副边绕组串联；
    所述绕组变压器，用于将电源提供的电压进行变压，并将变压后的电压在所述电源提供的电压的正半周期从所述第一副边绕组输出，将变压后的电压在所述电源提供的电压的负半周期从所述第二副边绕组输出；
    所述升压电路，用于在所述电源提供的电压的正半周期对所述第一副边绕组输出的电压进行升压，在所述电源提供的电压的负半周期对所述第二副边绕组输出的电压进行升压。
  2. 根据权利要求1所述的谐振变换器，其特征在于，每组所述副边绕组对应的所述升压电路包括第一充放电电路和第二充放电电路；所述第一充放电电路的第一端与所述第一副边绕组的第一端连接，所述第二充放电电路的第一端与所述第二副边绕组的第二端连接，所述第一充放电电路的第二端和所述第二充放电电路的第二端均接地；
    所述第一充放电电路，用于在所述电源提供的电压的负半周期根据所述第一副边绕组输出的电压进行充电，并在正半周期放电以对所述第一副边绕组输出的电压进行升压；
    所述第二充放电电路，用于在所述电源提供的电压的正半周期根据所述第二副边绕组输出的电压进行充电，并在负半周期放电以对所述第二副边绕组输出的电压进行升压。
  3. 根据权利要求2所述的谐振变换器，其特征在于，所述第一充放电电路包括第一电容、第一整流管和第二整流管；所述第一电容的第一端与所述第一副边绕组的第一端连接，所述第一电容的第二端与所述第一整流管的第一端连接，所述第一整流管的第二端为所述谐振变换器的输出端；所述第二整流管的第一端接地，所述第二整流管的第二端与所述第一电容的第二端、所述第一整流管的第一端连接；其中，
    所述第一电容在所述电源提供的电压的负半周期根据所述第一副边绕组输出的电压进行充电，在所述电源提供的电压的正半周期放电；
    所述第一整流管在所述电源提供的电压的正半周期导通，在所述电源提供的电压的负半周期关断；
    所述第二整流管在所述电源提供的电压的正半周期关断，在所述电源提供的电压的负半周期导通。
  4. 根据权利要求3所述的谐振变换器，其特征在于，所述第一整流管和所述第二整流管均为二极管；或者，所述第一整流管和所述第二整流管均为开关管。
  5. 根据权利要求2-4任一项所述的谐振变换器，其特征在于，所述第二充放电电路包括第二电容、第三整流管和第四整流管；所述第二电容的第一端与所述第二副边绕组的第二端连接，所述第二电容的第二端与所述第四整流管的第一端连接，所述第四整流管的第二端为所述谐振变换器的输出端；所述第三整流管的第一端接地，所述第三整流管的第二端与所述第二电容的第二端、所述第四整流管的第一端连接；其中，
    所述第二电容在所述电源提供的电压的正半周期根据所述第二副边绕组输出的电压进行充电，在所述电源提供的电压的负半周期放电；
    所述第三整流管在所述电源提供的电压的正半周期导通，在所述电源提供的电压的负半周期关断；
    所述第四整流管在所述电源提供的电压的正半周期关断，在所述电源提供的电压的负半周期导通。
  6. 根据权利要求5所述的谐振变换器，其特征在于，所述第三整流管和所述第四整流管均为二极管；或者，所述第三整流管和所述第四整流管均为开关管。
  7. 根据权利要求2所述的谐振变换器，其特征在于，所述升压电路还包括第三充放电电路；所述第三充放电电路的第一端连接在所述第一充放电电路的第二端，所述第三充放电电路的第二端接地；
    所述第三充放电电路，用于在所述正半周期与所述第一充放电电路叠加对所述第一副边绕组的输出电压进行升压，并在所述负半周期与所述第二充放电电路叠加对所述第二副边绕组的输出电压进行升压。
  8. 根据权利要求7所述的谐振变换器，其特征在于，所述第三充放电电路包括第三电容；所述第三电容的第一端连接在所述第一充放电电路的第二端，所述第三电容的第二端接地。
  9. 一种电源，其特征在于，所述电源包括如权利要求1-8任一项所述的谐振变换器。
  10. 一种终端，其特征在于，所述终端包括所述权利要求9所述的电源。
  11. 一种升压方法，其特征在于，所述方法包括：
    将电源提供的电压进行变压，在所述电源提供的电压的正半周期得到第一电压，在所述电源提供的电压的负半周期得到第二电压；
    在所述电源提供的电压的正半周期对所述第一电压进行升压，在所述电源提供的电压的负半周期对所述第二电压进行升压。
PCT/CN2022/086351 2021-05-14 2022-04-12 谐振变换器、电源、终端和升压方法 WO2022237431A1 (zh)

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