WO2016143121A1 - 交流回転機の制御装置および電動パワーステアリングの制御装置 - Google Patents
交流回転機の制御装置および電動パワーステアリングの制御装置 Download PDFInfo
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- WO2016143121A1 WO2016143121A1 PCT/JP2015/057313 JP2015057313W WO2016143121A1 WO 2016143121 A1 WO2016143121 A1 WO 2016143121A1 JP 2015057313 W JP2015057313 W JP 2015057313W WO 2016143121 A1 WO2016143121 A1 WO 2016143121A1
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/08—Arrangements for controlling the speed or torque of a single motor
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/10—Arrangements for controlling torque ripple, e.g. providing reduced torque ripple
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- B—PERFORMING OPERATIONS; TRANSPORTING
- B62—LAND VEHICLES FOR TRAVELLING OTHERWISE THAN ON RAILS
- B62D—MOTOR VEHICLES; TRAILERS
- B62D5/00—Power-assisted or power-driven steering
- B62D5/04—Power-assisted or power-driven steering electrical, e.g. using an electric servo-motor connected to, or forming part of, the steering gear
- B62D5/0457—Power-assisted or power-driven steering electrical, e.g. using an electric servo-motor connected to, or forming part of, the steering gear characterised by control features of the drive means as such
- B62D5/046—Controlling the motor
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- G—PHYSICS
- G01—MEASURING; TESTING
- G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
- G01R17/00—Measuring arrangements involving comparison with a reference value, e.g. bridge
- G01R17/02—Arrangements in which the value to be measured is automatically compared with a reference value
- G01R17/04—Arrangements in which the value to be measured is automatically compared with a reference value in which the reference value is continuously or periodically swept over the range of values to be measured
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- G—PHYSICS
- G01—MEASURING; TESTING
- G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
- G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
- G01R19/0046—Arrangements for measuring currents or voltages or for indicating presence or sign thereof characterised by a specific application or detail not covered by any other subgroup of G01R19/00
- G01R19/0053—Noise discrimination; Analog sampling; Measuring transients
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P23/00—Arrangements or methods for the control of AC motors characterised by a control method other than vector control
- H02P23/0004—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
- H02P25/024—Synchronous motors controlled by supply frequency
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
Definitions
- the present invention relates to a control device for an AC rotating machine and a control device for an electric power steering that realizes improvement in current detection accuracy.
- a conventional motor control device In a conventional motor control device, the maximum phase switching noise that makes current detection impossible is mixed into current detection of two phases other than the non-detectable phase by pasting the duty of the current non-detectable phase to 100%. Is preventing. Further, this conventional motor control device estimates the phase current value of the undetectable phase based on the phase current values of the two phases other than the undetectable phase (see, for example, Patent Document 1).
- the PWM pulse on-timing is adjusted in the first half of the PWM1 period to secure a current detection section, and the PWM output width is adjusted in the second half (for example, Patent Documents). 2). By making such adjustments, both current detection accuracy and output can be ensured.
- Japanese Patent No. 5396948 Japanese Patent Laid-Open No. 11-4594 Japanese Patent No. 5161985 Japanese Patent No. 5178768
- the prior art has the following problems.
- the motor control device disclosed in Patent Document 1 has a high potential side switching element when the on-time of the maximum phase low potential side switching element is smaller than the phase current detection time ts. Is kept on and the low-potential side switching element is kept off, so that switching noise is not mixed into the phase current values of the two phases other than the maximum phase.
- Patent Document 1 in order to secure the phase current detection time ts, it is necessary to detect the phase current after ts / 2 from the center of one carrier cycle. For example, if Dth is 90% and the modulation rate is up to 100%, the mixing of noise can be prevented by using Patent Document 1.
- the on-time of the high-potential side switching element becomes longer than the on-time of the low-potential side switching element. For this reason, the heat generation state is biased. Further, when used in a state where the modulation rate exceeds 100%, the DUTY of the intermediate phase exceeds Dth. For this reason, at the current detection timing, the current detection accuracy deteriorates due to the influence of the switching noise of the intermediate phase.
- the frequency converter for an AC motor disclosed in Patent Document 2 can detect a current of two phases out of three phases without being affected by switching noise, and therefore has a good current detection accuracy.
- voltage harmonics occur due to the PWM output shifting back and forth.
- the influence of the voltage harmonics causes deterioration in performance such as noise or vibration.
- the present invention has been made in order to solve the above-described problems, and provides a control device for an AC rotating machine and a control device for an electric power steering that do not deteriorate current detection accuracy due to the influence of switching noise of other phases.
- the purpose is to obtain.
- a control device for an AC rotating machine includes a DC power source that outputs a DC voltage, an AC rotating machine having m sets of n-phase windings, where m is a natural number, n is a natural number of 3 or more, and m sets of A current detector for detecting each current value of the n-phase winding, a high potential side switching element, and a low potential side switching element, and the high potential side switching element and the low potential side switching element based on the on / off signal Is controlled by switching the DC voltage into an AC voltage and applying it to the winding, and the voltage command based on the difference between the current command of the AC rotating machine and the current detection value by the current detector.
- a current detector configured to output a current flowing through a current detection resistance element inserted in series with a low potential side switching element corresponding to at least the (n-1) phase of the power converter. Based on this, when detecting the current flowing through the n-phase winding, the current is detected at a fixed timing of two or more times in one cycle of the carrier wave signal, and a current detection value that does not include an error caused by switching noise is acquired. Is.
- the present invention even in a modulation scheme in which only a detection current including an error due to the effect of switching noise can be obtained at a single current detection timing, the fixed timing of two or more times in one cycle of the carrier signal. By detecting the current, current detection that does not include an error caused by switching noise can be realized. As a result, it is possible to obtain an AC rotating machine control device and an electric power steering control device that do not deteriorate current detection accuracy due to the influence of switching noise of other phases.
- Embodiment 1 of this invention It is a figure which shows the whole structure of the power converter device in Embodiment 1 of this invention. It is a flowchart which shows a series of arithmetic processing by the offset calculator in Embodiment 1 of this invention. In Embodiment 1 of this invention, it is the figure which showed three-phase applied voltage Vu1 ', Vv1', Vw1 'in case a modulation factor is 100%. It is operation
- 5 is a flowchart showing a series of operations for calculating detected currents Iu1, Iv1, and Iw1 when data at the first timing is selected in the arithmetic unit according to the first embodiment of the present invention.
- 6 is a flowchart showing a series of operations for calculating detected currents Iu1, Iv1, and Iw1 when data at the second timing is selected in the arithmetic unit according to the first embodiment of the present invention.
- Embodiment 1 of this invention it is a figure which shows the applied voltage at the time of employ
- Embodiment 2 of this invention it is the figure which showed three-phase applied voltage Vu1 ', Vv1', Vw1 'in case a modulation factor is 102%. It is the figure which showed which timing data is used in each electrical angle based on the determination result by the calculator in Embodiment 2 of this invention. It is another figure which showed which timing data is used in each electrical angle based on the determination result by the calculator in Embodiment 2 of this invention. In Embodiment 2 of this invention, it is the figure which showed the number of the detectable phases with respect to each voltage phase of 1 period of carrier wave signals. It is a figure which shows the whole structure of the power converter device in Embodiment 3 of this invention.
- Embodiment 3 of this invention it is the electric current detection value in 1 period of electrical angles when the AC rotary machine is carrying out fixed rotation with a certain control command. In Embodiment 3 of this invention, it is the electric current detection value in the range of the electrical angle 60deg when the AC rotary machine is carrying out fixed rotation with a certain control command. It is a figure which shows the whole structure of the power converter device in Embodiment 6 of this invention. It is a flowchart which shows a series of arithmetic processing by the offset calculator in Embodiment 6 of this invention. It is a flowchart which shows a series of arithmetic processing by the offset calculator in Embodiment 6 of this invention.
- Embodiment 6 of this invention it is the figure which showed 1st three-phase applied voltage Vu1 ', Vv1', Vw1 'in case a modulation factor is 100%.
- Embodiment 6 of this invention it is the figure which showed 2nd three-phase applied voltage Vu2 ', Vv2', Vw2 'in case a modulation factor is 100%.
- FIG. 1 is a diagram showing an overall configuration of a power conversion device according to Embodiment 1 of the present invention.
- the AC rotating machine 1 is a three-phase AC rotating machine in which three-phase windings U1, V1, and W1 are housed in a stator of the rotating machine. Examples of such a three-phase AC rotating machine include a permanent magnet synchronous rotating machine, an induction rotating machine, and a synchronous reluctance rotating machine.
- the present invention is not limited to any rotation as long as the AC rotating machine has a three-phase winding. A machine may be used.
- the DC power supply 2 outputs DC voltage Vdc to power converter 4a.
- the DC power supply 2 includes all devices that output a DC voltage, such as a battery, a DC-DC converter, a diode rectifier, and a PWM rectifier.
- the smoothing capacitor 3 is connected in parallel with the DC power source 2 and realizes a stable DC current by suppressing fluctuations in the bus current.
- an equivalent series resistance Rc and a lead inductance Lc exist in addition to the true capacitor capacitance C.
- the power converter 4a uses an inverse conversion circuit, that is, an inverter, and based on the on / off signals Qup1, Qun1, Qvp1, Qvn1, Qwp1, Qwn1, the high potential side switching elements Sup1, Svp1, Swp1, and the low potential side
- the switching elements Sun1, Svn1, and Swn1 are turned on / off.
- the power converter 4a converts the DC voltage Vdc input from the DC power source 2 and applies the voltage to the three-phase windings U1, V1, and W1 of the AC rotating machine 1.
- the AC rotating machine 1 is energized with currents Iu1, Iv1, and Iw1.
- the on / off signals Qup1, Qun1, Qvp1, Qvn1, Qwp1, and Qwn1 are on / off signals for turning on and off the switching elements Sup1, Sun1, Svp1, Svn1, Swp1, and Swn1, respectively, in the power converter 4a. is there.
- semiconductor switches Sup1 to Swn1 semiconductor switches such as IGBTs, bipolar transistors, and MOS power transistors and diodes connected in antiparallel can be used.
- the voltage command calculator 6 calculates the three-phase voltage commands Vu1, Vv1, and Vw1 related to the voltage applied to the three-phase winding for driving the AC rotating machine 1, and outputs it to the offset calculator 7a.
- the current command of the AC rotating machine 1 is set as a control command, and currents Iu1, Iv1, Iw1 flowing through a three-phase winding detected by a current detector 11a described later are
- current feedback control for calculating the three-phase voltage commands Vu1, Vv1, and Vw1 by proportional-integral control can be used. Since such a control method is a well-known technique, detailed description is abbreviate
- the offset calculator 7a calculates the three-phase applied voltages Vu1 ', Vv1', Vw1 'based on the three-phase voltage commands Vu1, Vv1, Vw1.
- FIG. 2 is a flowchart showing a series of calculation processing by the offset calculator 7a according to Embodiment 1 of the present invention.
- the offset calculator 7a assigns the three-phase voltage commands Vu1, Vv1, and Vw1 to the maximum phase Vmax1, the intermediate phase Vmid1, and the minimum phase Vmin1 in descending order.
- step S121 the offset calculator 7a determines whether or not the modulation factor is 90% or less, and if true (YES), executes step S122, and if the determination result is false (NO), the step. S123 is executed.
- the offset calculator 7a may perform the determination process in step S121 using the difference between the maximum phase Vmax1 and the minimum phase Vmin1 as shown in FIG. Good.
- the offset calculator 7a subtracts the maximum phase Vmax1 from all the three-phase voltage commands Vu1, Vv1, and Vw1, and adds 0.4 times the DC voltage Vdc to 3 Phase applied voltages Vu1 ′, Vv1 ′, and Vw1 ′ are calculated.
- the voltage commands Vu1, Vv1, and Vw1 are offset so that the voltage of the phase corresponding to the maximum phase matches 0.4 Vdc.
- 0.4 Vdc is equal to the maximum applied voltage that can secure the phase current detection time. Therefore, by executing step S122, all of the voltage commands Vu1, Vv1, and Vw1 are set so that the applied voltage of the phase corresponding to the maximum phase among the three-phase applied voltages matches the maximum value of the carrier wave signal 0.4Vdc. Is offset.
- step S123 when the process proceeds to step S123, the difference between the maximum phase Vmax1 and the minimum phase Vmin1 exceeds 0.9Vdc, and any voltage causes an offset of ⁇ 0.5Vdc to 0.
- the three-phase applied voltages Vu1 ′, Vv1 ′, and Vw1 ′ cannot be accommodated up to 4Vdc. Therefore, in this case, the voltage commands Vu1, Vv1, and Vw1 are offset so that the voltage of the phase corresponding to the minimum phase matches ⁇ 0.5 Vdc.
- the modulation scheme when step S123 is executed is defined as “lower solid modulation”.
- the modulation method according to step S122 can be realized.
- the modulation rate exceeds 90%, the difference between the maximum phase Vmax1 and the minimum phase Vmin1 as shown in FIG. If the determination is made using, there may be a case where step S123 is performed depending on the angle.
- the threshold value of the modulation factor can be designed according to the actual machine.
- FIG. 3 is a diagram showing three-phase applied voltages Vu1 ′, Vv1 ′, and Vw1 ′ when the modulation factor is 100% in the first embodiment of the present invention.
- the horizontal axis represents the voltage phase ⁇ v [deg], and the vertical axis represents the ratio to the DC voltage Vdc.
- the three-phase voltage commands Vu1, Vv1, and Vw1 are sinusoidal waveforms having an amplitude of Vdc / ⁇ 3 with reference to 0.
- the applied voltage of the phase corresponding to the minimum phase is a lower solid modulation that is always ⁇ 0.5 Vdc, and corresponds to the maximum phase every 60 degrees.
- the applied voltage of the phase is 0.5 Vdc.
- Patent Document 1 obtains a two-phase current that is not affected by switching noise by setting the maximum phase to 0.5 Vdc.
- the on-time of the high-potential side switching element becomes longer than the on-time of the low-potential side switching element. For this reason, when the heat generation state is biased and is often used at a high rotation speed, the heat resistance performance of the low potential side switching element is sufficient, but the current limit is limited by the heat resistance performance of the high potential side switching element. Take it.
- the on / off signal generator 8 outputs on / off signals Qup1, Qun1, Qvp1, Qvn1, Qwp1, and Qwn1 based on the three-phase applied voltages Vu1 ', Vv1', and Vw1 '.
- FIG. 4 is an operation explanatory diagram of the on / off signal generator 8 according to Embodiment 1 of the present invention.
- C1 is a carrier wave signal, which is a triangular wave with a period Tc that has a minimum value of ⁇ 0.5 Vdc at t1 and t3 and a maximum value of 0.5 Vdc at t2 that is intermediate between t1 and t3.
- the first timing ts1 and the second timing ts2 indicate the current detection timing.
- the current detector 11a illustrated in FIG. 1 includes a current detection resistor element 9a and a calculator 10a. By providing the current detection resistance element 9a in series with each phase of the low potential side switching elements Sun1, Svn1, and Swn1 of the power converter 4a, the arithmetic unit 10a detects the current flowing through the three-phase winding.
- the arithmetic unit 10a detects the currents Iu11, Iv11, and Iw11 that flow through the three-phase winding at the first timing ts1, and the currents Iu12, Iv12, and Iw12 that flow through the three-phase winding at the second timing ts2. Is detected.
- the computing unit 10a calculates the detected currents Iu1, Iv1, and Iw1 from Iu11, Iv11, Iw11, and Iu12, Iv12, and Iw12 obtained from the current flowing through the current detection resistor element 9a.
- ti be the time required for the current detector 11a to detect the current. Specifically, this ti is determined by taking into account the ringing convergence time included in the detection waveform, the conversion time of the analog / digital converter, and the time required for sampling / holding, and the time for energizing the current detecting resistor element 9. Corresponds to the lower limit.
- the on / off signals Qup1, Qun1, Qvp1, Qvn1 related to the power converter 4a are between ts1-ti and ts1.
- Qwp1, Qwn1 need not be switched from 0 to 1 and from 1 to 0.
- the current detector 11a detects the on / off signals Qup1, Qun1, Qvp1 related to the power converter 4a between ts2-ti and ts2. , Qvn1, Qwp1, and Qwn1 need not be switched from 0 to 1 and from 1 to 0.
- the on / off signal is switched from 0 to 1 or from 1 to 0 after t2. This is because it is affected by generated noise.
- Vmax ' is 0.5 Vdc
- no switching operation occurs in one cycle of the carrier signal, and the high potential side switching element is always kept on.
- the maximum phase current cannot be detected, a two-phase current not including switching noise is detected.
- the first predetermined value is 0.4 Vdc, and there is no applied voltage that is 0.4 Vdc or more and less than 0.5 Vdc in any of the three phases. It is assumed that a current detection value not including switching noise can be obtained at the first timing ts1.
- the second timing ts2 may satisfy the following expression (3).
- FIG. 5 is a flowchart showing a series of arithmetic processing by the arithmetic unit 10a according to Embodiment 1 of the present invention. Specifically, the flowchart of FIG. 5 shows a procedure for determining which timing data is used as the current detection value.
- step S130 the arithmetic unit 10a determines whether or not the U-phase applied voltage Vu1 ′ is not less than 0.4 Vdc and less than 0.5 Vdc, and if true (YES), executes step S133, and if false (NO). Step S131 is executed.
- step S131 the arithmetic unit 10a determines whether or not the V-phase applied voltage Vv1 ′ is not less than 0.4 Vdc and less than 0.5 Vdc, and if true (YES), executes step S134, and if false (NO). Step S132 is executed.
- step S132 the arithmetic unit 10a determines whether or not the W-phase applied voltage Vw1 ′ is not less than 0.4 Vdc and less than 0.5 Vdc, and if true (YES), executes step S135, and if false (NO). Step S136 is executed.
- the arithmetic unit 10a uses the data detected at the second timing ts2 as the current detection value when the processing proceeds to step S133, step S134, and step S135, and the first timing when the processing proceeds to step S136.
- Data detected at ts1 is used as a current detection value.
- the determination is performed in the order of U, V, and W. However, the determination may be performed in another order.
- FIG. 6 is a diagram showing which timing data is used in each electrical angle based on the determination result by the arithmetic unit 10a in the first embodiment of the present invention.
- [1] indicates a range for selecting the detection result at the first timing ts1
- [2] indicates a range for selecting the detection result at the second timing ts2.
- [1] is selected when the maximum phase is 0.5 Vdc. I mean.
- FIG. 7 is a flowchart showing a series of operations for calculating the detection currents Iu1, Iv1, and Iw1 when the first timing data is selected in the arithmetic unit 10a according to the first embodiment of the present invention.
- the arithmetic unit 10a determines whether or not the U-phase applied voltage Vu1 ′ is less than 0.4 Vdc. If true (YES), execute step S141, and if false (NO), execute step S143. Execute.
- step S141 the arithmetic unit 10a determines whether or not the V-phase applied voltage Vv1 ′ is less than 0.4 Vdc. If true (YES), execute step S142, and if false (NO), execute step S144. Execute.
- step S142 the arithmetic unit 10a determines whether or not the W-phase applied voltage Vw1 ′ is less than 0.4 Vdc. If true (YES), the process executes step S146, and if false (NO), step S145 is performed. Execute.
- the calculation unit 10a can detect all currents, and sets each detected value as a detected current.
- the determination is performed in the order of U, V, and W, but the determination may be performed in another order.
- FIG. 8 is a flowchart showing a series of operations for calculating the detection currents Iu1, Iv1, and Iw1 when the second timing data is selected in the arithmetic unit 10a according to the first embodiment of the present invention.
- the arithmetic unit 10a determines whether or not the U-phase applied voltage Vu1 ′ is equal to or higher than 0.4 Vdc. If true (YES), execute step S153, and if false (NO), execute step S152. Execute.
- step S151 the arithmetic unit 10a determines whether or not the V-phase applied voltage Vv1 ′ is equal to or higher than 0.4 Vdc. If true (YES), execute step S154. If false (NO), execute step S152. Execute.
- step S152 the arithmetic unit 10a determines whether or not the W-phase applied voltage Vw1 ′ is 0.4 Vdc or more. If true (YES), step S155 is executed. If false (NO), step S156 is executed. Execute.
- step S156 the calculation unit 10a can detect all currents and sets each detected value as a detected current. However, since the applied voltage of any phase is 0.4 Vdc or more at the time selected in FIG. 5, step S156 is not actually used. In the flowchart of FIG. 8, the determination is performed in the order of U, V, and W. However, the determination may be performed in another order.
- the range where [2] is selected includes the case where it is affected by switching noise.
- the condition of the applied voltage so as not to be affected by the switching noise at the second timing ts2 is expressed by the following expression (4) from the same concept as the above expression (1). Here, it is 0.196 Vdc or less.
- the applied voltage of the V phase that is the intermediate phase is 0.27 Vdc. That is, the current detection value at the second timing ts2 includes an error due to V-phase switching noise.
- the region [2] before and after the electrical angle at which the intermediate phase and the maximum phase are equal is the corresponding location, and there are six regions in one cycle of the carrier wave signal.
- FIG. 9 shows the applied voltage when the voltage superimposition method (two-phase modulation) different from the lower solid modulation is adopted in the first embodiment of the present invention using the offset calculator 7b instead of the offset calculator 7a.
- the horizontal axis represents the voltage phase ⁇ v [deg], and the vertical axis represents the ratio to the DC voltage Vdc. Since two-phase modulation is a known technique, description of the flowchart and the like is omitted.
- the detection current is calculated based on the current value detected at the first timing, and at other times, the detection current is calculated based on the current value detected at the second timing.
- control device for an AC rotating machine in the first embodiment can be applied to a control device for an electric power steering using an AC rotating machine that generates a torque that assists the steering torque of the steering system.
- control device for an electric power steering control device that can constitute a steering system with a small torque ripple and noise.
- FIG. FIG. 10 is a diagram showing an overall configuration of the power conversion device according to Embodiment 2 of the present invention.
- the configuration of FIG. 10 in the second embodiment is different in the control unit 5c, the offset calculator 7c, the calculator 10b, and the current detector 11b. Therefore, these differences will be mainly described below.
- Tc is 50 ⁇ s and ti is 5 ⁇ s will be described as an example.
- the offset calculator 7c calculates the three-phase applied voltages Vu1 ', Vv1', Vw1 'based on the three-phase voltage commands Vu1, Vv1, Vw1.
- FIG. 11 is a flowchart showing a series of arithmetic processing by the offset calculator 7c in the second embodiment of the present invention.
- the offset calculator 7c assigns the three-phase voltage commands Vu1, Vv1, and Vw1 to the maximum phase Vmax1, the intermediate phase Vmid1, and the minimum phase Vmin1 in descending order.
- step S221 the offset calculator 7c determines whether or not the modulation factor is 90% or less, and if true (YES), executes step S222, and if false (NO), executes step S223. To do.
- the offset calculator 7c subtracts the maximum phase Vmax1 from all the three-phase voltage commands Vu1, Vv1, and Vw1, and adds 0.4 times the DC voltage Vdc to 3 Phase applied voltages Vu1 ′, Vv1 ′, and Vw1 ′ are calculated.
- the voltage commands Vu1, Vv1, and Vw1 are offset so that the voltage of the phase corresponding to the maximum phase matches 0.4 Vdc.
- 0.4 Vdc is equal to the maximum applied voltage that can secure the phase current detection time. Therefore, by executing step S222, all of the voltage commands Vu1, Vv1, and Vw1 are set so that the applied voltage of the phase corresponding to the maximum phase among the three-phase applied voltages matches the maximum value of the carrier wave signal 0.4Vdc. The voltage is offset.
- step S223 when the process proceeds to step S223, the difference between the maximum phase Vmax1 and the minimum phase Vmin1 exceeds 0.9Vdc, and any voltage causes an offset of ⁇ 0.5Vdc to 0.
- the three-phase applied voltages Vu1 ′, Vv1 ′, and Vw1 ′ cannot be accommodated up to 4Vdc. Therefore, in this case, the voltage commands Vu1, Vv1, and Vw1 are offset so that the voltage of the phase corresponding to the maximum phase matches 0.5 Vdc.
- the modulation method when step S223 is executed is defined as “upper solid modulation”. In the state where the modulation rate is up to 90%, the modulation method according to step S222 can be realized. However, when the modulation rate exceeds 90%, there is a case where step S223 occurs depending on the angle.
- the phase current detection time is 5 ⁇ s, and 90% is set as the threshold value.
- the modulation factor threshold value can be designed in accordance with the actual machine.
- step S221 was determined by the modulation rate, it may be determined by the difference between the maximum phase Vmax1 and the minimum phase Vmin1.
- FIG. 12 is a diagram showing three-phase applied voltages Vu1 ', Vv1', and Vw1 'when the modulation factor is 100% in the second embodiment of the present invention.
- the horizontal axis represents the voltage phase ⁇ v [deg], and the vertical axis represents the ratio to the DC voltage Vdc.
- the three-phase voltage commands Vu1, Vv1, and Vw1 are sinusoidal waveforms having an amplitude of Vdc / ⁇ 3 with reference to 0.
- step S222 By executing step S222, as shown in FIG. 12, the applied voltage of the phase corresponding to the maximum phase is always solid modulation of 0.5 Vdc, and the phase corresponding to the minimum phase is set every 60 degrees. The applied voltage of ⁇ 0.5 Vdc.
- the applied voltage of the phase corresponding to the intermediate phase exceeds 0.4 Vdc in the vicinity of 30 deg, 150 deg, and 300 deg. Therefore, when the uppermost modulation is performed, the current detection accuracy deteriorates due to the influence of the switching noise of the intermediate phase.
- Patent Document 1 in this region, a three-phase current that is not affected by switching noise is obtained by using a lower solid modulation without setting the maximum phase to 0.5 Vdc.
- FIG. 13 is a diagram showing three-phase applied voltages Vu1 ′, Vv1 ′, and Vw1 ′ when the modulation factor is 102% in the second embodiment of the present invention.
- the current detector 11b illustrated in FIG. 10 includes a current detection resistor element 9a and a calculator 10b.
- the arithmetic unit 10b detects the current flowing through the three-phase winding.
- the arithmetic unit 10b detects the currents Iu11, Iv11, and Iw11 that flow through the three-phase winding at the first timing ts1, and the currents Iu12, Iv12, and Iw12 that flow through the three-phase winding at the second timing ts2. Is detected.
- the computing unit 10b calculates the detection currents Iu1, Iv1, and Iw1 from Iu11, Iv11, Iw11, and Iu12, Iv12, and Iw12 obtained from the current flowing through the current detection resistance element 9a.
- the first timing ts1 is set to 2.4 ⁇ s after t2, and the second timing ts2 is set to 2.6 ⁇ s before t2.
- the implementation contents in the computing unit 10b will be described.
- FIG. 14 is a diagram showing which timing data is used in each electrical angle based on the determination result by the arithmetic unit 10b according to the second embodiment of the present invention.
- [1] indicates a range for selecting the detection result at the first timing ts1
- [2] indicates a range for selecting the detection result at the second timing ts2.
- a current detection value that does not include switching noise can be obtained by adopting the detection value at the first timing ts1. it can. In other areas, the detection value at the second timing ts2 can be used to obtain a current detection value that does not include switching noise. As a result, a current detection value that is not affected by switching noise can be obtained in the entire region.
- the switching element on the low potential side of the minimum phase is a timing different from the first timing, and when any one of the applied voltages of each phase is not less than the first predetermined value and less than the maximum value of the carrier signal.
- FIG. 15 is another diagram showing which timing data is used at each electrical angle based on the determination result by the arithmetic unit 10b in the second embodiment of the present invention, and the modulation rate is 102%.
- the waveform of the applied voltage in the case of is shown.
- the applied voltage of the minimum phase in the section detected at the second timing is small, but since the switching operation is performed at a timing further away from the second timing ts2, the current is affected by the switching noise even if the modulation rate is increased. The detection accuracy does not deteriorate.
- n is a natural number of 4 or more
- the n-phase application in which all the voltages of the n-phase voltage command are equally shifted so that the maximum phase voltage of the n-phase application voltages is equal to the maximum value of the carrier wave signal.
- FIG. 16 is a diagram showing the number of detectable phases with respect to each voltage phase of one period of the carrier signal in the second embodiment of the present invention. The number of detectable phases in one cycle of the carrier signal changes as shown in FIG.
- the current detection value of the minimum phase and one phase can be obtained.
- current detection values that are not affected by switching noise cannot be obtained.
- the region where ⁇ v is in the vicinity of 30 deg will be described.
- the following expression (5) is used by using the detection currents of all three phases obtained last time, the detection current of one phase obtained this time, and the angle change amount ⁇ from the previous time to the current time. ) May be used to calculate the remaining two-phase current.
- Iu1_old, Iv1_old, and Iw1_old in the above equation (5) are the detection currents Iu1, Iv1, and Iw1 obtained in the past, respectively.
- ⁇ represents an angle changed from when Iu1_old, Iv1_old, and Iw1_old are obtained until the current of one phase is detected this time.
- FIG. 17 is a diagram showing an overall configuration of the power conversion device according to Embodiment 3 of the present invention.
- the configuration of FIG. 17 in the third embodiment is different in the arithmetic unit 10c and the current detector 11c. Therefore, these differences will be mainly described below.
- Tc is 50 ⁇ s and ti is 5 ⁇ s will be described as an example.
- the case where the first timing ts1 is set to 2.4 ⁇ s after t2 and the second timing ts2 is set to 2.6 ⁇ s before t2 is the same as in the first and second embodiments.
- the implementation content in the computing unit 10c will be described.
- the timing for detecting the current is different by 5 ⁇ s between the first timing and the second timing, this causes an error in the current detection value.
- the three-phase detection currents Iu1, Iv1, and Iw1 are actually sine waves having a primary electrical angle when viewed macroscopically, but are actually realized by a 20 kHz PWM signal. For this reason, these three-phase detection currents Iu1, Iv1, and Iw1 are sine waves including a harmonic ripple of 20 kHz. That is, due to the 20 kHz harmonic ripple component, even if the current detection value at the first timing and the current detection value at the second timing are in the same state, a difference occurs in the current detection value.
- the detection current is obtained from the current detection value at the second timing while the detection current is obtained from the current detection value at the second timing in most regions. There are 3 times per cycle. At this time, if the current detection value at the second timing is used as it is, a third-order electric angle ripple is generated due to an error caused by a difference in detection timing.
- FIG. 18 is a current detection value in one cycle of electrical angle when the AC rotating machine is rotating at a constant speed with a certain control command in the third embodiment of the present invention.
- FIG. 19 is a current detection value in a range of an electrical angle of 60 deg when the AC rotating machine is rotating at a constant speed with a certain control command in the third embodiment of the present invention. Specifically, FIG. The section from 60 deg to 120 deg is enlarged and displayed.
- Lines 300 to 302 in FIG. 19 indicate the following.
- Line 300 Waveform of current detection value when detected before Tofs1 ( ⁇ s) from the center of one period of carrier wave signal
- Line 301 Waveform of current detection value when detected at the center of one period of carrier wave signal
- Line 302 Carrier wave signal 1 Waveform of current detection value when detected after Tofs2 ( ⁇ s) from the center of the cycle
- variable K is a parameter determined by the specifications of the AC rotating machine and the load voltage.
- Iu11 and Iu12 may be corrected to Iu11 'and Iu12' as shown in the following equation (9).
- V phase and W phase are corrected to Iv11 ′ and Iv12 ′ as in the following formula (10), and the W phase is corrected to Iw11 as in the following formula (11). What is necessary is just to correct
- the arithmetic unit 10c corrects the current detection value to be equivalent to the current detection value at the reference timing, and Iu11 ′, Iu12 ′, Iv11 ′, Iv12 ′, Iw11 ′, Iw12 ′. Are used to calculate currents Iu1, Iv1, and Iw1.
- K may be simplified and may be a constant, or in the case of fine adjustment, it may be a variable in accordance with the state such as the rotational speed, load voltage, and specification variation.
- the detection current of the reference timing is based on the coefficient proportional to the time difference Tofs1 between the reference timing and the first timing and the time difference Tofs2 between the reference timing and the second timing. It can be corrected considerably.
- the arithmetic unit 10c corrects the reference timing as the center of the carrier signal.
- the correction formula is expressed by the following formula (13).
- the arithmetic unit 10d corrects the current detection value corresponding to the current detection value at the first timing based on the equation (13) to obtain Iu11, Iu12 ′, Iv11, Iv12.
- the currents Iu1, Iv1, Iw1 are calculated using ', Iw11, Iw12'.
- K may be simplified and may be a constant, or in the case of fine adjustment, it may be a variable in accordance with the state such as the rotational speed, load voltage, and specification variation.
- the current detection value at the first timing is used in most areas. For this reason, with the reference timing as the first timing, the current detection value at the second timing is corrected to be equivalent to the current detection value at the first timing by the above equation (13), thereby reducing the correction frequency in one electrical angle cycle.
- the effect that it is possible can be acquired. Since the correction coefficients Tofs1K and Tofs2K are smaller than 1, the same effect can be obtained by the following equation (14).
- Embodiment 4 FIG.
- the case where the detection current is calculated by performing the correction for matching the reference timing by the correction formula using the correction coefficient set in advance has been described.
- the fourth embodiment a case will be described in which a correction amount according to a state in an actual machine is calculated online and corrected.
- FIG. 16 there is a section in which a two-phase current detection value can be obtained at the second timing in one electrical angle cycle. As can be seen from FIG. 14, in this section, a two-phase current detection value can be obtained even at the first timing.
- the correction processing in the fourth embodiment will be specifically described by taking an area from 50 deg to 130 deg as an example. In this region, Iv11, Iw11, Iv12, and Iw12 can be detected. Since the fundamental waves of the three-phase current are considered to be equal, the following equation (15) holds.
- the arithmetic unit 10e uses (1 + Tofs1K) / (1-Tofs2K) in the electrical angle region from 50 deg to 90 deg by using the following equation (16) based on the relationship between Iv11 and Iv12. ) Is calculated.
- the computing unit 10e can obtain the correction coefficient in the same way in other electrical angle regions.
- the arithmetic unit 10e calculates (1 + Tofs1K) / (1-Tofs2K) using the following equation (17) based on the relationship between Iw11 and Iw12 in the electrical angle region from 90 deg to 130 deg.
- the computing unit 10e can obtain the correction coefficient (1 + Tofs1K) / (1-Tofs2K) by calculating the ratio between the current detection value at the first timing and the current detection value at the second timing. Furthermore, the computing unit 10e can correct the current detection value at the second timing to be equivalent to the current detection value at the first timing as shown in the above equation (13) using the correction coefficient. As a result, by providing the computing unit 10e according to the fourth embodiment, it is possible to obtain an excellent effect that can be achieved by finely adjusting the correction amount online and reducing errors due to timing differences.
- the reference timing may be set at the center of one period of the carrier signal.
- the correction coefficient is obtained by the time difference Tofs1 of the first timing, the time difference Tofs2 of the reference timing and the second timing, and the variable K obtained by the above equation (18).
- the computing unit 10e calculates the ratio of the current detection value at the first timing to the current detection value at the second timing, thereby multiplying the time difference Tofs1 between the first timing and the time difference Tofs2 between the reference timing and the second timing. K can be obtained.
- the arithmetic unit 10e uses the correction coefficient obtained by multiplying the time difference Tofs1 of the first timing, the time difference Tofs2 of the reference timing and the second timing, and the variable K using the above equations (9) to (11) or ( As in 12), the current detection value at the first timing and the current detection value at the second timing can be corrected to be equivalent to the current detection value at the reference timing.
- the correction coefficient can also be obtained using a plurality of detectable current values. Also in this case, it is possible to obtain an unprecedented excellent effect of finely adjusting the correction amount online to reduce errors due to timing differences.
- Embodiment 5 FIG.
- the case where the correction coefficient (1 + Tofs1K) / (1-Tofs2K) or the coefficient K in the above equation (13) is calculated online to reduce the error due to the timing difference has been described.
- the fifth embodiment a case will be described in which correction is performed by a method different from that of the fourth embodiment.
- the difference between the latest current detection values in the past obtained in a state where both the first timing and the second timing can be detected is added to the current detection value of the second timing detected this time, and the current at the first timing is detected. Correction can be made corresponding to the detected value. Even with such a correction process, it is possible to obtain an excellent effect that has not been achieved in the past, by finely adjusting the correction amount online and reducing errors due to timing differences.
- the reference timing may be set at the center of one period of the carrier signal.
- the current detection value at the first timing is equivalent to the current detection value at the reference timing, and the current detection value at the second timing is converted to the reference timing.
- the reference timing has been described as the center of one period of the carrier signal, it is needless to say that correction can be performed in the same manner even if the reference timing is set to another timing. Even with such a correction process, it is possible to obtain an excellent effect that has not been achieved in the past, by finely adjusting the correction amount online and reducing errors due to timing differences.
- Embodiment 6 FIG.
- the case where the control device of the present invention is applied to an AC rotating machine having one winding set has been described.
- the control device of the present invention may be applied to an AC rotating machine having a plurality of winding sets. Therefore, in the sixth embodiment, a case will be described in which the control device of the present invention is applied to an AC rotating machine having two sets of three-phase windings having no phase difference.
- Tc is 50 ⁇ s
- ti is 5 ⁇ s
- the first predetermined value is 0.4 Vdc.
- FIG. 20 is a diagram showing an overall configuration of the power conversion device according to the sixth embodiment of the present invention.
- the first three-phase windings U1, V1, W1 and the second three-phase windings U2, V2, W2 are housed in the stator of the rotating machine without being electrically connected. It is a three-phase AC rotating machine.
- Examples of such a three-phase AC rotating machine include a permanent magnet synchronous rotating machine, an induction rotating machine, and a synchronous reluctance rotating machine.
- the present invention is not limited to any AC rotating machine having two three-phase windings. You may use the rotating machine of.
- the DC power supply 2 outputs DC voltage Vdc to first power converter 4a and second power converter 4b.
- the DC power supply 2 includes all devices that output DC voltage, such as a battery, a DC-DC converter, a diode rectifier, and a PWM rectifier.
- the smoothing capacitor 3 is connected in parallel with the DC power source 2 and realizes a stable DC current by suppressing fluctuations in the bus current.
- an equivalent series resistance Rc and a lead inductance Lc exist in addition to the true capacitor capacitance C.
- the first power converter 4a uses an inverse conversion circuit, that is, an inverter, on the basis of the on / off signals Qup1, Qun1, Qvp1, Qvn1, Qwp1, Qwn1, and the high potential side switching elements Sup1, Svp1, Swp1 and The low potential side switching elements Sun1, Svn1, Swn1 are turned on / off.
- the first power converter 4a converts the DC voltage Vdc input from the DC power supply 2 and applies a voltage to the three-phase windings U1, V1, and W1 of the AC rotating machine 1a. As a result, currents Iu1, Iv1, and Iw1 are energized in the AC rotating machine 1a.
- the on / off signals Qup1, Qun1, Qvp1, Qvn1, Qwp1, and Qwn1 are on / off signals for turning on and off the switching elements Sup1, Sun1, Svp1, Svn1, Swp1, and Swn1, respectively, in the power converter 4a. is there.
- semiconductor switches Sup1 to Swn1 semiconductor switches such as IGBTs, bipolar transistors, and MOS power transistors and diodes connected in antiparallel can be used.
- the second power converter 4b uses high voltage side switching elements Sup2, Svp2, Swp2 based on on / off signals Qup2, Qun2, Qvp2, Qvn2, Qwp2, Qwn2 using an inverse conversion circuit, that is, an inverter.
- the low potential side switching elements Sun2, Svn2, and Swn2 are turned on / off.
- the second power converter 4b converts the DC voltage Vdc input from the DC power supply 2 and applies a voltage to the three-phase windings U2, V2, and W2 of the AC rotating machine 1a.
- currents Iu2, Iv2, and Iw2 are energized in the AC rotating machine 1a.
- the on / off signals Qup2, Qun2, Qvp2, Qvn2, Qwp2, and Qwn2 are turned on / off to turn on and off the switching elements Sup2, Sun2, Svp2, Svn2, Swp2, and Swn2, respectively, in the second power converter 4b. Off signal.
- semiconductor switches Sup2 to Swn2 semiconductor switches such as IGBTs, bipolar transistors, and MOS power transistors and diodes connected in antiparallel can be used.
- the voltage command calculator 6 includes a first three-phase voltage command Vu1, Vv1, Vw1, and a second three-phase winding related to a voltage applied to the first three-phase winding for driving the AC rotating machine 1a.
- the second three-phase voltage commands Vu2, Vv2, and Vw2 related to the voltage applied to are calculated and output to the offset calculators 7d and 7e, respectively.
- the current command of the AC rotating machine 1a is set as a control command and detected by the first current detector 11f.
- the second three-phase voltage command Vu2 is controlled by proportional integral control.
- Vv2, and Vw2 and current feedback control can be used. Since such a control method is a well-known technique, detailed description is abbreviate
- the offset calculator 7d calculates the first three-phase applied voltages Vu1 ', Vv1', and Vw1 'based on the first three-phase voltage commands Vu1, Vv1, and Vw1.
- FIG. 21 is a flowchart showing a series of calculation processes by the offset calculator 7d in the sixth embodiment of the present invention.
- the offset calculator 7d assigns the first three-phase voltage commands Vu1, Vv1, and Vw1 to the first maximum phase Vmax1, the first intermediate phase Vmid1, and the first minimum phase Vmin1 in descending order.
- step S601 the offset calculator 7d determines whether or not the first modulation rate is 90% or less and the second modulation rate is 90% or less. If true (YES), step S602 is executed. If it is false (NO), step S603 is executed. As shown in FIG. 8 of Patent Document 3, the difference between the first maximum phase Vmax1 and the first minimum phase Vmin1 and the difference between the second maximum phase Vmax2 and the second minimum phase Vmin2 described later are used. Thus, it may be determined whether or not Vmax1 ⁇ Vmin1 ⁇ 0.9Vdc and Vmax2 ⁇ Vmin2 ⁇ 0.9Vdc.
- step S602 the offset calculator 7d subtracts the first maximum phase Vmax1 from all the voltages of the first three-phase voltage commands Vu1, Vv1, and Vw1, and adds 0.4 times the DC voltage Vdc.
- the first three-phase applied voltages Vu1 ′, Vv1 ′, and Vw1 ′ are calculated.
- the first voltage commands Vu1, Vv1, and Vw1 are offset so that the voltage of the phase corresponding to the first maximum phase matches 0.4 Vdc.
- 0.4 Vdc is equal to the maximum applied voltage that can secure the phase current detection time. Therefore, by executing step S602, the first voltage is set so that the applied voltage of the phase corresponding to the first maximum phase among the first three-phase applied voltages matches the maximum value of the carrier wave signal 0.4Vdc. All the voltages of the commands Vu1, Vv1, and Vw1 are offset.
- the difference between the first maximum phase Vmax1 and the first minimum phase Vmin1 or the difference between the second maximum phase Vmax2 and the second minimum phase Vmin2 exceeds 0.9 Vdc.
- at least one of the first three-phase applied voltage and the second three-phase applied voltage cannot be set between ⁇ 0.5 Vdc and 0.4 Vdc, regardless of the voltage offset. Therefore, in this case, the first voltage commands Vu1, Vv1, and Vw1 are offset by performing upper modulation so that the voltage of the phase corresponding to the first maximum phase matches 0.5 Vdc. .
- the phase current detection time is assumed to be 5 ⁇ s and 90% as a threshold value as in the examples of Patent Document 1 and Patent Document 3.
- the modulation factor threshold value can be designed in accordance with the actual machine.
- the offset calculator 7e calculates the second three-phase applied voltages Vu2 ', Vv2', and Vw2 'based on the second three-phase voltage commands Vu2, Vv2, and Vw2.
- FIG. 22 is a flowchart showing a series of calculation processes performed by the offset calculator 7e according to the sixth embodiment of the present invention.
- the offset calculator 7e substitutes the second three-phase voltage commands Vu2, Vv2, and Vw2 in descending order into the second maximum phase Vmax2, the second intermediate phase Vmid2, and the second minimum phase Vmin2.
- step S611 the offset calculator 7e determines whether or not the first modulation rate is 90% or less and the second modulation rate is 90% or less. If true (YES), step S612 is executed. If it is false (NO), step S613 is executed. Note that, as shown in FIG. 8 of Patent Document 3, Vmax1 ⁇ Vmin1 using the difference between the first maximum phase Vmax1 and the first minimum phase Vmin1 and the difference between the second maximum phase Vmax2 and the second minimum phase Vmin2. Whether or not ⁇ 0.9 Vdc and Vmax2 ⁇ Vmin2 ⁇ 0.9 Vdc may be determined. However, the determination methods of the offset calculator 7d and the offset calculator 7e should be aligned.
- the offset calculator 7e subtracts the second maximum phase Vmax2 from all the voltages of the second three-phase voltage commands Vu2, Vv2, and Vw2, and adds 0.4 times the DC voltage Vdc.
- the second three-phase applied voltages Vu2 ′, Vv2 ′, and Vw2 ′ are calculated.
- the second voltage commands Vu2, Vv2, and Vw2 are offset so that the voltage of the phase corresponding to the second maximum phase matches 0.4 Vdc.
- 0.4 Vdc is equal to the maximum applied voltage that can secure the phase current detection time. Therefore, by executing step S612, the second voltage is set such that the applied voltage of the phase corresponding to the second maximum phase among the second three-phase applied voltages matches the maximum value of the carrier wave signal 0.4Vdc. All the voltages of the commands Vu2, Vv2, and Vw2 are offset.
- the difference between the first maximum phase Vmax1 and the first minimum phase Vmin1 or the difference between the second maximum phase Vmax2 and the second minimum phase Vmin2 exceeds 0.9 Vdc.
- at least one of the first three-phase applied voltage and the second three-phase applied voltage cannot be set between ⁇ 0.5 Vdc and 0.4 Vdc, regardless of the voltage offset. Therefore, in this case, the second voltage commands Vu2, Vv2, and Vw2 are offset by performing upper modulation so that the voltage of the phase corresponding to the second maximum phase matches 0.5 Vdc. .
- FIG. 23 is a diagram showing first three-phase applied voltages Vu1 ', Vv1', and Vw1 'when the modulation factor is 100% in the sixth embodiment of the present invention.
- the horizontal axis represents the voltage phase ⁇ v [deg], and the vertical axis represents the ratio to the DC voltage Vdc.
- the first three-phase voltage commands Vu1, Vv1, and Vw1 are sinusoidal waveforms having an amplitude of Vdc / ⁇ 3 with reference to 0.
- the applied voltage of the phase corresponding to the first maximum phase is always the upper solid modulation of 0.5 Vdc, and the first minimum phase is set every 60 degrees.
- the applied voltage of the phase corresponding to is ⁇ 0.5 Vdc.
- the applied voltage of the phase corresponding to the first intermediate phase exceeds 0.4 Vdc in the vicinity of 30 deg, 150 deg, and 300 deg. Therefore, when the uppermost modulation is performed, the detection accuracy of the current detection value obtained at the first timing is deteriorated due to the influence of the switching noise of the first intermediate phase.
- the first timing and the second timing are performed twice for the first three-phase winding. It is a technical feature that current detection is performed over a wide range.
- FIG. 24 is a diagram showing second three-phase applied voltages Vu2 ′, Vv2 ′, and Vw2 ′ when the modulation factor is 100% in the sixth embodiment of the present invention.
- the horizontal axis represents the voltage phase ⁇ v [deg], and the vertical axis represents the ratio to the DC voltage Vdc.
- the second three-phase voltage commands Vu2, Vv2, and Vw2 are sinusoidal waveforms having an amplitude of Vdc / ⁇ 3 with reference to 0.
- the applied voltage of the phase corresponding to the second maximum phase is always the upper solid modulation of 0.5 Vdc, and the second minimum phase is set every 60 degrees.
- the applied voltage of the phase corresponding to is ⁇ 0.5 Vdc.
- the applied voltage of the phase corresponding to the second intermediate phase exceeds 0.4 Vdc in the vicinity of 30 deg, 150 deg, and 300 deg. Therefore, when the uppermost modulation is performed, the detection accuracy of the current detection value obtained at the first timing is deteriorated due to the influence of the switching noise of the first intermediate phase.
- the first timing and the second timing are performed twice for the second three-phase winding. It is a technical feature that current detection is performed over a wide range.
- the on / off signal generator 8 outputs on / off signals Qup1, Qun1, Qvp1, Qvn1, Qwp1, Qwn1 based on the first three-phase applied voltages Vu1 ′, Vv1 ′, Vw1 ′, and the second three-phase signals
- On / off signals Qup2, Qun2, Qvp2, Qvn2, Qwp2, and Qwn2 are output based on the applied voltages Vu2 ′, Vv2 ′, and Vw2 ′. Since it is the same as that described with reference to FIG. 4 in the first embodiment, detailed description thereof is omitted.
- the first current detector 11f is composed of a first current detecting resistance element 9a and a computing unit 10f.
- the arithmetic unit 10f has the first three-phase winding. Detect the current flowing through the wire.
- the computing unit 10f detects currents Iu11, Iv11, and Iw11 that flow through the first three-phase winding at the first timing ts1, and flows through the first three-phase winding at the second timing ts2. Currents Iu12, Iv12, and Iw12 are detected.
- the computing unit 10f calculates detection currents Iu1, Iv1, and Iw1 from Iu11, Iv11, Iw11, and Iu12, Iv12, and Iw12 obtained from the current flowing through the first current detection resistor element 9a.
- the undetectable two-phase current detection value may be estimated from the one-phase current detection value for the first three-phase winding. Further, as described in the third to fifth embodiments, the current detection value at the second timing may be corrected to be equivalent to the current detection value at the first timing.
- the second current detector 11g is composed of a second current detecting resistor element 9b and a computing unit 10g.
- the arithmetic unit 10g has the second three-phase winding. Detect the current flowing through the wire.
- the arithmetic unit 10g detects currents Iu21, Iv21, and Iw21 flowing through the second three-phase winding at the first timing ts1, and flows through the second three-phase winding at the second timing ts2. Currents Iu22, Iv22, and Iw22 are detected.
- the computing unit 10g calculates detection currents Iu2, Iv2, and Iw2 from Iu21, Iv21, Iw21, and Iu22, Iv22, and Iw22 obtained from the current flowing through the second current detection resistor element 9b.
- the undetectable two-phase current detection value may be estimated from the one-phase current detection value for the second three-phase winding. Further, as described in the third to fifth embodiments, the current detection value at the second timing may be corrected to be equivalent to the current detection value at the first timing.
- the first timing ts1 is when the first three-phase applied voltage and the second three-phase applied voltage are less than the first predetermined value or the maximum value of the carrier signal.
- the timing may be such that all the low potential side switching elements of the respective phases other than the first maximum phase and the second maximum phase are turned on. For example, it may be 2.4 ⁇ s after the center of one period of the carrier signal.
- the first three-phase applied voltage and the second three-phase applied voltage are shown to have exactly the same waveform. However, in the actual machine, the first three-phase applied voltage and the second three-phase applied voltage are different depending on the respective specifications of the first three-phase winding and circuit and the second three-phase winding and circuit. There is a difference in phase or amplitude.
- FIG. 25 is a diagram showing a waveform when a phase difference occurs between the first three-phase applied voltage and the second three-phase applied voltage in Embodiment 6 of the present invention.
- the electrical angle region that can be detected at the first timing ts1 is reduced compared to the case of FIG. 14 due to the influence of the phase difference of the three-phase applied voltage.
- the current detection value obtained by the first current detection resistor element 9a is not affected by the switching noise caused by the first three-phase applied voltage, but the switching noise caused by the second three-phase applied voltage is not affected. Will be affected. As a result, current detection accuracy deteriorates.
- the minimum value of the first intermediate phase and the second intermediate phase in the region where the second timing is used is Vmid_min, and ts2 may be set to satisfy the following expression (24).
- Vmid_min is 0.3 Vdc
- ts2 is 5 ⁇ s or more.
- the second timing may be set 5 ⁇ s or more before the center of one period of the carrier signal.
- the detected current is calculated based on the current value detected at the first timing when each phase applied voltage is less than the first predetermined value or the maximum value of the carrier signal, In other cases, the detection current is calculated based on the current value detected at the second timing.
- a current detection value that does not include switching noise can be obtained, and vibration and noise generated from an AC rotating machine having a plurality of n-phase windings (n is a natural number of 3 or more) having no phase difference can be reduced. It is possible to obtain a remarkable effect that has not been achieved in the past.
- Embodiment 7 FIG.
- the control device of the present invention is applied to an AC rotating machine having two sets of three-phase windings having no phase difference.
- the seventh embodiment a case will be described in which the control device of the present invention is applied to an AC rotating machine having a phase difference of 30 deg between two sets of three-phase windings. Therefore, the seventh embodiment is different from the previous sixth embodiment only in whether there is a phase difference.
- FIG. 26 is a diagram showing the first three-phase applied voltages Vu1 ′, Vv1 ′, Vw1 ′ and the second three-phase applied voltages Vu2 ′, Vv2 ′, Vw2 ′ in the seventh embodiment of the present invention.
- the detection current is calculated using the current detection value at the first timing
- the detection current is calculated using the current detection value at the second timing.
- a two-phase current detection value that cannot be detected may be estimated from the one-phase current detection value. Further, as described in the third to fifth embodiments, the current detection value at the second timing may be corrected to be equivalent to the current detection value at the first timing.
- the first timing ts1 is the same as when the number of winding sets is one, when the first three-phase applied voltage and the second three-phase applied voltage are less than the first predetermined value or the maximum value of the carrier wave signal.
- the timing may be such that all the low potential side switching elements of the respective phases other than the first maximum phase and the second maximum phase are turned on. For example, it may be 2.4 ⁇ s after the center of one period of the carrier signal.
- the second timing ts2 since it is desired to use the second timing ts2 when current detection is not possible at the first timing ts1, it may be set to a timing at which an effective current detection value can be obtained other than the above.
- the electrical angle region that can be detected at the first timing ts1 is greatly reduced compared to the case of FIG. 14 due to the influence of the phase difference of the three-phase applied voltage.
- the second timing is set to 2.6 ⁇ s before the center of one period of the carrier signal as shown in FIG. 14, the current detection value obtained by the first current detection resistor element 9a is the first 3 Since there is a sufficient phase difference without being affected by the switching noise due to the phase applied voltage, it is not affected by the switching noise due to the second three-phase applied voltage. For this reason, current detection accuracy does not deteriorate.
- the second timing may be set based on the same concept as that of one winding group. For example, the second timing may be set 2.6 ⁇ s before the center of one period of the carrier signal.
- the detected current is calculated based on the current value detected at the first timing when each phase applied voltage is less than the first predetermined value or the maximum value of the carrier signal, In other cases, the detection current is calculated based on the current value detected at the second timing.
- a current detection value that does not include switching noise can be obtained, and vibration and noise generated from an AC rotating machine having a plurality of n-phase (n is a natural number of 3 or more) windings having a phase difference can be reduced. It is possible to obtain a remarkable effect that has not been achieved in the past.
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Abstract
Description
特許文献1のモータ制御装置は、相電流検出時間tsを確保するために、最大相の低電位側スイッチング素子のオン時間が、相電流検出時間tsよりも小さい場合には、高電位側スイッチング素子をオン、低電位側スイッチング素子をオフとしたままとすることにより、最大相以外の2相の相電流値にスイッチングノイズが混入しないようにしている。
図1は、本発明の実施の形態1における電力変換装置の全体構成を示す図である。交流回転機1は、3相巻線U1、V1、W1が回転機の固定子に納められている3相交流回転機である。このような3相交流回転機としては、永久磁石同期回転機、誘導回転機、同期リラクタンス回転機等が挙げられるが、本発明は、3相巻線を有する交流回転機ならば、何れの回転機を用いてもよい。
図10は、本発明の実施の形態2における電力変換装置の全体構成を示す図である。先の実施の形態1における図1の構成と比較すると、本実施の形態2における図10の構成は、制御部5c、オフセット演算器7c、演算器10b、電流検出器11bが異なっている。そこで、これらの相違点を中心に、以下に説明する。なお、本実施の形態2でも、Tcを50μs、tiを5μsとした場合を例にして、説明する。
図17は、本発明の実施の形態3における電力変換装置の全体構成を示す図である。先の実施の形態2における図10の構成と比較すると、本実施の形態3における図17の構成は、演算器10c、電流検出器11cが異なっている。そこで、これらの相違点を中心に、以下に説明する。なお、本実施の形態3でも、Tcを50μs、tiを5μsとした場合を例にして、説明する。
ライン300:搬送波信号1周期の中央からTofs1(μs)前で検出した場合の電流検出値の波形
ライン301:搬送波信号1周期の中央で検出した場合の電流検出値の波形
ライン302:搬送波信号1周期の中央からTofs2(μs)後で検出した場合の電流検出値の波形
先の実施の形態3では、あらかじめ設定した補正係数を用いた補正式により、基準タイミングを合わせる補正を行い、検出電流を演算する場合について説明した。これに対して、本実施の形態4では、実機での状態に合わせた補正量をオンラインで算出して、補正する場合について説明する。
先の実施の形態4では、上式(13)の補正係数(1+Tofs1K)/(1-Tofs2K)あるいは係数Kを、オンラインで算出して、タイミング違いによる誤差を低減する場合について説明した。これに対して、本実施の形態5では、先の実施の形態4とは別の手法による補正を行う場合について説明する。
先の実施の形態1~5では、1組の巻線組を有する交流回転機に本発明の制御装置を適用した場合について説明した。しかしながら、本発明の制御装置は、複数組の巻線組を有する交流回転機に適用してもよい。そこで、本実施の形態6では、2組の位相差のない3相巻線を有する交流回転機に対して、本発明の制御装置を適用した場合について説明する。なお、本実施の形態6においても、先の実施の形態1~5と同様に、Tcを50μs、tiを5μs、第1の所定値を0.4Vdcとして説明する。
先の実施の形態6では、2組の位相差のない3相巻線を有する交流回転機に対して、本発明の制御装置を適用した場合について説明した。これに対して、本実施の形態7では、2組の3相巻線に30degの位相差がある交流回転機に対して、本発明の制御装置を適用する場合について説明する。したがって、本実施の形態7は、先の実施の形態6とは、位相差があるかないかのみが異なっている。
Claims (10)
- 直流電圧を出力する直流電源と、
mを自然数、nを3以上の自然数として、m組のn相巻線を有する交流回転機と、
前記m組の前記n相巻線のそれぞれの電流値を検出する電流検出器と、
高電位側スイッチング素子および低電位側スイッチング素子を有し、オン/オフ信号に基づいて前記高電位側スイッチング素子および前記低電位側スイッチング素子がスイッチング制御されることで、前記直流電圧を交流電圧に変換して前記巻線に印加する電力変換器と、
前記交流回転機の電流指令と前記電流検出器による電流検出値との差分に基づいて、電圧指令を演算するとともに、前記電圧指令に基づいて演算した印加電圧と搬送波信号とを比較することにより、前記電力変換器の前記高電位側スイッチング素子および前記低電位側スイッチング素子に前記オン/オフ信号を出力する制御部と
を有し、
前記電流検出器は、前記電力変換器の少なくとも(n-1)相分の低電位側スイッチング素子に直列に挿入された電流検出用抵抗素子に流れる電流に基づいて、前記n相巻線を流れる電流を検出する際に、前記搬送波信号の1周期において2回以上の固定タイミングで電流を検出し、スイッチングノイズにより生じる誤差を含まないような前記電流検出値を取得する
交流回転機の制御装置。 - 前記電力変換器は、前記m組のそれぞれにおいて、演算により求めた前記n相のそれぞれに対する前記印加電圧に関して、最大相の電圧指令に対応して演算される印加電圧が前記搬送波信号の最大値と等しくなるように、すべての電圧指令を等しくシフトすることで、前記印加電圧をオフセット補正する
請求項1に記載の交流回転機の制御装置。 - 前記交流回転機の角度を検出する角度検出器
をさらに有し、
前記電流検出器は、前記m組のそれぞれにおいて、今回の電流検出タイミングにおいて、n相のうちの少なくとも1相の電流を検出することができ、多くとも(n-1)相の電流が検出不可能な場合には、過去の電流検出タイミングにおいて取得したn相検出電流と、前記今回の電流検出タイミングにおいて検出することができた1相の検出電流と、前記角度検出器によって検出された前記過去の電流検出タイミングにおける前記交流電動機の角度と前記今回の電流検出タイミングにおける前記交流電動機の角度との差分である角度変化量と、に基づいて、前記今回の電流検出タイミングにおいて検出不可能であった前記(n-1)相の電流を演算により推定する
請求項1または2に記載の交流回転機の制御装置。 - 前記電流検出器は、前記m組のそれぞれにおいて、前記印加電圧の前記n相電圧のそれぞれが、第1の所定値未満または前記搬送波信号の最大値に等しい第1状態である場合には、少なくとも電圧指令が最大である相以外の各相の低電位側スイッチング素子が全てオンしている第1タイミングで得た前記電流検出値に基づいて検出電流を演算し、前記第1状態以外の場合には、前記第1タイミングとは異なるタイミングであって、少なくとも電圧指令が最小である相の低電位側スイッチング素子がオンしている第2タイミングで得た前記電流検出値に基づいて前記検出電流を演算する
請求項1から3のいずれか1項に記載の交流回転機の制御装置。 - 前記電流検出器は、
前記第1タイミングにおける電流検出値を、基準タイミングと前記第1タイミングとの時間差に比例する第1係数に基づいて、前記基準タイミングでの電流検出値に補正し、
前記第2タイミングにおける電流検出値を、基準タイミングと前記第2タイミングとの時間差に比例する第2係数に基づいて、前記基準タイミングでの電流検出値に補正する
請求項4に記載の交流回転機の制御装置。 - 前記電流検出器は、
前記第1タイミングおよび前記第2タイミングのいずれにおいても2相以上が検出可能な場合には、各相における電流検出値の比を用いて、前記第1係数と前記第2係数との比を算出し、前記第1タイミングにおける電流検出値または前記第2タイミングにおける電流検出値の少なくとも一方を前記基準タイミングにおける電流検出値に補正する
請求項5に記載の交流回転機の制御装置。 - 前記電流検出器は、検出可能相のうち最大振幅相の電流検出値の比を用いて、前記第1係数と前記第2係数との比を算出する
請求項6に記載の交流回転機の制御装置。 - 前記電流検出器は、
前記基準タイミングを前記第1タイミングとし、前記第1タイミングにおける電流検出値は補正せず、前記第2タイミングにおける電流検出値を前記第1タイミングにおける電流検出値に補正する
請求項5から7に記載の交流回転機の制御装置。 - 前記電流検出器は、
前記基準タイミングを前記第1タイミングとし、過去のタイミングにおいて、前記第1タイミングおよび前記第2タイミングのいずれにおいても電流検出値が検出可能であった相に関して、今回のタイミングにおいて前記第1タイミングで電流検出値が検出不可能であり、前記第2タイミングで電流検出値が検出可能である場合には、前記今回のタイミングにおける第2タイミングで電流検出値に対して、過去のタイミングにおける第1タイミングでの電流検出値から第2タイミングでの電流検出値を引いた値を加算することで、前記今回のタイミングにおける第2タイミングでの電流検出値を前記今回のタイミングにおける第1タイミングの電流検出値に補正する
請求項5に記載の交流回転機の制御装置。 - 請求項1から9のいずれか1項に記載の交流回転機の制御装置を備え、
前記制御部は、ステアリング系の操舵トルクを補助するトルクを、前記交流回転機が発生するように、前記電圧指令を演算する
電動パワーステアリングの制御装置。
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WO2021255813A1 (ja) | 2020-06-16 | 2021-12-23 | 三菱電機株式会社 | 電力変換装置、及び電動パワーステアリング装置 |
JP7471414B2 (ja) | 2020-06-16 | 2024-04-19 | 三菱電機株式会社 | 電力変換装置、及び電動パワーステアリング装置 |
WO2022130480A1 (ja) * | 2020-12-15 | 2022-06-23 | 三菱電機株式会社 | 電力変換装置 |
JP7504230B2 (ja) | 2020-12-15 | 2024-06-21 | 三菱電機株式会社 | 電力変換装置 |
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US20180022378A1 (en) | 2018-01-25 |
JP6266161B2 (ja) | 2018-01-24 |
JPWO2016143121A1 (ja) | 2017-06-08 |
EP3270502A4 (en) | 2018-11-21 |
EP3270502A1 (en) | 2018-01-17 |
EP3270502B1 (en) | 2023-10-11 |
CN107438943B (zh) | 2020-09-04 |
US10666169B2 (en) | 2020-05-26 |
CN107438943A (zh) | 2017-12-05 |
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