WO2011127754A1 - 双模压控振荡器、频率综合器及无线接收装置 - Google Patents

双模压控振荡器、频率综合器及无线接收装置 Download PDF

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Publication number
WO2011127754A1
WO2011127754A1 PCT/CN2011/000658 CN2011000658W WO2011127754A1 WO 2011127754 A1 WO2011127754 A1 WO 2011127754A1 CN 2011000658 W CN2011000658 W CN 2011000658W WO 2011127754 A1 WO2011127754 A1 WO 2011127754A1
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Prior art keywords
controlled oscillator
voltage controlled
frequency
unit
oscillator unit
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English (en)
French (fr)
Inventor
李巍
周谨
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Fudan University
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Fudan University
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Priority to US13/266,759 priority Critical patent/US8362809B2/en
Priority to CN201180001738.4A priority patent/CN102439845B/zh
Publication of WO2011127754A1 publication Critical patent/WO2011127754A1/zh
Anticipated expiration legal-status Critical
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B27/00—Generation of oscillations providing a plurality of outputs of the same frequency but differing in phase, other than merely two anti-phase outputs
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B19/00—Generation of oscillations by non-regenerative frequency multiplication or division of a signal from a separate source
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
    • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
    • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
    • H03B5/1206—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device using multiple transistors for amplification
    • H03B5/1212—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device using multiple transistors for amplification the amplifier comprising a pair of transistors, wherein an output terminal of each being connected to an input terminal of the other, e.g. a cross coupled pair
    • H03B5/1215—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device using multiple transistors for amplification the amplifier comprising a pair of transistors, wherein an output terminal of each being connected to an input terminal of the other, e.g. a cross coupled pair the current source or degeneration circuit being in common to both transistors of the pair, e.g. a cross-coupled long-tailed pair
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
    • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
    • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
    • H03B5/1228—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device the amplifier comprising one or more field effect transistors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
    • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
    • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
    • H03B5/1237—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator
    • H03B5/124—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator the means comprising a voltage dependent capacitance
    • H03B5/1243—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator the means comprising a voltage dependent capacitance the means comprising voltage variable capacitance diodes
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
    • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
    • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
    • H03B5/1237—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator
    • H03B5/1262—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator the means comprising switched elements
    • H03B5/1265—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator the means comprising switched elements switched capacitors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B2200/00—Indexing scheme relating to details of oscillators covered by H03B
    • H03B2200/006—Functional aspects of oscillators
    • H03B2200/0078—Functional aspects of oscillators generating or using signals in quadrature

Definitions

  • the invention belongs to the technical field of radio frequency wireless receiver integrated circuits, and relates to a dual-mode Voltage-Controlled Oscillator (DMVCO) capable of operating in a wideband mode or a quadrature mode, and A frequency synthesizer and a wireless receiving device of the DMVCO are used.
  • DMVCO Voltage-Controlled Oscillator
  • Figure 1 shows the spectral distribution of some communication standards in the current space environment.
  • mobile communication is one of the most important applications of wireless technology.
  • GSM Global System of Mobile Communication
  • the third generation mobile communication standards include the Chinese standard TD-SCDMA, the European standard WCDMA and the North American standard CDMA2000. Therefore, between the second generation mobile communication and the third generation mobile communication, between the different standards of the third generation mobile communication, wireless signals supporting mobile communication of different frequencies and different standards will coexist in space for a long time.
  • wireless data communication especially wireless broadband data communication technology, is also rapidly developing.
  • Bluetooth Bluetooth
  • Wi-Fi Wireless Fidelity
  • UWB Ultra Wideband
  • Wireless navigation applications are no exception, and global positioning systems such as GPS (Global Positioning System) / Galileo / Beidou are rapidly evolving. This is especially true for broadcast applications, such as DTMB (Digital Television Terrestrial Multimedia Broadcasting), CMMB (China Mobile Multimedia Broadcasting), DVB-H (Handheld Digital Multimedia Broadcasting) and other national and international standards, and Industry standards have matured, or are in the process of industrialization.
  • DTMB Digital Television Terrestrial Multimedia Broadcasting
  • CMMB China Mobile Multimedia Broadcasting
  • DVB-H Wideheld Digital Multimedia Broadcasting
  • Multi-Mode, multi-mode wireless receiver is especially important, which is in line with the important trend of the development of software-defined radio (SDR), which can greatly improve the weight of wireless receivers. Structure. Therefore, a multi-mode, reconfigurable radio receiver is tentatively proposed in the prior art.
  • the design of a wide-band range high-performance frequency synthesizer is one of the keys; further, the frequency synthesizer can be provided with a wide tuning range (Tunning Range, TR) orthogonality.
  • the Voltage-Controlled Oscillator (VC0) which is a local oscillator signal for modulation and demodulation of wireless signals, is the core of the technology.
  • multi-mode wireless receivers are usually integrated on a single circuit board. Such a simple stacking will result in an increase in power consumption, weight and area of the wireless receiver, and a significant increase in cost. Therefore, the development of wireless receivers in a multi-mode, single-chip direction is an inevitable trend.
  • SSBM single sideband mixer
  • PPFs polyphase filters
  • SSBM Single Sideband Mixer
  • the present invention proposes a DMVCO and frequency synthesizer that can operate in either a wideband mode or an orthogonal mode.
  • a dual mode voltage controlled oscillator comprising: a first voltage controlled oscillator unit and a second voltage controlled oscillator unit, wherein the first voltage controlled oscillator unit operates in a first frequency band, The second voltage controlled oscillator operates in a second frequency band, and the first frequency band and the second frequency band have an overlapping frequency band;
  • Each of the voltage controlled oscillator units includes a quadrature coupling module
  • the dual mode voltage controlled oscillator is operatively operable in a wideband mode or an orthogonal mode; wherein, when operating in the wideband mode, the corresponding frequency is generated only by the first voltage controlled oscillator unit or the second voltage controlled oscillator unit Differential signal
  • the first voltage controlled oscillator unit and the second voltage controlled oscillator unit simultaneously operate in the overlapping frequency band, by the first voltage controlled oscillator unit and the second voltage control
  • An orthogonal coupling module of the oscillator unit couples the first voltage controlled oscillator unit with the second voltage controlled oscillator unit and generates a quadrature signal of a corresponding frequency.
  • the first voltage controlled oscillator unit and the second voltage controlled oscillator unit have substantially the same frequency tuning characteristics at the overlap frequency band.
  • each of the voltage controlled oscillator units further includes:
  • a switched capacitor array for controlling the same frequency of the respective tuning curves of the first voltage controlled oscillator unit and the second voltage controlled oscillator unit in the overlapping frequency band;
  • a switched variable capacitor array for controlling the same tuning gains of respective tuning curves of the first voltage controlled oscillator unit and the second voltage controlled oscillator unit in the overlapping frequency band.
  • the switched capacitor array comprises a plurality of switched capacitor units, each switch capacitor unit correspondingly controlling a frequency of a one-tone tuning curve in the overlapping frequency band.
  • the switched variable capacitor array comprises a plurality of switched variable capacitor units, each of the switched variable capacitor units correspondingly controlling a tuning gain of a tuning curve in the overlapping frequency band.
  • the switched capacitor array further comprises a plurality of switched capacitor units using binary weights, correspondingly controlling the frequency of the tuning curve of the voltage controlled oscillator outside the overlapping frequency band.
  • each of the voltage controlled oscillator units further includes:
  • a fixed variable capacitance module having a linearization function for controlling the linearity of a tuning curve of the first voltage controlled oscillator unit and the second voltage controlled oscillator unit.
  • each of the voltage controlled oscillator units further includes:
  • phase shifting network module for providing a phase shift of substantially 90° to reduce frequency mismatch and amplitude mismatch of said first voltage controlled oscillator unit and said second voltage controlled oscillator unit in said overlapping frequency band Sensitivity.
  • a dual mode voltage controlled oscillator is provided, each of the voltage controlled oscillator units further comprising an on-chip inductor and a parasitic capacitor.
  • each of the voltage controlled oscillator units further includes:
  • a negative resistance module that provides the energy required to oscillate the resonant cavity of the voltage controlled oscillator unit
  • the negative resistance module includes two transistors that are cross-coupled with each other and a first current source array.
  • the current flowing through the cross-coupled transistor is adjusted by controlling the first current source array to control the first The output signal amplitude of the voltage controlled oscillator unit or the second voltage controlled oscillator unit.
  • a first switch is disposed between the first current source array and the cross-coupled transistor, and when the dual-mode voltage controlled oscillator operates in a wideband mode, the first switch is controlled to select the first Any one of a voltage controlled oscillator unit and a second voltage controlled oscillator unit.
  • the orthogonal coupling module includes two orthogonal coupling transistors, wherein a gate of each of the first voltage controlled oscillator unit/second voltage controlled oscillator unit The resonant cavity of the second voltage controlled oscillator unit / first voltage controlled oscillator unit The corresponding output signal is coupled.
  • a second current source array for providing a bias current to the quadrature coupling transistor is disposed in each of the voltage controlled oscillator units.
  • a second switch is disposed between the orthogonal coupling transistor and the second current source array, and the dual mode voltage controlled oscillator is implemented in the broadband mode and the orthogonal by controlling the switch Switch between modes.
  • the first frequency band is substantially 3 GHz to 4.8 GHz
  • the second frequency band is substantially 4 GHz to 6 GHz
  • the overlapping frequency band is substantially 4 GHz to 4.8 GHz.
  • a frequency synthesizer comprising a local oscillator generator, the local oscillator generator comprising:
  • Quadrature Single Sideband Mixer (QSSBM)
  • QSSBM Quadrature Single Sideband Mixer
  • the frequency synthesizer is operable to operate in a wideband mode or an orthogonal mode; when operating in the wideband mode, the differential signal generated by the first voltage controlled oscillator unit or the second voltage controlled oscillator unit passes the division Processing to generate a first orthogonal local oscillator signal; when operating in the orthogonal mode, inputting a quadrature signal obtained by orthogonal coupling processing and a quadrature signal output by the divider to the orthogonal
  • the single sideband mixer performs upconversion processing to generate a second quadrature local oscillator signal;
  • the first orthogonal local oscillator signal is used to cover a first output frequency band having a relatively low frequency
  • the second orthogonal local oscillation signal is used to cover a second output frequency band having a relatively high frequency
  • the local oscillator generator further includes a first stage frequency gate and a second stage frequency gate;
  • the first stage frequency gate selects a differential signal of a corresponding frequency of the output of the first voltage controlled oscillator unit or the second voltage controlled oscillator unit to the divider, and the second stage frequency gating The quadrature signal output by any of the dividers is selected to output the first quadrature local oscillator signal.
  • the first output frequency band may be substantially 0.4 GHz to 3 GHz
  • the second output frequency band may be substantially 5 GHz to 6 GHz.
  • a wireless receiving apparatus comprising any of the frequency synthesizers described above is provided.
  • the technical effect of the present invention is that the frequency synthesizer covers the lower output frequency band (for example
  • the frequency synthesizer and DMVCO operate in wideband mode, and DMVCO operates at a relatively low frequency, so the power consumption of the divider and buffer used by the frequency synthesizer can be greatly reduced;
  • the frequency synthesizer is When covering a higher output band (eg 5-6 GHz), the frequency synthesizer and DMVCO operate in quadrature mode, using two voltage-controlled oscillators operating in the overlap band in the DMVCO to provide the quadrature signals required by the QSSBM, The use of additional PPF and QVCO is avoided. Therefore, the frequency synthesizer structure using DMVCO is not complicated, not only the output frequency range can be guaranteed, but also the power consumption is not increased, and the output frequency range and power consumption requirements are also taken into consideration. . Moreover, the QSSBM used in the frequency synthesizer can suppress the image signal during mixing, improve the spectral purity of the voltage controlled oscillator, and have good frequency spuriousness.
  • Figure 1 is the spectrum distribution of some communication standards in the current space environment
  • FIG. 2 is a schematic structural diagram of a broadband orthogonal DMVCO according to an embodiment of the present invention
  • FIG. 3 is a schematic diagram showing a tuning curve distribution of the voltage controlled oscillators A and B shown in FIG. 2.
  • FIG. 4 is a block diagram showing the structure of a local oscillator generator of a frequency synthesizer according to an embodiment of the present invention, and the voltage controlled oscillator thereof is used. DMVCO as shown in Figure 2;
  • FIG. 5 is a schematic view showing the working principle of the DMVCO shown in Figure 2;
  • FIG. 6 is a schematic diagram of spectrum planning of the frequency synthesizer shown in FIG. 4;
  • FIG. 7 is a schematic structural view of a switched capacitor unit used in the switched capacitor array 120 of FIG. 2;
  • FIG. 8 is a schematic structural view of a switched variable capacitor unit used in the switched variable capacitor array 130 of FIG. 2;
  • FIG. 9 is a schematic diagram showing the structure of the fixed variable capacitance module 140 having the linearization function shown in FIG. 2;
  • FIG. 10 is a schematic structural diagram of the phase shift network module 150 shown in FIG. 2;
  • Figure 11 is a schematic illustration of the output signal of the DMVCO of the embodiment of Figure 2 in an orthogonal mode. 11 000658
  • a wideband quadrature dual-mode voltage-controlled oscillator refers to a DMVCO that can operate in a wideband mode or an orthogonal mode, respectively, where "wideband mode” refers to the division of one of the DMVCO's voltage-controlled oscillator output signals.
  • wideband mode refers to the division of one of the DMVCO's voltage-controlled oscillator output signals.
  • orthogonal mode refers to coupling two voltage controlled oscillators in a DMVCO together to produce a quadrature signal.
  • the following performance metrics must be considered: the frequency range (ie, the tuning range of the VCO, or the output frequency range of the frequency synthesizer), phase noise, frequency spurs, power dissipation, and chip area.
  • FIG. 2 is a schematic structural diagram of a broadband orthogonal DMVCO according to an embodiment of the present invention.
  • Figure 3 shows the tuning curve distribution of the voltage controlled oscillators A and B shown in Figure 2.
  • FIG. 4 is a block diagram showing the structure of a local oscillator generator of a frequency synthesizer according to an embodiment of the present invention. The voltage controlled oscillator uses DMVCO as shown in FIG. 2.
  • Figure 5 is a schematic diagram showing the operation of the DMVCO shown in Figure 2
  • Figure 6 is a schematic diagram of the spectrum planning of the frequency synthesizer shown in Figure 4.
  • the DMVCO 100 includes a voltage controlled oscillator A operating at a high frequency band and a voltage controlled oscillator B operating at a low frequency band.
  • the voltage controlled oscillators A and B have substantially the same circuit structure, and both of them include the same components, Their components will be described in detail.
  • 0.4-3GHZ and 5 are implemented by two VCOs in the DMVCO and the frequency gate, divider, and quadrature single sideband mixing in FIG.
  • the tuning range of -6 GHz (shown in Figure 4) can cover the frequency distribution of common communication standards as shown in Figure 1.
  • voltage controlled oscillators A and B include on-chip inductor 110a and
  • the voltage controlled oscillators A and B are respectively tuned in the frequency range in which they operate to produce the tone shown in Figure 3. Harmonic curve; Obviously, the voltage controlled oscillator A operates in the high frequency band (HB), and the voltage controlled oscillator operates in the low band (Low Band, LB).
  • HB refers to the frequency range of 4-6 GHz
  • LB refers to the frequency range of 3-4.8 GHz
  • the two are in the overlapping frequency range in the frequency range of 4-4.8 GHz.
  • VCO HB refers to voltage controlled oscillation.
  • A, VCO LB refers to the voltage controlled oscillator. It should be noted that, according to the target frequency band (frequency range) that the frequency synthesizer using the DMVCO needs to cover, the voltage controlled oscillators A and B can be set to operate differently from the above embodiments. Other frequency bands.
  • a negative resistance module 160 is connected across the LC cavity to provide the energy required to oscillate the resonant cavity of the voltage controlled oscillator A or B.
  • the negative resistance module 160 includes two transistors Mn that are cross-coupled with each other. Further, the negative resistance module 160 further includes a current source array 161, and the current flowing by the cross-coupled transistor Mn is controlled by the current source array 161, and crosses.
  • the gate voltage of the coupling transistor Mn is controlled by the gate bias V BIAS and provided by a low pass RC filter (such as R B and C B shown in Figure 2); the current flowing through the cross-coupled transistor Mn is controlled.
  • the size allows the voltage controlled oscillator A or B to always operate in the current-limited region without entering the voltage-constrained region, or without causing the oscillation amplitude to be too small due to the current being too small. Therefore, the performance and power consumption of the voltage controlled oscillator A/voltage controlled oscillator B can be better balanced.
  • a switch for example, VCOA_EN, VCOB_EN
  • VCOA_EN for example, VCOA_EN, VCOB_EN
  • voltage controlled oscillator A when DMVCO is operating in wideband mode, voltage controlled oscillator A produces a differential signal in the 4-6 GHz tuning range, and voltage controlled oscillator B is generated in the 3-4.8 GHz tuning range.
  • the difference signal, the differential signal output by any one of the voltage controlled oscillator A (ie, VCO HB) and the voltage controlled oscillator B (ie, VCO LB ) may be processed by a divider in the frequency synthesizer 10 to generate a corresponding frequency. Orthogonal signal.
  • the frequency strobe of the frequency synthesizer 10 is implemented by a multiplexer, and the first-order multiplexer (MUX2) 151 of the first stage selects one of the differential signals and then passes the multi-stage division.
  • the divider link module 17 formed by the divider performs a division process to generate quadrature signals in various frequency bands.
  • the five dividers (171 to 175) in the divider link module 17 are selected as Current Mode Logic (CML) divide by two dividers.
  • CML Current Mode Logic
  • a quadrature signal of 1-1.5 GHz can be generated; the differential signal generated by voltage controlled oscillator A is processed by dividers 172, 174 and 175 in divider link module 17 to produce 0.5- The orthogonal signal of 0.75 GHz; likewise, the differential signal generated by the voltage controlled oscillator B is processed by the divider 171 in the divider link module 17 to generate a quadrature signal of 1.5-2.4 GHz; the voltage controlled oscillator The differential signal generated by B is processed by the dividers 172 and 173 in the divider link module 17 to generate a 0.75-1.2 GHz quadrature signal; the differential signal generated by the voltage controlled oscillator B passes through the divider link module.
  • the dividers 172, 174, and 175 in 170 process to produce a quadrature signal of 0.375-0.6 GHz. Therefore, a quadrature signal of 0.4-3 GHz can be continuously generated, and a quadrature signal outputted by one of the dividers is selected by a second-stage three-select multiplexer (MUX3) 153, thereby generating a Covers the quadrature local oscillator signal in the band 0.4-3 GHz (for example, LO_I or LO_Q as shown in Figure 4).
  • MUX3 second-stage three-select multiplexer
  • the voltage controlled oscillator A and the voltage controlled oscillator B have overlapping frequency bands, there are corresponding overlapping frequency bands after the divider processing (for example, 1-1.5 GHz band and 0.75-1.2). In the GHz band, there is an overlap band at 1-1.2 GHz.
  • the quadrature signal processed by the voltage controlled oscillator B can be selected to cover the frequency band, so that the power consumption of the voltage controlled oscillator can be reduced, thereby reducing Power consumption of the frequency synthesizer.
  • the Frequency Synthesizer 10 when the DMVCO 100 is operating in the Width mode, the Frequency Synthesizer 10 also operates in the Wideband mode. With the MUX3 153's select output, the Frequency Synthesizer 10 can cover the output frequency range of 0.4-3 GHz. Therefore, when outputting the quadrature local oscillator signal in the frequency band of 0.4-3 GHz, only one of the voltage controlled oscillator A or the voltage controlled oscillator B can be operated, so that another voltage controlled oscillator can be stopped, greatly Reduce power consumption in wideband mode. Specifically, it can be realized by controlling a switch (for example, VCOA-EN, VCOB-EN) provided between the current source array and the cross-coupled transistor Mn.
  • a switch for example, VCOA-EN, VCOB-EN
  • DMVCO can also work in orthogonal mode.
  • the QVCO can be considered to be coupled by two identical VCOs, that is, the two VCOs have basically the same structure, operate in the same frequency range, and have the same frequency tuning characteristics, so that the two VCOs can be coupled. Produces precise quadrature output.
  • the DMVCO operates in the orthogonal mode to pass the frequency synthesizer 10 through the QSSBM (Quadature Single Sideband Mixer). 13 produces a quadrature signal of 5-6 GHz.
  • QSSBM Quadature Single Sideband Mixer
  • phase error of the quadrature signal input by QSSBM is small enough (the phase error is as small as possible)
  • voltage-controlled oscillation is required.
  • the frequency and slope of the tuning curves of the A and B in the overlapping frequency band are as consistent as possible, and the amplitudes of the output signals of the voltage controlled oscillators A and B in the overlapping frequency band need to be as uniform as possible, that is, the overall location.
  • the frequency tuning characteristics are as identical as possible in the overlapping bands.
  • a Orthogonal Coupling Module is set in each VCO of the DMVCO 100.
  • the voltage controlled oscillator A and the voltage controlled oscillator B operate in the overlapping frequency band (4-4.8 GHz), and the two are orthogonally coupled and output through the orthogonal coupling module.
  • DMVCO can be basically equivalent to QVCO.
  • the quadrature coupling modules of the voltage controlled oscillator A and the voltage controlled oscillator B each include two orthogonal coupling transistors Mc, the gate of each of the quadrature coupling transistors Mc in one voltage controlled oscillator and another voltage control
  • the corresponding output signal of the resonator of the oscillator is coupled; for example, as shown in Fig. 2, i+ is coupled with 1+, i_ is coupled with I-, q+ is coupled with Q+, and q- is coupled with Q-, so that voltage-controlled oscillation can be realized.
  • Orthogonal coupling between A and voltage controlled oscillator B Orthogonal coupling between A and voltage controlled oscillator B.
  • a voltage-controlled oscillator A and a voltage-controlled oscillator B are respectively provided with a switched capacitor array (Switched Capacitor Array) 120, and the switched capacitor array 120 is connected in parallel to the resonant cavity.
  • the voltage controlled oscillator A is divided into a 32-nine tuning curve at a tuning range of 4-6 GHz, and the voltage-controlled oscillator B is tuned at 3-4.8 GHz.
  • the range is divided into 32 tuning curves.
  • the switched capacitor array 120 is composed of a switched capacitor unit with binary weights (for example, five switched capacitor units generate 25 tuning curves).
  • the oscillation frequency of the voltage controlled oscillator is calculated by the following relation (1):
  • the frequency spacing between tuning curves in the overlapping frequency range is difficult to equal, making it difficult to guarantee two VCOs in the 4-4.8 GHz band.
  • the frequency tuning characteristics are equal in frequency parameters.
  • the capacitor is basically composed of 16 switched capacitor units and 4 switched capacitor units with binary weights.
  • 16 switched capacitor units control 16 tuning curves of the overlapping frequency band, and by design control of each switched capacitor unit in each switched capacitor array 120, two adjacent two adjacent overlapping bands can be guaranteed
  • the intervals of the tuning curves are the same, that is, it is easy to ensure that the output frequencies of the two voltage controlled oscillators are equal.
  • a 4-bit binary switched capacitor is used in addition to the overlap band to reduce parasitic effects.
  • FIG. 7 is a schematic structural view of a switched capacitor unit used in the switched capacitor array 120 of FIG.
  • the MIM capacitor ⁇ is controlled by an intermediate switch (consisting of two NMOS transistors), the control signal is Dn, and is formed by a pull-up tube (two PMOS transistors) and a pull-down tube (two NMOS transistors).
  • the bias level is provided.
  • the specific structure of the switched capacitor unit is not limited by the embodiment shown in FIG.
  • the tuning gains K vco of the two voltage-controlled oscillators must be ensured as much as possible, in Figure 3. Reflected as the "slope" of the corresponding tuning curve is the same.
  • the tuning gain of the voltage controlled oscillator A or B (ie, the slope of the tuning curve) is calculated by the following relation (2): Where K vc Uber is the slope of the corresponding tuning curve, f vc ., n the frequency corresponding to the tuning curve, C t . tn is the total capacitance of the cavity, C v is the variable capacitance, V etrl is the voltage controlled oscillator A or B's control voltage.
  • the traditional voltage-controlled oscillators use a fixed variable capacitor (without linearization function) physically fixed to the inductor-capacitor cavity to implement the voltage control function of the oscillator. Then, from the relation (2), it is not difficult to find out when the total capacitance of the cavity is C t .
  • n is small, that is, when the oscillation frequency of the voltage controlled oscillator is high, the slope of the tuning curve is large; when the total capacitance of the resonant cavity C t ⁇ is large, that is, when the oscillation frequency of the voltage controlled oscillator is low, The slope of the tuning curve is small. In order to offset the influence of the change of C tot .
  • a voltage-controlled oscillator A and a voltage-controlled oscillator B are provided with a switched variable capacitor array (Switched Varactor Array) 130, a switched variable capacitor.
  • the array 130 is connected in parallel to the resonant cavity.
  • FIG. 8 is a block diagram showing the structure of a switched variable capacitor unit used in the switched variable capacitor array 130 of FIG.
  • the switched variable capacitor array 130 includes 16 (since n in the relationship (2) is any integer from 1 to 16), such as
  • the switched variable capacitance unit shown in FIG. 8 performs slope adjustment on 16 tuning curves of the overlapping frequency bands, respectively.
  • Cv variable capacitance
  • C t ⁇ the switch through the switched variable capacitance unit 130 is The number of variable capacitors (Cv) connected to the Vctrl terminal is reduced to control the slope of the tuning curve.
  • the tuning gain K vco can be controlled by the switched variable capacitor array 130 so that the frequency tuning characteristics of the voltage controlled oscillators A and B in the overlapping frequency band are uniformly designed as a fixed K vco . Moreover, the frequency tuning characteristics of voltage controlled oscillators A and B can be fine tuned by the frequency band to compensate for fluctuations in process, voltage, temperature (PVT).
  • each switched variable capacitor unit in the switched variable capacitor array 130 is not limited by the embodiment shown in FIG. 8 of the present invention.
  • each switched variable capacitor unit can also be composed of two inverted MOS tube (I-MOS) variable capacitors, which can be fine-tuned in the frequency band by switching, or can be connected to the oscillator tuning voltage V ctrl To control the tuning gain of the VCO.
  • I-MOS inverted MOS tube
  • the bias current of the quadrature coupling transistor Mc is also provided by the current source array 170 as shown, so that the voltage controlled oscillators A and B can be easily ensured in the overlapping frequency band.
  • the amplitude of the oscillation is the same, which further makes the frequency tuning characteristics of the two overlapping bands more similar.
  • a switch (IQ_EN) can be placed between the quadrature coupling transistor M c and the current source array 170. By controlling the switch, the DMVCO can be switched between the wideband mode and the quadrature mode.
  • the switched capacitor array 120 and the switched variable capacitor array 130 are both digitally variable capacitor arrays, and it is difficult to implement each tuning shown in FIG. Continuous tuning control of the curve. Therefore, a fixed variable capacitance module 140 having a linearization function is also provided on the voltage controlled oscillators A and B.
  • FIG. 9 is a schematic structural view of the fixed variable capacitance module 140 having the linearization function shown in FIG. 2. As shown in FIG. 2 and FIG. 9, the fixed variable capacitance module 140 having the linearization function is also connected in parallel to the resonant cavity. In the fixed variable capacitance module 140, the voltage division network formed by the resistor R2 is used. The set variable capacitor C VAR is biased at three different bias voltages (V DD , V DD /2 and V ss ), respectively. The variable capacitor C VAR and the bias point can pass between the large resistor R1. Isolation. By being thus biased, the linearity of the respective tuning curves of the voltage controlled oscillators A and B can be greatly improved.
  • variable capacitance array 130 and the fixed variable capacitance module 140 with linearization function can realize the same frequency tuning characteristics of the voltage controlled oscillators A and B in the overlapping frequency band, that is, the tuning curve shown in FIG. 3 is in the overlapping frequency band. Closer; however, voltage-controlled oscillators A and B are very sensitive to frequency mismatch and amplitude mismatch when implementing QVCO in overlapping bands. To reduce sensitivity to these mismatches, voltage control The phase shift network module 150 is also disposed on the oscillators A and B.
  • FIG. 10 is a schematic structural diagram of the phase shift network module 150 shown in FIG.
  • the large signal equivalent transconductance of the quadrature coupling transistor Mc can be calculated by the following relation (3): Where G m — Mc is the equivalent transconductance of the quadrature coupling transistor Mc, g m — Me is the transconductance of the quadrature coupling transistor Mc, and 1 is the phase shift resistance in the phase shift network module 150, ( ⁇ is phase The phase shift capacitance in the network module 150 is shifted.
  • phase shift As can be seen from relation (3), adjusting the resistance R s and / or the capacitance C s to an appropriate value provides a close to 90. Phase shift.
  • the phase shift provided greatly reduces the sensitivity of the quadrature mode DMVCO to frequency and amplitude mismatch, so that the orthogonal mode DMVCO achieves better phase noise performance and a more accurate quadrature output signal.
  • FIG. 11 is a diagram showing the output signal of the DMVCO of the embodiment shown in Figure 2 in the orthogonal mode. Among them, the orthogonal signals generated by DMVCO in the orthogonal mode (two paths of I and Q, respectively) are schematically shown in Fig. 11.
  • a QSSBM 13 is provided, and when the frequency synthesizer 10 operates in the orthogonal mode, the quadrature signal output from the DMVCO 100 and the quadrature signal output from the divider 173 are input to the QSSBM 13, QSSBM 13
  • the up-conversion process can output a quadrature local oscillator signal of 5-6 GHz. Therefore, as shown in Fig. 6, the frequency synthesizer 10 can cover the frequency band of 5-6 GHz through the mixer (MIX). It will be appreciated that in other embodiments, the quadrature signals output by other dividers in the divider chain may also be input to the QSSBM 13.
  • DMVCO tuning range is relatively low working frequency band, the frequency synthesizer
  • the power consumption of the divider and buffer used can be large Large reduction;
  • the frequency synthesizer covers a higher output band eg 5-6 GHz
  • the frequency synthesizer and DMVCO operate in quadrature mode, using two voltage-controlled oscillators in the DM VCO that operate in overlapping bands.
  • the quadrature signal required by QSSBM avoids the use of additional PPF and QVCO.
  • the structure of the frequency synthesizer using DMVCO is not complicated, not only the output frequency range can be guaranteed, but also the power consumption is not increased.
  • the QS SBM used in the frequency synthesizer can suppress the image signal during mixing, improve the spectral purity of the voltage controlled oscillator, and have good frequency spur.
  • the frequency synthesizer covering the 0.4-3 GHz and 5-6 GHz frequency bands and the DMVCO used by the frequency synthesizer are described.
  • Those skilled in the art can define the radio through software according to the above teachings and enlightenment. Technology to achieve coverage of other bandwidth ranges. Therefore, the DMVCO and the frequency synthesizer of the invention have good reconfigurability.
  • the local oscillator generator generally further includes a phase locked loop (PLL) module (not shown), and the phase locked loop module can be used to control and stabilize the output of the DMVCO. frequency.
  • PLL phase locked loop
  • a radio receiving device e.g., a radio receiver
  • other specific components included in the radio receiving device will not be described in detail herein.

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Description

双模压控振荡器、 频率综合器及无线接收装置 技术领域
本发明属于射频无线接收机集成电路技术领域， 涉及可工作于宽 带模式（ wideband mode )或正交模式 ( quadrature mode ) 的双模压控 振荡器（ Dual-Mode Voltage-Controlled Oscillator, DMVCO ) 、 以及使 用该 DMVCO的频率综合器和无线接收装置。 背景技术
随着无线通信技术的快速发展， 越来越多的无线通信设备进入到 人们的日常生活当中。 在这种背景下， 为了追求更快的数据传输率以 及更高的频谱利用率， 越来越多的无线通信标准出现在现在的空间环 境中。
图 1所示为目前空间环境中的一些通信标准的频谱分布。 例如， 移动通信是无线技术的最重要应用之一， 在目前广泛使用的第二代移 动通信 GSM ( Global System of Mobile Communication, 全球移动通信 系统）制式移动通信的基础上， 移动通信正在全面进入第三代移动通 信的时代。第三代移动通信标准包括中国标准 TD-SCDMA,欧洲标准 WCDMA以及北美标准 CDMA2000。 于是， 第二代移动通信与第三代 移动通信之间， 第三代移动通信不同标准之间， 不同频率、 不同制式 的支持移动通信的无线信号将在空间长期共存。 同时， 随着人们对数 据高速无线传输方面的要求不断提高， 无线数据通信特别是无线宽带 数据通信技术也快速发展， 目前正广泛使用的通信标准有 Bluetooth (蓝牙） 和 Wi-Fi, 还有将来带宽更宽、 速度更快的通信标准 UWB ( Ultra Wideband, 超带宽） 等等。 无线导航应用也不例外， 全球定 位系统 GPS ( Global Positioning System, 全球定位系统） /伽利略 /北斗 等定位导航系统都在快速发展。 广播应用更是如此， DTMB ( Digital Television Terrestrial Multimedia Broadcasting,地面数字电视 多媒体 广播） 、 CMMB ( China Mobile Multimedia Broadcasting , 中国移动多 媒体广播） 、 DVB-H (手持数字多媒体广播）等国内和国际标准， 以 及行业标准都已经趋于成熟， 或正在产业化推进过程中。
在目前这种无线通信技术的发展趋势下， 一款能够支持多标准多
确 认 本 模式（即 Multi-Mode, 多模）的无线接收机就显得尤其重要， 这符合 软件定义无线电 （ Software-Defined Radio, SDR ) 这一无线电技术发 展的重要趋势， 可以大大提高无线接收机的可重构性。 因此， 现有技 术中尝试性提出了支持多模式、 可重构的无线电接收机。 然而， 为实 现这种无线电接收机， 一款宽频带范围的高性能频率综合器的设计是 关键之一；进一步，可以为该频率综合器提供一个宽调谐范围（Tuning Range, TR ) 的正交本振信号以进行无线信号的调制与解调的压控振 荡器 （ Voltage-Controlled Oscillator, VC0 )是技术的核心。
现在大多数的商用多模通信设备中， 支持多模式的无线接收机通 常是将各个通信模式的芯片分别集成在一块电路板上。 这样的简单堆 砌， 将造成无线接收机的功耗、 重量和面积增加， 成本也大大增加。 因此， 无线接收机向多模、 单芯片式方向发展是必然趋势。
M. Ingels等人在 ESSCIRC 2007 (第 436-439页）发表的、题为 "A CMOS 100 MHz to 6 GHz Software Defined Radio Aanalog Front-end with Integrated Pre-Power Amplifier" 的文章中， 提出了一种宽频带的 频率综合器。该频率综合器使用单边带混频器（ Single S ideband Mixer, SSBM ) 进行上变频操作来扩展频带， 并使用多个多相滤波器 （Poly Phase Filter, PPF ) 为单边带混频器 ( Single Sideband Mixer, SSBM ) 提供正交信号，然而为提高 PPF的工作频率范围， PPF的阶数必须增加， 势必会大大增加该频率综合器的功耗。
Davide Guermandi等人在 2005年 2月的 IEEE ISSCC Digest of Technical Papers上 (第 536-537页 )发表的、 题为 "A 0.75 to 2.2GHz Continuously-Tunable Quadrature VCO" 文章中， 提出了一种宽频带的 频率综合器。 该频率综合器同样使用 SSBM进行上变频操作来扩展频 带， 但通过使用正交压控振荡器 （ Quadrature Voltage-Controlled Oscillator, QVCO ) 为 SSBM提供正交信号。 但是 QVCO的调谐范围非 常有限， 并且功耗大、 谐波抑制能力差。
Pierluigi Nuzzo等人在 2008年 6月 "IEEE Radio Frequency Integrated Circuits Symp."论文集中 （第 303-306页） 的、 题为 "A 0.1 -5GHz Dual-VCO Software-Defined ΣΔ Frequency Synthesizer in 45nm Digital CMOS"文章中， 提出了一种宽频带的频率综合器。 在该 频率综合器中， 为了实现压控振荡器的宽调谐范围， 采用了两个分别 工作在高低两个不同频段的压控振荡器来覆盖一个非常宽的调谐范 围。 但是， 这样的一种架构需要压控振荡器工作在所需频率的两倍频 率处， 然后通过除二除法器（ Divider-by-2 )产生镜像来抑制接收机所 需要的正交信号。 由于高频除法器（有时还包括高频緩冲器）的使用， 这种架构的频率综合器在高频段需要消耗大量的功耗， 频率综合器的 带宽和功耗难以兼顾。 发明内容
为克服以上所述及的缺点或者其它缺点， 本发明提出一种既可工 作于宽带模式也可工作于正交模式的 DMVCO和频率综合器。
按照本发明的一方面， 提供一种双模压控振荡器， 其包括第一压 控振荡器单元和第二压控振荡器单元， 所述第一压控振荡器单元工作 于第一频段， 所述第二压控振荡器工作于第二频段， 所述第一频段与 所述第二频段之间具有交叠频段；
每个所述压控振荡器单元均包括正交耦合模块；
所述双模压控振荡器可操作地工作于宽带模式或正交模式； 其中， 工作于所述宽带模式时， 仅通过第一压控振荡器单元或第 二压控振荡器单元产生相应频率的差分信号；
工作于所述正交模式时， 所述第一压控振荡器单元和第二压控振 荡器单元同时工作在所述交叠频段， 通过所述第一压控振荡器单元和 第二压控振荡器单元的正交耦合模块， 将所述第一压控振荡器单元与 所述第二压控振荡器单元耦合在一起， 并产生相应频率的正交信号。
按照本发明提供的双模压控振荡器的优选实施例中， 所述第一压 控振荡器单元和第二压控振荡器单元在所述交叠频段的频率调谐特 性基本相同。
较佳地， 每个所述压控振荡器单元还包括：
开关式电容阵列， 其用于控制所述第一压控振荡器单元和所 述第二压控振荡器单元在所述交叠频段内的相应调谐曲线的所 在频率相同； 以及
开关式可变电容阵列， 其用于控制所述第一压控振荡器单元 和所述第二压控振荡器单元在所述交叠频段内的相应调谐曲线 的调谐增益相同。 8
较佳地， 所述开关式电容阵列包括多个开关式电容单元， 每个开 关式电容单元对应控制所述交叠频段内的一奈调谐曲线的所在频率。
较佳地， 所述开关式可变电容阵列包括多个开关式可变电容单 元， 每个开关式可变电容单元对应控制所述交叠频段内的一条调谐曲 线的调谐增益。
较佳地， 所述开关式电容阵列还包括多个采用二进制权重的开关 式电容单元， 对应控制所述压控振荡器在所述交叠频段外的调谐曲线 的所在频率。
在又一较佳实例中， 每个所述压控振荡器单元还包括：
具有线性化功能的固定式可变电容模块， 其用于控制所述第 一压控振荡器单元和所述第二压控振荡器单元的调谐曲线的线 性度。
在还一较佳实例中， 每个所述压控振荡器单元还包括：
相移网络模块， 其用于提供基本为 90°的相移以降低所述第 一压控振荡器单元和第二压控振荡器单元在所述交叠频段对频 率失配及幅度失配的敏感性。
按照本发明提供双模压控振荡器的又一实施例， 每个所述压控振 荡器单元还包括片上电感和寄生电容。
较佳地， 每个所述压控振荡器单元还包括：
负阻模块， 其为所述压控振荡器单元的谐振腔提供振荡所需 要的能量；
其中， 所述负阻模块包括两个相互交叉耦合的晶体管以及第一电 流源阵列； 通过控制所述第一电流源阵列来调整流经所述交叉耦合晶 体管的电流大小， 以控制所述第一压控振荡器单元或第二压控振荡器 单元的输出信号幅度。
较佳地， 所述第一电流源阵列与所述交叉耦合的晶体管之间设置 第一开关， 所述双模压控振荡器工作于宽带模式时， 控制所述第一开 关以选择所述第一压控振荡器单元和第二压控振荡器单元中的任意 一个。
较佳地， 所述正交耦合模块包括两个正交耦合晶体管， 其中， 第 一压控振荡器单元 /第二压控振荡器单元中的每个所述正交耦合晶体 管的栅极与第二压控振荡器单元 /第一压控振荡器单元的所述谐振腔 的相应输出信号耦合。
较佳地， 每个所述压控振荡器单元中设置用于为所述正交耦合晶 体管提供偏置电流的第二电流源阵列。
较佳地， 所述正交耦合晶体管与所述第二电流源阵列之间设置第 二开关， 通过控制所述开关以实现所述双模压控振荡器在所述宽带模 式和所述正交模式之间切换。
具体地， 所述第一频段基本为 3GHz至 4.8GHz, 所述第二频段基 本为 4GHz至 6GHz, 所述交叠频段基本为 4GHz至 4.8GHz。
按照本发明的又一方面，提供一种频率综合器， 包括本振发生器， 所述本振发生器包括：
以上所迷及的任意一种双模压控振荡器，
正交单边带混频器 ( Quadrature Single Sideband Mixer , QSSBM ) , 以及
一级或一级以上的除法器；
其中， 所述频率综合器可操作地工作于宽带模式或正交模式； 工作于所述宽带模式时， 第一压控振荡器单元或第二压控振荡器 单元所产生差分信号通过所述除法器处理以产生第一正交本振信号； 工作于所述正交模式时， 将正交耦合处理后输出得到的正交信号 和所述除法器所输出的正交信号输入至所述正交单边带混频器进行 上变频处理以产生第二正交本振信号；
所迷第一正交本振信号用于覆盖频率相对低的第一输出频段， 所 述第二正交本振信号用于覆盖频率相对高的第二输出频段。
按照本发明提供的频率综合器的一实施例， 所述本振发生器还包 括第一级频率选通器和第二级频率选通器；
其中， 所述第一级频率选通器选择第一压控振荡器单元或第二压 控振荡器单元所述输出的相应频率的差分信号至所述除法器， 所述第 二级频率选通器选择任一所述除法器所输出的正交信号以输出所述 第一正交本振信号。
较佳地， 所述第一输出频段可以基本为 0.4GHz至 3GHz， 所述第 二输出频段可以基本为 5 GHz至 6GHz。
按照本发明的再一方面， 提供一种无线接收装置， 其包括以上所 述及的任意一种频率综合器。 本发明的技术效果是， 频率综合器在覆盖较低输出频段 （例如
0.4-3GHZ ) 时， 频率综合器和 DMVCO工作于宽带模式， DMVCO工 作在相对较低的频率上， 因此频率综合器所使用的除法器和緩沖器的 功耗就能大大降低； 频率综合器在覆盖较高输出频段（例如 5-6GHz ) 时， 频率综合器和 DMVCO工作于正交模式， 利用 DMVCO中的两个 工作于交叠频段的压控振荡器来提供 QSSBM所需要的正交信号， 避 免了额外的 PPF和 QVCO的使用， 因此， 使用 DMVCO的频率综合 器结构并不复杂， 不但其输出频率范围可得到保证， 而且功耗并不增 加， 同时兼顾了输出频率范围和功耗的要求。 并且， 频率综合器中使 用的 QSSBM可以抑制混频时的镜像信号， 提高压控振荡器的频谱纯 度， 频率杂散性好。 附图说明
从结合附图的以下详细说明中， 将会使本发明的上述和其它目的 及优点更加完全清楚，其中，相同或相似的要素采用相同的标号表示。
图 1是目前空间环境中的一些通信标准的频谱分布；
图 2是按照本发明一实施例提供的宽带正交 DMVCO的结构示意 图；
图 3是图 2所示压控振荡器 A和 B的调谐曲线分布示意图； 图 4是按照本发明一实施例提供的频率综合器的本振发生器的模 块结构示意图， 其压控振荡器使用如图 2所示的 DMVCO;
图 5是图 2所示的 DMVCO的工作原理示意图；
图 6是图 4所示频率综合器的频谱规划示意图；
图 7是图 2所示开关式电容阵列 120中所使用的开关式电容单元 的结构示意图；
图 8是图 2所示开关式可变电容阵列 130中所使用的开关式可变 电容单元的结构示意图；
图 9是图 2所示具有线性化功能的固定式可变电容模块 140的结 构示意图；
图 10是图 2所示的相移网络模块 150的结构示意图；
图 11是图 2所示实施例的 DMVCO在正交模式下输出信号的示 意图。 11 000658
具体实施方式
下面介绍的是本发明的多个可能实施例中的一些， 旨在提供对本 发明的基本了解。 并不旨在确认本发明的关键或决定性的要素或限定 所要保护的范围。 容易理解， 根据本发明的技术方案， 在不变更本发 明的实质精神下， 本领域的一般技术人员可以提出可相互替换的其它 实现方式。 因此， 以下具体实施方式以及附图仅是对本发明的技术方 案的示例性说明， 而不应当视为本发明的全部或者视为对本发明技术 方案的限定或限制。
本文中， 宽带正交双模压控振荡器是指可分别地工作于宽带模式 或正交模式的 DMVCO, 其中， "宽带模式" 是指对 DMVCO的其中 一个压控振荡器输出信号进行除法处理以产生正交信号； "正交模 式，，是指将 DMVCO中的两个压控振荡器耦合在一起以产生正交信号。
在设计 VCO及频率综合器时， 必须兼顾考虑以下性能指标： 频 率范围 （即 VCO的调谐范围、 或频率综合器的输出频率范围） 、 相 位噪声、 频率杂散、 功耗以及芯片面积等。
图 2所述为按照本发明一实施例提供的宽带正交 DMVCO的结构 示意图。图 3所示为图 2所示的压控振荡器 A和 B的调谐曲线分布示 意图。 图 4所示为按照本发明一实施例提供的频率综合器的本振发生 器的模块结构示意图， 其压控振荡器使用如图 2所示的 DMVCO。 图 5所示为图 2所示的 DMVCO的工作原理示意图， 图 6所示为图 4所 示频率综合器的频谱规划示意图。
参阅图 2所示， DMVCO 100包括工作在高频段的压控振荡器 A 和工作低频段的压控振荡器 B，压控振荡器 A和 B的电路结构基本相 同， 它们都包括相同部件， 以下将详细描述它们的各部件。 在该实施 例中 , 为支持各种通信标准协议， 通过 DMVCO中的两个 VCO以及 图 2中的频率选通器、除法器和正交单边带混频的处理，实现 0.4-3GHZ 以及 5-6GHZ的调谐范围 （如图 4所示） ， 从而可以覆盖如图 1所示 的常见通信标准的频谙分布。
参阅图 2和图 3， 压控振荡器 A和 B分别包括片上电感 110a和
1 10b, 同时还包括在片上电感两端的寄生电容（图中未示出） 。 压控 振荡器 A和 B分别在其所工作的频段范围内调谐产生如图 3所示的调 谐曲线； 明显地， 压控振荡器 A工作于高频段（High Band, HB ) , 压控振荡器 Β工作于低频段（Low Band, LB )。 在下文中， HB是指 4-6GHz的频率范围， LB是指 3-4.8GHZ的频率范围，二者在 4-4.8GHz 频率范围为交叠频段， 在附图中， VCO HB是指压控振荡器 A, VCO LB是指压控振荡器 ^ 需要说明是， 根据使用该 DMVCO的频率综 合器需要覆盖的目标频段（频率范围）， 可以设置压控振荡器 A和 B 工作于不同于以上实施例的其它频段。
继续如图 2所示， 在 LC谐振腔的两端并联负阻模块 160, 其用 来为压控振荡器 A或 B的谐振腔提供振荡所需要的能量。在该实施例 中， 负阻模块 160包括两个相互交叉耦合的晶体管 Mn; 进一步， 负 阻模块 160还设置电流源阵列 161， 交叉耦合晶体管 Mn所流过的电 流受电流源阵列 161控制， 交叉耦合晶体管 Mn的栅极电压通过栅极 偏压 VBIAS控制并通过低通 RC滤波器来提供（如图 2中所示的 RB和 CB ) ； 控制交叉耦合的晶体管 Mn所流过的电流大小使压控振荡器 A 或 B始终工作在电流受限区， 而不至于进入电压受限区、 或者不至于 由于电流太小而导致振荡幅度太小。 因此， 可以更好地平衡压控振荡 器 A/压控振荡器 B的性能和功耗。另外，在电流源阵列 161和交叉耦 合的晶体管 Mn之间设置开关 （例如， VCOA— EN, VCOB— EN ) ， 通过控 制该开关， 可控制相应的压控振荡器是否工作。
参阅图 2和图 4、 图 5和图 6， DMVCO工作于宽带模式时， 压控 振荡器 A 在 4-6GHz 调谐范围内产生差分信号， 压控振荡器 B 在 3-4.8GHZ调谐范围内产生差分信号， 压控振荡器 A (即 VCO HB )和 压控振荡器 B (即 VCO LB ) 中任意一个所述输出的差分信号可以被 频率综合器 10 中的除法器进行处理以产生相应频率的正交信号。 在 该实施例中， 频率综合器 10 的频率选通器通过多路复用器实现， 第 一级的二选一多路复用器 （MUX2 ) 151 选择其中一路差分信号， 然 后通过多级除法器所形成的除法器链路模块 17 进行除法处理， 以生 成各个频段范围的正交信号。 在该实施例中， 除法器链路模块 17 中 的五个除法器（ 171至 175 )选择为电流模式逻辑（Current Mode Logic, CML )除 2除法器。 例如， 压控振荡器 A所产生的差分信号经过除法 器链路模块 17中的除法器 171处理后， 可产生 2-3GHz的正交信号； 压控振荡器 A所产生的差分信号经过除法器链路模块 17中的除法器 172和 173处理后，可产生 1-1.5GHZ的正交信号；压控振荡器 A所产 生的差分信号经过除法器链路模块 17中的除法器 172、 174和 175处 理后， 可产生 0.5-0.75GHZ的正交信号； 同样地， 压控振荡器 B所产 生的差分信号经过除法器链路模块 17中的除法器 171处理后， 可产 生 1.5-2.4GHZ的正交信号；压控振荡器 B所产生的差分信号经过除法 器链路模块 17中的除法器 172和 173处理后，可产生 0.75- 1.2GHz的 正交信号； 压控振荡器 B所产生的差分信号经过除法器链路模块 170 中的除法器 172、 174和 175处理后， 可产生 0.375-0.6GHz的正交信 号。 因此， 可以连续产生 0.4-3GHZ 的正交信号， 通过第二级的三选 一多路复用器 （MUX3 ) 153 选择其中一个除法器输出的正交信号， 从而可以生成如图 6所示的覆盖频段 0.4-3GHz的正交本振信号 （例 如， 如图 4所示的 LO_I或 LO_Q )。 需要说明的是， 在该实例中， 由 于压控振荡器 A和压控振荡器 B存在交叠频段，在除法器处理以后也 相应存在交叠频段 （例如， 1-1.5GHZ 频段与 0.75-1.2GHz 频段在 1-1.2GHZ存在交叠频段），此时可以选择使用压控振荡器 B除法处理 后的正交信号来覆盖该频段， 这样， 可以减小压控振荡器的功耗， 进 而降低频率综合器的功耗。
因此， DMVCO 100工作在宽度模式时， 频率综合器 10也工作在 宽带模式， 通过 MUX3 153 的选择输出， 频率综合器 10 可以覆盖 0.4-3GHZ的输出频率范围。 因此， 在输出 0.4-3GHZ频段内的正交本 振信号时，仅通过压控振荡器 A或者压控振荡器 B中的一个工作即可， 因此， 另一压控振荡器可以停止工作， 大大降低宽带模式下的功耗。 具体地， 可以通过控制电流源阵列和交叉耦合的晶体管 Mn之间设置 的开关 （例如， VCOA— EN, VCOB—EN ) 来实现。
另外， DMVCO还可以工作于正交模式。 现有技术中， QVCO可 以认为由两个完全相同的 VCO耦合而成， 即两个 VCO的结构基本相 同、 工作于相同频段范围并具有相同的频率调谐特性， 因此， 两个 VCO之间可以耦合产生精准的正交输出。在本发明中， 压控振荡器 A 和 B均工作于 4-4.8GHZ频段时， DMVCO工作于正交模式以使频率 综合器 10通过 QSSBM ( Quadrature Single Sideband Mixer, 正交单边 带混频器） 13产生 5-6GHZ的正交信号。 为了保证 QSSBM所输入的 正交信号的相位误差足够小 （相位误差越小越好） ， 需要使压控振荡 器 A和 B在交叠频段内的调谐曲线的所在频率和斜率尽可能一致，并 且需要使压控振荡器 A和 B在交叠频段内的输出信号的幅度尽可能一 致， 即总体来所， 在交叠频段内频率调谐特性尽可能相同。
为满足正交模式工作的要求， DMVCO 100的每个 VCO中设置正 交耦合模块。 如图 2所示， 在该实施例中， 压控振荡器 A和压控振荡 器 B工作于交叠频段时（4-4.8GHZ ) , 二者通过正交耦合模块进行正 交耦合输出， 此时 DMVCO可以基本等同于 QVCO。 具体地， 压控振 荡器 A和压控振荡器 B的正交耦合模块均包括两个正交耦合晶体管 Mc, 一个压控振荡器中每个正交耦合晶体管 Mc的栅极与另一个压控 振荡器的谐振腔的相应输出信号耦合； 例如， 如图 2所示， i+与 1+ 耦合， i_与 I-耦合， q+与 Q+耦合， q-与 Q-耦合， 从而可以实现压控振 荡器 A和压控振荡器 B之间的正交耦合。
进一步参阅图 2和图 3， 压控振荡器 A和压控振荡器 B中均设置 有开关式电容阵列 ( Switched Capacitor Array ) 120， 开关式电容阵列 120并联接入谐振腔。 如图 3所示， 在该实例中， 通过开关式电容阵 列 120，压控振荡器 A在 4-6GHz的调谐范围被划分为 32奈调谐曲线， 压控振荡器 B在 3-4.8GHz的调谐范围被划分为 32条调谐曲线。现有 技术中， 开关式电容阵列 120由采用二进制权重的开关式电容单元组 成（例如 5个开关式电容单元生成 25条调谐曲线）。 并且， 压控振荡 器的振荡频率通过以下关系式 （ 1 ) 计算：
f 'vCO,n - ΙΓ 1 ζ ~ (、11 ) 其中， fv。。,n为对应调谐曲线的频率， Ctol.n为谐振腔的总电容， 且 C, = Cp +Ca n +Cv , 其中， Cp为总寄生电容（图 2中未示出） ， Ca n为开 关式电容阵列 120的总电容， 为所有可变电容（例如， 图 2中所示的 开关式可变电容阵列 130和具有线性化功能的固定式可变电容模块 140 ) 。
因此， 如果全部采用二进制加权开关式电容单元来控制每条调谐 曲线的频率， 那么交叠频段范围的调谐曲线之间的频率间隔是难以相 等的， 从而难以保证两个 VCO在 4-4.8GHZ频段的频率调谐特性的频 率参数方面相等。为了在 4-4.8GHZ频段使两个 VCO的调谐曲线一致， 在该开关式电容阵列 120中， 并不全部使用采用二进制权重的开关式 电容， 而是基本由 16个开关式电容单元和 4个采用二进制权重的开 关式电容单元组成。 其中， 16 个开关式电容单元控制交叠频段的 16 条调谐曲线， 通过对每个开关式电容阵列 120中的每个开关式电容单 元进行设计控制， 可以保证在交叠频段中相邻两条调谐曲线的间隔相 同， 即容易保证两个压控振荡器的输出频率相等。 另外， 在该实例中， 对应在交叠频段之外， 使用 4位二进制开关式电容以减小寄生效应。
图 7所示为图 2所示开关式电容阵列 120中所使用的开关式电容 单元的结构示意图。 如图 7所示， MIM电容（^通过中间的开关 （由 两个 NMOS管构成）控制，控制信号为 Dn，并通过上拉管（两个 PMOS 管构成）和下拉管（两个 NMOS管构成）提供偏置电平。 开关式电容 单元的具体结构不受图 7所示实施例限制。
继续参阅图 2和图 3 , 在交叠频段， 即使两个压控振荡器的输出 频率等， 还必须尽量保证两个压控振荡器的调谐增益 （tuning gain ) Kvco相等， 在图 3中反映为相应调谐曲线的 "斜率"相同。 压控振荡 器 A或 B的调谐增益 （即调谐曲线的斜率） 通过以下关系式 （2 ) 计 算：
Figure imgf000013_0001
其中， Kvc „为相应调谐曲线的斜率， fvc。,n调谐曲线对应的频 率， Ct。tn为谐振腔的总电容， Cv为可变电容， Vetrl为压控振荡器 A或 B的控制电压。
传统的压控振荡器都采用物理上固定接入电感电容谐振腔的固 定式可变电容（不具备线性化功能）来实现振荡器的压控功能。 那么， 从关系式（2 ) 中， 不难发现， 当谐振腔的总电容 Ct。t.n小的时候， 即压 控振荡器的振荡频率高的时候， 调谐曲线的斜率大； 当谐振腔的总电 容 Ct ^大的时候， 即压控振荡器的振荡频率低的时候， 调谐曲线的斜 率小。 为了抵消 Ctot.n的变化对压控振荡器调谐增益的影响， 压控振荡 器 A和压控振荡器 B中均设置开关式可变电容阵列 （ Switched Varactor Array ) 130, 开关式可变电容阵列 130并联接入谐振腔。
图 8所示为图 2所示开关式可变电容阵列 130中所使用的开关式可 变电容单元的结构示意图。 如图 8所示， 在该实例中， 开关式可变电 容阵列 130中包括 16个 （因关系式（2 ) 中的 n为 1至 16的任一整数） 如 图 8所示的开关式可变电容单元， 以分别对交叠频段的 16条调谐曲线 进行斜率调整。 当 0 增大时， 通过开关式可变电容单元 130的开关， 使更多的可变电容（Cv )接到 Vctrl端； 当 Ct ^减小时， 通过开关式可 变电容单元 130的开关， 減少接到 Vctrl端的可变电容（Cv ) 的数量， 从而实现对调谐曲线斜率的控制。因此，通过开关式可变电容阵列 130 可以控制调谐增益 Kvco , 使压控振荡器 A和 B在交叠频段中的频率调 谐特性被一致地设计为固定的 Kvco。 而且， 压控振荡器 A和 B的频率 调谐特性可被频段细调谐以补偿工艺、 电压、 温度（ Process, Voltage, Temperature, PVT ) 的波动。
需要说明的是， 开关式可变电容阵列 130中开关式可变电容单元 的具体结构不受本发发明图 8所示实施例限制。 例如， 每个开关式可 变电容单元还可以由 2个反型 MOS管 （I-MOS ) 可变电容组成， 其 可以通过开关转换实现频段细调谐、 或可以被连接到振荡器调谐电压 Vctrl以控制 VCO的调谐增益。
继续参阅图 2和图 3所示，正交耦合晶体管 Mc的偏置电流也采用如 图所示的电流源阵列 170来提供， 从而， 可以易于保证压控振荡器 A和 B在交叠频段的振荡幅度相同， 进一步使二者在交叠频段的频率调谐 特性更接近。 正交耦合晶体管 M c与电流源阵列 170之间可以设置开关 ( IQ_EN ) ， 通过控制该开关， 可以使 DMVCO在宽带模式和正交模 式之间切换。
继续参阅图 2和图 3所示，在以上实施例中，开关式电容阵列 120 和开关式可变电容阵列 130均为可数字式变化的电容阵列， 难以实现 对图 3所示的每条调谐曲线的连续调谐控制。 因此， 压控振荡器 A和 B上还均设置具有线性化功能的固定式可变电容模块 140。
图 9所示为图 2所示具有线性化功能的固定式可变电容模块 140 的结构示意图。 结合图 2和图 9所示， 具有线性化功能的固定式可变 电容模块 140同样并联接入谐振腔， 固定式可变电容模块 140中， 通 过由电阻 R2构成的分压网络， 三个偏置点的可变电容 CVAR被分别偏 置在三个不同的偏置电压 （VDD、 VDD/2 和 Vss ) 上， 可变电容 CVAR 和偏置点之间可以通过大电阻 R1进行隔离。 通过这样偏置后， 压控 振荡器 A和 B的相应调谐曲线的线性度可以得到大大提高。
继续参阅图 2和图 3所示， 尽管通过开关式电容阵列 120、 开关式 可变电容阵列 130和具有线性化功能的固定式可变电容模块 140可以 实现压控振荡器 A和 B在交叠频段的频率调谐特性基本相同， 即图 3所 示的调谐曲线在交叠频段比较接近； 但是，压控振荡器 A和 B在交叠频 段实现 QVCO的功能时， 对频率的失配和幅度的失配是非常敏感的， 为了降低对这些失配的敏感性，在压控振荡器 A和 B上还均设置有相移 网络模块 150, 图 10所示为图 2所示的相移网络模块 150的结构示意图。 在该实施例中， 使用了相移网络模块 150以后， 正交耦合晶体管 Mc的 大信号等效跨导可以通过以下关系式 （3 )计算：
Figure imgf000015_0001
其中， Gm— Mc为正交耦合晶体管 Mc的等效跨导， gm— Me为正交耦合 晶体管 Mc的跨导， 1^为相移网络模块 150中的相移电阻， （^为相移网 络模块 150中的相移电容。
从关系式（3 ) 中可以看出， 将电阻 Rs和 /或电容 Cs调整至合适值， 就能提供一个接近 90。的相移。 所提供的相移能大大降低正交模式的 DMVCO对频率及幅度失配的敏感性，以至于正交模式的 DMVCO获得 更好的相位噪音性能并获得更精确的正交输出信号。
参阅图 4和图 5， DMVCO 100工作于正交模式时， 压控振荡器 A 与压控振荡器 B同时工作在交叠频段， 并可以耦合在一起从而生成正 交信号。 图 11所示为图 2所示实施例的 DMVCO在正交模式下输出信号 的示意图。 其中， 正交模式下 DMVCO产生的正交信号 （分别为 I、 Q 两路）在图 11中示意性地给出。在频率综合器 10中，设置有 QSSBM 13 , 在频率综合器 10工作于正交模式时，将 DMVCO 100所输出的正交信号 与除法器 173输出的正交信号输入至 QSSBM 13， QSSBM 13进行上变 频处理， 从而可以输出 5-6GHz的正交本振信号。 因此， 如图 6所示， 频率综合器 10可以通过混频器（MIX )覆盖 5-6GHz的频段。 可以理解 的是， 在其它实施例中， 除法器链路中其它除法器所输出的正交信号 也可以输入至 QSSBM 13中。
由上述可知， 频率综合器在覆盖较低输出频段（例如 0.4- 3GHZ ) 时， 频率综合器和 DMVCO工作于宽带模式， DMVCO在相对较低频率 段调谐范围工作， 频率综合器所使用的除法器和緩冲器的功耗就能大 大降低； 频率综合器在覆盖较高输出频段（例如 5-6GHz )时， 频率综 合器和 DMVCO工作于正交模式，利用 DM VCO中的两个工作于交叠频 段的压控振荡器来提供 QSSBM所需要的正交信号， 避免了额外的 PPF 和 QVCO的使用， 因此， 使用 DMVCO的频率综合器的结构并不复杂， 不但其输出频率范围可以得到保证， 而且功耗并不增加， 同时兼顾了 输出频率范围和功耗的要求。频率综合器中使用的 QS SBM可以抑制混 频时的镜像信号， 提高压控振荡器的频谱纯度， 频率杂散性好。
需要说明的是， 以上实施例中对覆盖 0.4-3GHZ以及 5-6GHz频段 的频率综合器以及该频率综合器所使用的 DMVCO进行了说明， 本领 域技术人员根据以上教导和启示， 通过软件定义无线电的技术， 实现 对其它带宽范围的覆盖。 因此， 该发明 DMVCO、 频率综合器的可重 构性好。
进一步， 如图 4所示， 频率综合器 10中， 本振发生器一般地还包 括锁相环 （PLL )模块 （图中未示出） ， 锁相环模块可以用来控制并 稳定 DMVCO的输出频率。 应用图 4所示的频率综合器 10， 可以进一步 形成无线电接收装置（例如无线电接收机） ， 无线电接收装置所包括 的其它具体部件在此不再 详述。
以上例子主要说明了本发明的 DMVCO、 使用该 DMVCO的频率 综合器以及无线电接收装置。 尽管只对其中一些本发明的实施方式进 行了描述， 但是本领域普通技术人员应当了解， 本发明可以在不偏离 其主旨与范围内以许多其他的形式实施。 因此， 所展示的例子与实施 方式被视为示意性的而非限制性的， 在不脱离如所附各权利要求所定 义的本发明精神及范围的情况下， 本发明可能涵盖各种的修改与替 换。

Claims

权 利 要 求
1. 一种双模压控振荡器， 其特征在于， 包括第一压控振荡器单元 和第二压控振荡器单元， 所述第一压控振荡器单元工作于第一频段， 所述第二压控振荡器工作于第二频段， 所述第一频段与所述第二频段 之间具有交叠频段；
每个所述压控振荡器单元均包括正交耦合模块；
所述双模压控振荡器可操作地工作于宽带模式或正交模式； 其中， 工作于所述宽带模式时， 仅通过第一压控振荡器单元或第 二压控振荡器单元产生相应频率的差分信号；
工作于所述正交模式时， 所述第一压控振荡器单元和第二压控振 荡器单元同时工作在所述交叠频段， 通过所述第一压控振荡器单元和 第二压控振荡器单元的正交耦合模块， 将所述第一压控振荡器单元与 所述第二压控振荡器单元耦合在一起， 并产生相应频率的正交信号。
2. 如权利要求 1所述的双模压控振荡器， 其特征在于， 所述第一 压控振荡器单元和第二压控振荡器单元在所述交叠频段的频率调谐 特性基本相同。
3. 如权利要求 2所述的双模压控振荡器， 其特征在于， 每个所述 压控振荡器单元还包括：
开关式电容阵列， 其用于控制所述第一压控振荡器单元和所 述第二压控振荡器单元在所述交叠频段内的相应调谐曲线的所 在频率相同； 以及
开关式可变电容阵列， 其用于控制所述第一压控振荡器单元 和所述第二压控振荡器单元在所述交叠频段内的相应调谐曲线 的调谐增益相同。
4. 如权利要求 3所述的双模压控振荡器， 其特征在于， 所述开关 式电容阵列包括多个开关式电容单元， 每个开关式电容单元对应控制 所述交叠频段内的一条调谐曲线的所在频率。
5. 如权利要求 3所述的双模压控振荡器， 其特征在于， 所述开关 式可变电容阵列包括多个开关式可变电容单元， 每个开关式可变电容 单元对应控制所述交叠频段内的一条调谐曲线的调谐增益。
6. 如权利要求 4所述的双模压控振荡器， 其特征在于， 所述开关 式电容阵列还包括多个采用二进制权重的开关式电容单元， 对应控制 所述压控振荡器在所述交叠频段外的调谐曲线的所在频率。
7. 如权利要求 3所述的双模压控振荡器， 其特征在于， 每个所述 压控振荡器单元还包括：
具有线性化功能的固定式可变电容模块， 其用于控制所述第 一压控振荡器单元和所述第二压控振荡器单元的调谐曲线的线 性度。
8. 如权利要求 3或 7所述的双模压控振荡器， 其特征在于， 每个 所述压控振荡器单元还包括：
相移网络模块， 其用于提供基本为 90。的相移以降低所述第 一压控振荡器单元和第二压控振荡器单元在所述交叠频段对频 率失配及幅度失配的敏感性。
9. 如权利要求 2所述的双模压控振荡器， 其特征在于， 每个所述 压控振荡器单元还包括片上电感和寄生电容。
10. 如权利要求 1所述的汉模压控振荡器， 其特征在于， 每个所 述压控振荡器单元还包括：
负阻模块， 其为所述压控振荡器单元的谐振腔提供振荡所需 要的能量；
其中， 所述负阻模块包括两个相互交叉耦合的晶体管以及第一电 流源阵列； 通过控制所述第一电流源阵列来调整流经所述交叉耦合晶 体管的电流大小， 以控制所述第一压控振荡器单元或第二压控振荡器 单元的输出信号幅度。
1 1. 如权利要求 10所述的双模压控振荡器， 其特征在于， 所述第 一电流源阵列与所述交叉耦合的晶体管之间设置第一开关， 所述双模 压控振荡器工作于宽带模式时， 控制所述第一开关以选择所述第一压 控振荡器单元和第二压控振荡器单元中的任意一个。
12. 如权利要求 1所述的双模压控振荡器， 其特征在于， 所述正 交耦合模块包括两个正交耦合晶体管， 其中， 第一压控振荡器单元 / 第二压控振荡器单元中的每个所述正交耦合晶体管的栅极与第二压 控振荡器单元 /第一压控振荡器单元的谐振腔的相应输出信号耦合。
13. 如权利要求 12所述的双模压控振荡器， 其特征在于， 每个所 述压控振荡器单元中设置用于为所述正交耦合晶体管提供偏置电流 的第二电流源阵列。
14. 如权利要求 13所述的双模压控振荡器， 其特征在于， 所述正 交耦合晶体管与所述第二电流源阵列之间设置第二开关， 通过控制所 述开关以实现所述双模压控振荡器在所述宽带模式和所述正交模式 之间切换。
15. 如权利要求 1所述的双模压控振荡器， 其特征在于， 所述第 一频段基本为 3GHz至 4.8GHz,所述第二频段基本为 4GHz至 6GHz, 所述交叠频段基本为 4GHz至 4.8GHz。
16. 一种频率综合器， 包括本振发生器， 其特征在于， 所述本振 发生器包括：
如权利要求 1所述的双模压控振荡器，
正交单边带混频器， 以及
一级或一级以上的除法器；
其中， 所述频率综合器可操作地工作于宽带模式或正交模式； 工作于所述宽带模式时， 第一压控振荡器单元或第二压控振荡器 单元所产生差分信号通过所述除法器处理以产生第一正交本振信号； 工作于所述正交模式时， 将正交耦合处理后输出得到的正交信号 和所述除法器所输出的正交信号输入至所述正交单边带混频器进行 上变频处理以产生第二正交本振信号；
所述第一正交本振信号用于覆盖频率相对低的第一输出频段， 所 述第二正交本振信号用于覆盖频率相对高的第二输出频段。
17. 如权利要求 16所述的频率综合器， 其特征在于， 所述本振发 生器还包括第一级频率选通器和第二级频率选通器；
其中， 所述第一级频率选通器选择第一压控振荡器单元或第二压 控振荡器单元所述输出的相应频率的差分信号至所述除法器， 所述第 二级频率选通器选择任一所述除法器所输出的正交信号以输出所述 第一正交本振信号。
18. 如权利要求 16所述的频率综合器， 其特征在于， 所述第一输 出频段基本为 0.4GHz至 3GHz, 所述第二输出频段基本为 5 GHz至
6GHz。
19. 一种无线接收装置， 其特征在于， 包括如权利要求要求 16所 述的频率综合器。
PCT/CN2011/000658 2010-04-15 2011-04-14 双模压控振荡器、频率综合器及无线接收装置 Ceased WO2011127754A1 (zh)

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