Classifications

• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03F—AMPLIFIERS
  • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
  • H03F1/32—Modifications of amplifiers to reduce non-linear distortion
  • H03F1/3241—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03C—MODULATION
  • H03C3/00—Angle modulation
  • H03C3/38—Angle modulation by converting amplitude modulation to angle modulation
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
  • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
  • H04L27/36—Modulator circuits; Transmitter circuits
  • H04L27/362—Modulation using more than one carrier, e.g. with quadrature carriers, separately amplitude modulated
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
  • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
  • H04L27/36—Modulator circuits; Transmitter circuits
  • H04L27/366—Arrangements for compensating undesirable properties of the transmission path between the modulator and the demodulator
  • H04L27/367—Arrangements for compensating undesirable properties of the transmission path between the modulator and the demodulator using predistortion
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03F—AMPLIFIERS
  • H03F2200/00—Indexing scheme relating to amplifiers
  • H03F2200/331—Sigma delta modulation being used in an amplifying circuit

Definitions

• This invention relates to modulation systems and methods, and more particularly to IQ modulation systems and methods.
• Modulation systems and methods are widely used in transmitters to modulate information including voice and/or data onto a carrier.
• the carrier may be a final carrier or an intermediate carrier.
• the carrier frequency can be in UHF, NHF, RF, microwave or any other frequency band.
• Modulators are also referred to as “mixers” or “multipliers”. For example, in a mobile radiotelephone, a modulator is used for the radiotelephone transmitter.
• FIG. 1 illustrates a conventional IQ modulator.
• an IQ modulator 10 also referred to as a "quadraphase modulator” or a “quadrature modulator” includes a quadrature splitter 20, also known as a 90° phase shifter, and a pair of multipliers 16a, 16b coupled to the quadrature splitter.
• In-phase (I) data 11a and quadrature-phase (Q) data lib are coupled to a respective multiplier or mixer 16a, 16b.
• NCO Voltage Controlled Oscillator
• Digital input data is converted to analog data by I Digital-to-Analog Converter (DAC) 14a and Q DAC 14b, respectively.
• the outputs of the respective DACs 14a and 14b are applied to the respective low pass filters 12a and 12b to provide the respective I and Q data inputs 11a and lib.
• the modulator modulates the input data on a carrier 13, by summing the outputs of the multipliers 16a, 16b at a summing node 218.
• the modulated carrier is amplified by a power amplifier 22 and transmitted via an antenna 24. In modern radiotelephone communications, mobile radiotelephones continue to decrease in size, cost and power consumption.
• the power amplifier 22 of an IQ modulator may consume excessive power due to efficiency limitations therein. More specifically, it is known to provide a linear class-A or class- AB power amplifier 22 that may have efficiencies as low as 30 percent or less. Thus, large amounts of battery power may be wasted as heat. Moreover, the noise figure of a conventional IQ modulator may be excessive so that high cost Surface Acoustic Wave (SAW) filters may need to be used.
• SAW Surface Acoustic Wave
• the rTheta technique may require the oscillator phase lock loop to support the phase modulation bandwidth.
• radiotelephone signals such as TDMA and CDMA signals, it may be increasingly difficult to provide the requisite bandwidth in the oscillator phase lock loop.
• Power modulation systems include first and second power amplifiers, each including a signal input, a supply voltage input and a power output.
• the first and second power amplifiers are preferably class-C power amplifiers.
• a source of first, second, third and fourth reference frequency signals is also provided.
• the first and second reference frequency signals are inverted relative to one another, and the third and fourth reference frequency signals are inverted relative to one another.
• the first, second, third and fourth reference frequency signals are 0°, 180°, 90° and 270° phase shifted reference frequency signals, respectively.
• a switching system is also provided that selectively applies one of the first and second reference frequency signals to the signal input of the first power amplifier as a function of the polarity of one of the I and Q input signals.
• the switching system also selectively applies one of the third and fourth reference frequency signals to the signal input of the second power amplifier as a function of the polarity of the other of the I and Q input signals.
• a third amplifier preferably a class-D amplifier, is responsive to the one of the I and Q input signals to supply a first variable supply voltage to the supply input of the first amplifier.
• the supply voltage input preferably corresponds to the drain or collector voltage of the output stage of the first amplifier.
• a fourth amplifier also preferably a class-D amplifier, is responsive to the other of I and Q input signals to supply a second variable supply voltage to the supply input of the second amplifier.
• a coupler couples the power outputs of the first and second power amplifiers to a load such as a radiotelephone antenna. Accordingly, power IQ modulation is provided that can use class-C and class-D amplifiers that are highly efficient.
• the first, second, third and fourth reference frequency signals may be generated by a controlled oscillator that produces the first reference frequency signal.
• a first inverter is responsive to the controlled oscillator to produce the second reference frequency signal.
• a phase shifter is responsive to the controlled oscillator to produce the ⁇ ird reference frequency signal.
• a second inverter is responsive to the phase shifter to produce the fourth reference frequency signal.
• the switching system may include a first switch that is responsive to the one of the I and Q input signals being positive, to couple the first reference frequency signal to the first power amplifier.
• the first switch is also responsive to the one of the I and Q input signals being negative, to couple the second reference frequency signal to the first power amplifier.
• a second switch is responsive to the other of the I and Q input signals being positive, to couple the third reference frequency signal to the second power amplifier.
• the second switch is also responsive to the other of the I and Q input signals being negative to couple the fourth reference frequency signal to the second power amplifier.
• a first polarity detector may be provided, that is coupled between the one of the I and Q input signals and the first switch, to detect whether the one of the I and Q input signals is positive or negative.
• a second polarity detector may also be provided, that is coupled between the other of I and Q input signals and the second switch, to detect whether the other of the I and Q input signals is positive or negative.
• a first analog-to-digital converter also may be coupled between the one of the
• a second analog-to-digital converter also may be provided, that is coupled between the other of the I and Q input signals and the fourth amplifier, such that the fourth amplifier is responsive to a digital representation of the other of the I and Q signals to supply a second variable voltage supply to the supply input of the second amplifier.
• the analog-to-digital converters are preferably delta-sigma analog-to-digital converters.
• the coupler may include a matching network that couples the first and second power amplifiers to a load such as a radiotelephone antenna.
• the coupler preferably isolates the first and second power amplifiers from one another while coupling the first and second power amplifiers to the load.
• the coupler may include a first quarter wavelength transmission line that couples the first power amplifier to the matching network and a second quarter wavelength transmission line that couples the second power amplifier to the matching network.
• the coupler may include a first isolator that couples the first power amplifier to the matching network and a second isolator that couples the second power amplifier to the matching network.
• the matching network may also be a voltage-to-current converting matching network, such as a quarter-wave transmission line or discrete component equivalent thereof.
• These coupling techniques also are illustrated in Application Serial No. 09/054,063, filed April 2, 1998, entitled “Hybrid Chireix/Doherty Amplifiers and Methods " to the present co-inventor Dent assigned to the assignee of the present invention, the disclosure of which is hereby incorporated herein by reference, and Application Serial No. 09/054,060, filed April 2, 1998 entitled “Power Waveform Synthesis Using Bilateral Devices " to the present co- inventor Dent, assigned to the assignee of the present invention, the disclosure of which is hereby incorporated herein by reference.
• the first and second class-C amplifiers may have a nonlinear amplitude and/or phase response as a function of the respective first and second variable supply voltages.
• Power modulation according to the invention may include amplitude distortion and phase distortion compensation that is responsive to the I and Q input signals, to modify the I and Q input signals to reduce the nonlinear amplitude response.
• compensation systems and methods are responsive to the I and Q input signals, to generate predistorted I and Q signals, and to apply the predistorted I and Q signals to the switching system and to the third and fourth amplifiers, to thereby compensate for the nonlinear distortion.
• a lookup table may be provided that is responsive to the I input signal, to produce a first amplitude distortion compensating value and a first phase compensating value, and that is responsive to the Q input signal to produce a second amplitude distortion compensating value and a second phase compensating value.
• a pair of lookup subtables may also be provided.
• a summer is also provided that adds the first amplitude distortion compensating value and the second phase compensating value to produce the predistorted I signal, and that subtracts the first phase compensating value from the second amplitude distortion compensating value, to produce the predistorted Q signal.
• a separate adder and subtracter also may be provided. Compensation systems and methods according to the invention also may be used separately to precompensate for distortion in an amplifier that produces distortion.
• the present invention can provide power IQ modulation at high efficiency to thereby allow reduced size, cost and/or power consumption of a mobile radiotelephone or other transmitter.
• Wide bandwidth oscillator phase lock loops need not be used, and high performance SAW filters may not be needed to reduce transmitted noise.
• Associated power modulation methods are also provided.
• Figure 1 is a block diagram of a conventional IQ modulator.
• Figures 2-5 are block diagrams of power IQ modulation systems and methods according to the present invention.
• FIG. 6 is a block diagram of compensation systems and methods according to the present invention. Detailed Description of Preferred Embodiments
• the present invention may be embodied as systems (apparatus) or methods.
• the present invention may take the form of an entirely hardware embodiment or an embodiment combining software and hardware aspects. Accordingly, individual blocks and combinations of block in the drawings support combinations of means for performing the specified functions and combinations of steps for performing the specified functions. Each of the blocks of the drawings may be embodied in many different ways, as is well known to those of skill in the art.
• power IQ modulation systems and methods will now be described. As shown in Figure 2, power modulation systems and methods 200 modulate and amplify in-phase I and quadrature-phase Q input signals, in respective I and Q paths.
• first and second power amplifiers 210a and 210b are included.
• Each of the first and second power amplifiers 210a and 210b includes a respective signal input 212a, 212b, supply input 214a, 214b and power output 216a, 216b.
• a reference frequency source 220 generates first, second, third and fourth reference frequency signals 222a-222d.
• the first and second reference frequency signals 222a and 222b are inverted relative to one another and the third and fourth reference frequency signals 222c and 222d are inverted relative to one another.
• the first through fourth reference frequency signals 222a-222d are 0°, 180°, 90° and 270° phase shifted reference frequency signals, respectively,
• the reference frequency source 220 preferably includes a controlled oscillator such as a voltage controlled oscillator 224 that produces the first reference frequency signal 222a
• a phase-locked loop may also be included.
• a first inverter 226 is responsive to the controlled oscillator 224 to produce the second reference frequency signal 222b.
• a phase shifter 228 is responsive to the controlled oscillator 224 to produce the third reference frequency signal 222c.
• a second inverter 232 is responsive to the phase shifter 228 to produce the fourth reference frequency signal 222d.
• many other techniques may be used to generate a reference frequency or phase inverted reference frequency signals under control of a sign bit.
• a switching system 240 selectively applies one of the first and second reference frequency signals 222a or 222b to the signal input 212a of the first power amplifier 210a as a function of the polarity of one of the I and Q input signals, for example the Q input signal.
• the switching system 240 also selectively applies one of the third and fourth reference frequency signals 222c or 222d to the signal input 212b of the second power amplifier 210b as a function of the polarity of the other of the I and Q input signals, for example, the I input signal.
• First and second drivers 242a, 242b may be inserted between the switching system 240 and the respective class-C amplifier 210a, 210b to overdrive the class-C amplifier 210a, 210b, if necessary, to achieve the desired gain or drive power level for amplifiers 210a, 210b.
• the switching system preferably comprises a first switch such as a first Single-Pole Double-Throw (SPDT) switch 244a that is responsive to the one of the I and Q input signals being positive to couple the first reference frequency signal 222a to the first power amplifier 210a, and that is responsive to the one of the I and Q input signals being negative to couple the second reference frequency signal 222b to the first power amplifier 210a.
• a second switch such as a second SPDT switch 244b, is responsive to the other of the I and Q input signals being positive to couple the third reference frequency signal 222 c to the second power amplifier 210b, and that is responsive to the other of the I and Q input signals being negative to couple the fourth reference signal 222d to the second power amplifier 210b.
• a balanced modulator may be used instead of SPDT switches to selectively invert a reference signal.
• the switching system 240 also preferably comprises a first polarity detector 246a that is coupled between the one of the I and Q input signals and the first switch 244a, to detect whether the one of the I and Q input signals is positive or negative.
• a second polarity detector 246b is coupled between the other of the I and Q input signals and the second switch 244b, to detect whether the other of the I and Q input signals is positive or negative.
• a third amplifier 250a preferably a class-D amplifier, is responsive to the one of the I and Q input signals, for example the Q input signal, to supply a first variable supply voltage to the supply input 214a of the first amplifier 210a.
• a fourth amplifier 250b is responsive to the other of the I and Q input signals, for example the I input signal, to supply a second variable supply voltage to the supply input 214b of the second amplifier 210b. In order to generate the appropriate inputs for the third and fourth amplifiers
• a first analog-to-digital converter preferably a delta- sigma analog-to-digital converter 260a
• the third amplifier is responsive to a digital representation to the one of the I and Q signals to supply a first variable supply voltage to the supply input 214a of the first amplifier 210a.
• a second analog- to-digital converter preferably a second delta-sigma analog-to-digital converter 260b, is coupled between the other of the I and Q input signals and the fourth amplifier 250b, such that the fourth amplifier 250b is responsive to a digital representation of the other of the I and Q signals to supply a second variable supply voltage input 214b of the second amplifier 210b.
• low pass filters 262a, 262b, 264a and 264b may filter the Q and I inputs and the outputs of the third and fourth amplifiers 250a and 250b respectively.
• I and Q signals are generated in digital form by a Digital Signal Processor (DSP).
• the DSP can include lowpass filters 262a and 262b in digital form.
• Delta-Sigma converters 260a and 260b can then convert the filtered digital sample streams from the DSP from a Pulse Coded Modulation (PCM) format to a Delta-Sigma representation by an entirely digital logic process and system.
• PCM Pulse Coded Modulation
• a coupler 270 couples the power outputs 216a and 216b of the first and second power amplifiers 210a and 210b to a load 272 such as radiotelephone antenna.
• the coupler 270 includes a matching network that couples the first and second power amplifiers to the load 272 while preferably isolating the first and second power amplifiers 210a and 210b from one another.
• the frequency of the controlled oscillator 224 is preferably set to the transmitted carrier frequency.
• the 90° phase shifter 228 preferably creates an identical signal but with a phase shift of 90° from the original signal.
• the original signal and the 90° shifted signal are then sent to inverters 226 and 232, which are amplifiers with a gain of-1.
• four reference frequency signals 222a-222d are present. They preferably represent +sin and -sin, and +cos and -cos of the controlled oscillator 224.
• the two switches 244a and 244b are controlled to select a polarity of the sin for one channel and a polarity of the cos for the other channel. These signals proceed independently to the driver amplifier stages 242a and 242b and the power amplifier outputs stages 210a and 210b.
• the control of the polarity of the sin or cos signals preferably coincides with the amplitude of the I or Q channel becoming zero. This can reduce introduction of polarity change induced noise into the output signal spectrum.
• the two power amplifier output stages are preferably coupled with series blocking capacitors in the matching network 270 so that the supply voltages, preferably corresponding to drain voltages of the class-C amplifiers, can be controlled independent of each other.
• the distance from the class-C amplifiers 210a and 210b to the matching network 270 is short relative to the wavelength of the carrier frequency.
• the class-C amplifier supply voltages 214a and 214b preferably are created in an efficient manner by class-D amplifiers 250a and 250b, using low pass filters 246a and 246b to remove the switching noise.
• the input signals to the class-D amplifiers 250a and 250b are created by a delta-sigma process so that the noise in the bandwidth of the low pass filters 246a and 246b can be reduced.
• the Class D amplifiers are preferably capable both of sourcing and sinking current so that current can even flow from the amplifiers 210a and 210b back to the supply during periods of a modulation cycle.
• the power amplifiers 210a and 210b preferably are saturated class-C amplifiers rather than linear power amplifiers. Higher efficiency may therefore be obtained. Out-of-band noise can be reduced, which can eliminate the need for additional SAW filters in the chain. Finally, a large modulation bandwidth can be supported. This bandwidth may be limited only by the filter bandwidth of the low pass filters 264a and 264b that follow the class-D amplifier stages 250a and 250b, respectively.
• FIG 3 is a block diagram of a second embodiment of power IQ modulation systems and methods according to the invention.
• Power modulation systems and methods 200' of Figure 3 are similar to those of Figure 2, except that a respective quarter wavelength transmission line 310a and 310b of appropriate impedance is included in each branch of the final output stage between the power output 216a and 216b of the class-C amplifiers 210a and 210b and the matching network 270, as disclosed in the above-incorporated applications.
• a pair of isolators 410a and 410b may be included between the power outputs 216a and 216b and the matching network 270.
• matching network 270 may be two separate matching networks, each connected to an isolator 410 that increases the impedance from 50 ohms at the isolator to 100 ohms at the point where the two matching networks are joined and attached to the feedline to antenna 272.
• FIG. 5 illustrates yet another embodiment 200'" wherein the matching network 270 is replaced by a pair of voltage-to-current (VI) converters and matching networks 570a and 570b.
• VI voltage-to-current
• each of the embodiments of Figures 3-5 may have various advantages and disadvantages.
• the embodiment of Figure 3 uses quarter wavelength transmission lines 310a and 310b to convert the predominantly voltage source of the power amplifier stages 210a and 210b to currents. The currents can then be added in parallel and supplied to the load through the matching network 270.
• the embodiment of Figure 4 uses a pair of isolators 410a and 410b to combine the outputs from the two power amplifier output stages 210a and 210b.
• the isolators may reduce interaction between the two power amplifier stages 210a and 210b at the expense of overall efficiency.
• the embodiment of Figure 5 uses the matching network 570a and 570b to convert the voltage output of the class-C amplifiers 210a and 210b to current for addition in the load.
• Class-C amplifiers 210a, 210b can have a nonlinear relationship between supply voltage and output level, resulting in distortion. Such amplifiers can also suffer from Amplitude Modulation to Phase Modulation (AM-to-PM) conversion, whereby modulation of the amplitude produces undesired modulation of the phase.
• AM-to-PM Amplitude Modulation to Phase Modulation
• the invention can also include methods and systems to compensate for such signal distortion.
• Such precompensation can be either feedforward precompensation, where I and Q signals are precompensated for the known distortion that the amplifiers will introduce, or feedback precompensation where the phase or amplitude of the output signal is sensed, compared to desired values to obtain error quantities, the error quantities then being used to generate corrected I, Q values that will result in the desired output signal being produced more accurately.
• Feedback systems can also be either real-time feedback systems, or "off-line” or “learning” systems which gradually adjust the way in which I, Q values are precompensated so as to produce the desired output signal more accurately.
• output signal amplitude may be measured using a power detector, but output phase may be more difficult to measure, as phase generally is measured relative to a phase reference signal. Therefore, although a feedback technique of controlling amplitude linearity may be quite practical, a feedforward technique of compensating for AM-to-PM conversion may be more practical.
• a preferred technique of compensating distortion comprises use of a digital signal processor to generate compensated I and Q signals.
• a technique of precompensating conventional amplifiers as well as AM-to-PM conversion that depends on the total output signal amplitude can comprise using the digital signal processor to compute the desired total amplitude of the output signal from the square root of I 2 + Q 2 .
• the desired amplitude is then applied as an address to a prestored look-up table to obtain an amplitude compensating value that is used to scale both the I and Q values to obtain amplitude-compensated I and Q values which, when then used for modulation, can result in the correct, desired output signal amplitude being produced.
• Either the original desired amplitude or the compensated amplitude may then be applied to a prestored phase compensating table to obtain a phase compensating value that compensates for AM-to-PM conversion.
• the phase compensating value is then used in complex multiplication with the amplitude-compensated I and Q values to obtain phase and amplitude compensated I and Q values that, when used for modulation, can result in the correct, desired signal amplitude being produced.
• F “ (A)/A cos( ⁇ ) and F " (A)/ A sin( ⁇ ) are purely functions of the amplitude A, they can be precomputed and stored in a look-up table as F1(A) and F2(A), and then used as above. Thus, only four multiplications may be used to correct both phase and amplitude distortion.
• each amplifier amplifies only the I part or the Q part. Therefore it may be expected that amplitude distortion in the I- amplif ⁇ er would be dominantly dependent on only the I value, while amplitude distortion in the Q-amplifier would be dominantly dependent on the Q value.
• AM-to-PM conversion in the I-amplifier would change the signal phase away from zero (the cosine wave phase) by a phase error amount depending dominantly on the I-value
• AM-to-PM conversion in the Q amplifier would change its signal phase away from 90 degrees (the sine wave phase) by a phase error amount depending dominantly on the Q value.
• a different precompensation structure may be used to compensate for two different values of AM-to-PM conversion and amplitude distortion, one dependent only on the I-value and the other dependent only on the Q value.
• the AM-to-PM conversion of the I amplifier is described by ⁇ (I) then the magnitude of the I value will be reduced to Icos( ⁇ (I)) and an additive error component Isin( ⁇ (I)) will be added to the Q value.
• ⁇ (Q) describing AM-to-PM conversion of the Q amplifier by ⁇ (Q), the magnitude of Q will be reduced to Qcos( ⁇ (Q)), while an additive error Qsin( ⁇ (Q)) will be subtracted from the I component.
• a block diagram of precompensation according to the invention is shown in Figure 6.
• the desired I value is applied to a look-up table 600a to obtain a compensated I value F1(I), also referred to as a first amplitude distortion compensating value 610a, and an AM-to-PM compensating value F2(I), also referred to as a first phase compensating value 620a.
• the desired Q value is applied to a look-up table 600b, which can be the same look-up table if the amplifiers 210a, 210b and their associated high-level class-D modulators 250a, 250b are identical.
• the two look-up tables denoted by 600a, 600b in Figure 6 can be different subtables of a single lookup table system 600.
• the Q value addresses the lookup table 600b, to produce an amplitude- precompensated Q value F1(Q), also referred to as a second amplitude distortion compensating value 610b, and an AM-to-PM compensating value F2(Q), also referred to as a second phase compensating value 620b.
• the value F2(Q) is added to the value F1(I) in adder 630a while the value F2(I) is subtracted from the value F1(Q) in subtractor 630b of a summer 630.
• Summer 630 produces precompensated or predistorted I and Q values which, when applied to modulate respective amplifiers, will result in an output after combining the amplifier outputs that more accurately follows the desired I and Q values.
• the compensated I and Q values may be used to modulate respective amplifiers by applying the modulus (magnitude) of the compensated I value to the high-level, class-D I-modulator 260b, 250b, 264b while applying the sign of the compensated I value to select either inverted or not-inverted phase of cosine wave reference signal from generator 224.
• the modulus (magnitude) of the compensated Q value is applied to the Q-modulator 260a, 250a, 264a while the sign of the compensated Q value selects either inverted or not-inverted sine wave drive from reference 224 shifted in phase by 90 degrees 228.
• the above techniques which can compensate either for total-amplitude dependent distortion or distortion dependent on the I and Q components separately may be used together, if distortion of both types exists.
• the two techniques may be applied in any order, by ensuring that the look-up tables precomputed for the first- applied technique account for distortion already compensated by the second-applied technique.
• high efficiency power IQ modulation systems and methods may be provided to thereby allow a reduction in the cost, size and/or power consumption of a cellular radiotelephone or other transmitter.

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• 1999
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