Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04J—MULTIPLEX COMMUNICATION
  • H04J1/00—Frequency-division multiplex systems
  • H04J1/02—Details
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04J—MULTIPLEX COMMUNICATION
  • H04J4/00—Combined time-division and frequency-division multiplex systems
  • H04J4/005—Transmultiplexing
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H17/00—Networks using digital techniques
  • H03H17/02—Frequency selective networks
  • H03H17/0211—Frequency selective networks using specific transformation algorithms, e.g. WALSH functions, Fermat transforms, Mersenne transforms, polynomial transforms, Hilbert transforms
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H17/00—Networks using digital techniques
  • H03H17/02—Frequency selective networks
  • H03H17/0248—Filters characterised by a particular frequency response or filtering method
  • H03H17/0251—Comb filters

Definitions

• Such a communication unit structure is essentially equivalent to a single radio frequency (RF) and intermediate frequency (IF) band analog signal processing portion followed by or preceded by a single digital processing portion which manipulates the digitized signal as if it represented multiple communication channels.
• RF radio frequency
• IF intermediate frequency
• another possible technique for providing this type of communication unit structure is through the use of Discrete Fourier Transforms (DFT's) in a DFT bank and Inverse Discrete Fourier Transforms (IDFT's) in an IDFT bank or similar digital filtering techniques, to synthesize a series of adjacent narrow bandwidth channels.
• DFT's Discrete Fourier Transforms
• IDFT's Inverse Discrete Fourier Transforms
• a number of current and future planned information signal coding and channelization standards (i.e., open air interface standards) exist. These coding and channelization standards which have channels allocated in equal portions of frequency bandwidth include channelization structures based on frequency division multiple access, time division multiple access, and frequency hopping code division multiple access.
• Some of the current and future planned coding and channelization standards have names including: Advanced Mobile Phone Service (AMPS), Narrow Advanced Mobile Phone Service (NAMPS), Total Access Communication System (TACS), Japanese Total Access Communication System (JTACS), United States Digital Cellular (USDC), Japan Digital Cellular (JDC), Groupe Special Mobile (GSM), Frequency Hopping Spread Spectrum (FH-SS), Cordless Telephone 2 (CT2), Cordless Telephone 2 Plus (CT2 Plus), and Cordless Telephone 3 (CT3).
• AMPS Advanced Mobile Phone Service
• NAMPS Narrow Advanced Mobile Phone Service
• TACS Total Access Communication System
• JTACS Japanese Total Access Communication System
• USDC Digital Cellular
• JDC Japan Digital Cellular
• GSM Groupe Special Mobile
• these multi-channel communication units include a digital signal processing portion which may be reprogrammed, at will, through software during the manufacturing process or in the field after installation such that these multi-channel communication units may operate in accordance with any one of several information signal coding and channelization standards.
• a commutator combines portions of the plurality of digitized signals into a composite digitized signal. Then, an digital-to- analog converter generates a composite analog transmission signal from the composite digitized signal. Finally, transmitter transmits the composite analog transmission signal over a frequency band.
• FIG. 1 is a block diagram showing a preferred embodiment receiver in accordance with the present invention.
• FIG. 8 is a block diagram showing a preferred embodiment fifth order differencer in accordance with the present invention for use in the preferred embodiment receiver and transmitter shown in FIGs. 1 through 6. Detailed Description
• each cell or service region (served by a base site) are typically assigned according to a channel reuse separation pattern (e.g., a 21- cell or 24-cell reuse pattern).
• a channel reuse separation pattern e.g., a 21- cell or 24-cell reuse pattern.
• This type of channel assignment based on a reuse separation pattern reduces co-channel and adjacent channel interference. Therefore, each base site is designed to receive only a subset of all available communication channels (i.e., 1/21 or 1/24 of all available channels). It will be appreciated by those skilled in the art that other reuse separation patterns may be used by an AMPS or NAMPS communication system and as such each base site will need to be designed to receive the number of channels available for serving the cell or service region served by the base site.
• the receiver preferably includes an input such as an antenna 100 which intercepts electromagnetic radiation within the frequency band of interest and transducer (not shown) for converting the intercepted electromagnetic radiation into an electrical signal.
• an input such as an antenna 100 which intercepts electromagnetic radiation within the frequency band of interest and transducer (not shown) for converting the intercepted electromagnetic radiation into an electrical signal.
• transducer not shown
• other types of input devices may be used for intercepting or capturing electromagnetic radiation.
• a wave guide, a coaxial cable, an optical fiber, or an infrared frequency transducer may be used to intercept electromagnetic radiation for subsequent input into the preferred embodiment receiver.
• the receiver preferably consists of a front end (analog portion) consisting of elements known to those skilled in the art that provide a low noise figure and ample protection to undesired out-of-band signals. These elements consist of a preselect band pass filter 102.
• the IF band electrical signal 114 is input to another down converter.
• This down converter consists of a mixer 118, a frequency agile local oscillator 116, and a low pass filter 122.
• the electrical signal 114 is operatively coupled to one input of the mixer 118 and the output of the oscillator 116 is provided to another input of mixer 118.
• Mixer 118 mixes the IF band electrical signal 114 down to a baseband electrical signal 120.
• the baseband electrical signal 120 is filtered by a low pass filter 122 to eliminate undesired frequency components into a conditioned baseband electrical signal 124.
• the conditioned baseband electrical signal 124 preferably is operatively coupled to a digitizing device 126 (i.e., analog-to-digital (A/D) converter) which digitizes a portion of the baseband electrical signal 124 into a digitized signal 128.
• the digitized signal 128 represents the electromagnetic radiation within the intercepted frequency band.
• FIG. 2 shows a graphical representation 128' of this digitized signal 128 which contains a plurality of channels (i.e., N channels) contained in frequency divided passbands.
• the sampling rate of the digitizing device 126 preferably is set to be within the Nyquist criterion of at least twice the highest desired frequency with the baseband electrical signal 124.
• Commutator 130 receives the digitized signal 128 and operatively couples portions of the digitized signal 128 to a discrete Fourier transform pre-filter bank including filters 132, 134, 136, and 138 (i.e., f ⁇ , f2, f3, and fN).
• filters 132, 134, 136, and 138 i.e., f ⁇ , f2, f3, and fN.
• filters 132, 134, 136, and 138 are shown in FIG. 1 ; however, it will be appreciated by those skilled in the art that the principles described herein may be applied to more or less filters in the filter bank.
• Discrete Fourier transformer 148 preferably operates at the same or higher sampling rate of the filters 132, 134, 136, and 138.
• the discrete Fourier transformer 148 performs one transform (e.g., fast
• FFTs allow the receiver structure to be implemented in a computationally efficient manner while providing for a means to reduce the effects of errors associated with discrete Fourier transforms.
• Some of these potential error sources in discrete Fourier transformers include: errors due to aliasing (i.e., errors resulting from overlapping of the decimated signal in the frequency domain), spectral distortion from leakage (i.e., spectral energy leaking from one frequency to another), and the picket-fence effect which results in important spectral components being missed (i.e., the discrete sampling omits particular spectral components of a signal which occurs between sample points).
• aliasing i.e., errors resulting from overlapping of the decimated signal in the frequency domain
• spectral distortion from leakage i.e., spectral energy leaking from one frequency to another
• the picket-fence effect which results in important spectral components being missed (i.e., the discrete sampling omits particular spectral components of a
• the errors due to aliasing may be reduced by increasing the sampling rate of the discrete Fourier transformer 148 and/or pre-filtering to minimize high-frequency spectral content of the inputs to the discrete Fourier transformer 148.
• the spectral distortion due to leakage may be reduced by increasing the number of DFT points through operating at a higher sampling rate or increasing the window width using window functions that have Fourier transforms with low sidelobes, and/or eliminating large periodic components by filtering before performing window functions.
• N post-filters 158, 160, 162, and 164 may be computationally simple (e.g., the filters may be Infinite Impulse Response (IIR) filters), if phase distortion is not critical.
• IIR Infinite Impulse Response
• these post-filters need only be implemented for that subset of desired channels.
• the N decimating post-filters 158, 160, 162, and 164 preferably are designed with respect to at least one signal characteristic including: optimum selectivity, controlled phase response, and controlled amplitude response.
• each conditioned output 166, 168, 170, and 172 may need to be convolutionally decoded, maximum likelihood sequence estimated, or vocoded to retrieve the voice or data present in the particular conditioned output.
• the pre-filters 132, 134, 136, and 138 receive their inputs from a commutator 130 and operate in parallel to form the inputs 140, 142, 144, and 146, respectively, for the DFT filter bank 148.
• pre-filters 132, 134, 136, and 138 incorporate a bank of Hogenauer's Cascaded Integrator Comb (CIC) filters.
• CIC Hogenauer's Cascaded Integrator Comb
• This type of filter is generally described in an article by E.B. Hogenauer, "An Economical Class of Digital Filters for Decimation and Interpolation", IEEE Trans. Acoust. Speech. Signal Processin g , vol. ASSP-20, No. 2, April, 1981 , pp. 155-162.
• the CIC filters have advantages over typical polyphase filters in that CIC filters consist of cascaded first order integrators and combs (i.e., differencers), which allows these filters to be implemented without multipliers.
• M-1 y (MNT) ⁇ exp(-j2 ⁇ f/M)w(MNT + ⁇ )
• modified CIC filter 138 receives a portion of the digitized signal 128 at an input.
• the portion of the digitized signal received by modified CIC filter 138 is determined by commutator 130 which operatively couples portions of the digitized signal 128 to each of the modified CIC filters 132, 134, 136, and 138 (i.e., f-i , f 2 , f3, and f ).
• This digitized signal portion 128/130 is input to a fifth order integrator 174 which integrates bits within received in the digitized signal portion 128/130 over an infinite time period to produce part of Z(M[NT]- ⁇ ) of (eq. 7).
• differencer 198 is shown in FIG. 8 consisting of five cascaded 1st order differencers.
• the output of decimator 186 is provided to summer 801 which subtracts the output of delay element 802 from the output of decimator 186.
• the summer output 803 is then provided to the next differencer stage, consisting of summer 804 and delay element 805, which subtracts the output of delay 805 from summer output 803 to produce summer output 806.
• This differencing operation is repeated three more time utilizing summers 807, 810, and 813 as well as delay elements 808, 811 , and 814 to produce the fifth order differencer 198 output 208.
• each of the other fifth order differencers 200, 202, 204, and 206 may be implemented in a manner similar to that which has been described in reference to FIG. 8 for differencer 198.
• the integrator 174 result 176 is input to differencer 200 after being delayed for one sample time by delay element 178 and subsequently decimated to a lower sample rate by a decimating device 188.
• a delayed and decimated version of result 176 is input to differencer 202 after being delayed for two sample times by delay elements 178 and 180 and decimated by a decimating device 190.
• a delayed and decimated version of result 176 is input to differencer 204 after being delayed for three sample times by delay elements 178, 180, and 182 and decimated by a decimating device 192.
• decimating and delay functions performed by elements 178, 180, 182, 184, 186, 188, 190, 192, and 194 alternatively may be implemented as a single decimating device coupled to integrator 174 result 176 followed by four delay elements without departing from the scope and spirit of the present invention.
• the scaled differencer outputs from multipliers 218, 220, 222, 224, and 226 preferably are then summed together by summer 228 such that the scaled differencer outputs based on the present sample plus four previous samples are summed together (i.e., a fifth order summation). Finally, the result from summer 228 is provided as the output 146 of the fifth order pre-filter 138.
• FIGs. 4-6 The principles described above in reference to a preferred embodiment receiver, as shown in FIGs. 1-3, may also be applied to a preferred embodiment transmitter in accordance with the present invention, as shown in FIGs. 4-6.
• the following description of the preferred embodiment transmitter refers to four channels (i.e., channels 1 , 2, 3, and N).
• channels 1 , 2, 3, and N the channels 1 , 2, 3, and N.
• the principles described herein may be readily applied to more or less transmitter channels without departing from the scope and spirit of the present invention.
• N inputs 300, 302, 304, and 306 may be input to pre- filters 308, 310, 312, and 314, respectively (i.e., pre-filters g-i , g 2 , g3, and gN).
• pre-filters 308, 310, 312, and 314 perform baseband filtering (i.e., condition) N inputs 300, 302, 304, and 306 with respect to at least one signal characteristic including: optimum selectivity, controlled phase response, and controlled amplitude response.
• the pre-filters 308, 310, 312, and 314 interpolate the N inputs 300, 302, 304, and 306 to increase the sampling rate such that an subsequent inverse discrete Fourier transformer 324 can operate at a rate higher that the of the N inputs 300, 302, 304, and 306 (e.g., the FFTs may operate at a rate which is three times greater than the filter sampling rate).
• This interpolation may be accomplished by zero- padding the N inputs 300, 302, 304, and 306 (e.g. inserting two zero value samples between each sample of the inputs).
• the pre-filters 308, 310, 312, and 314, like the receiver post-filters 158, 160, 162, and 164, preferably are computationally simple to implement.
• N outputs 326, 328, 330, and 332 are post-filtered by filters 334, 336, 338, and 340 (i.e., post-filters fi , f 2 , f3, and fN).
• These post filters operate similar to polyphase filters in known inverse discrete Fourier transform post-filter banks in that they receive their inputs from a inverse discrete Fourier transformer, and run in parallel to form the input to a commutator 350.
• One such polyphase filter bank is shown and described in the previously cited related U.S. Patent Application Serial No. 07/966,630 entitled "A Method and Means for Transmultiplexing Signals Between Signal Terminals and Radio Frequency Channels" which was filed on October 22, 1992.
• the fifth order filter design is used to implement each of the parallel modified CIC post-filters 334, 336, 338, and 340 (i.e., f-
• the modified fifth order CIC filter shown in FIG. 6 will be described as modified CIC filter 340 (f
• the preferred embodiment modified CIC filter 340 receives the
• the differencer 354 result 356 (i.e., differencer output) preferably is input to a fifth order integrator 362 after being upconverted to a higher sample rate (e.g., seven times the sample rate of the differencer 354 result 356) by an interpolating device 358. It will be appreciated by those skilled in the art that this interpolation of result 356 allows the subsequent parts of (eq. 8) to be implemented within post-filter 340 at a higher sampling rate such that the individual transmit channels can be further separated (i.e., separated by one or more other transmit channels).
• the fifth order integrator 362 which integrates bits within the interpolated differencer result 360 over an infinite time period to produce Z ⁇ (NT-k) of (eq. 8).
• the preferred embodiment fifth order integrator 362 was previously described in reference to FIG. 7. Subsequently, the output 364 of integrator 362 is scaled by a factor arj with multiplier 374. Similarly, the integrator 362 result 364 is scaled by a factor a-i with multiplier 376 after being delayed for one sample time by delay element 366. Further, a delayed version of result 364 is scaled by a factor a 2 with multiplier 378 after being delayed for two sample times by delay elements 366 and 368 .
• a delayed version of result 364 is scaled by a factor a3 with multiplier 380 after being delayed for three sample times by delay elements 366, 368, and 370.
• a delayed version of result 364 is scaled by a factor & 4 with multiplier 382 after being delayed for four sample times by delay elements 366, 368, 370, and 372 .
• the scaled integrator outputs from multipliers 374, 376, 378, 380, and 382 preferably are then summed together by summer 384 such that the scaled integrator outputs based on the present sample plus four previous samples are summed together (i.e., a fifth order summation). Finally, the result from summer 348 is provided as the output of the post-filter 340.
• the conditioned baseband electrical signal 360 is input to an up converter.
• This up converter consists of a mixer 364, a frequency agile local oscillator 362, and a band pass filter 368.
• the conditioned baseband electrical signal 360 is operatively coupled to one input of the mixer 364 and the output of the oscillator 362 is provided to another input of mixer 364.
• Mixer 364 mixes the baseband electrical signal 360 up to an IF band electrical signal 366. Subsequently, the IF band electrical signal 366 is filtered by a band pass filter 368 to eliminate undesired frequency components into a conditioned IF band electrical signal 370.
• the conditioned IF band electrical signal 370 is input to another up converter.
• the up converter consists of a mixer 374, a frequency agile local oscillator 372, and a band pass filter 378.
• the electrical signal 370 is operatively coupled to one input of the mixer 374 and the output of the oscillator 372 is provided to another input of mixer 374.
• Mixer 374 mixes the IF band electrical signal 370 up to a radio frequency (RF) band electrical signal 376. Subsequently, the RF band electrical signal 376 is filtered by a band pass filter 378 to eliminate undesired frequency components into a conditioned RF band electrical signal.
• RF radio frequency
• the conditioned RF band electrical signal preferably is output by a device such as antenna 380 which amplifies and radiates the conditioned RF band electrical signal as electromagnetic radiation within the frequency band of interest.
• a device such as antenna 380 which amplifies and radiates the conditioned RF band electrical signal as electromagnetic radiation within the frequency band of interest.
• other types of output devices may be used for transmitting or radiating electromagnetic radiation.
• a wave guide, a coaxial cable, an optical fiber, or an infrared frequency transducer may be used to transmit electromagnetic radiation.
• the output of the mixer 106 is a down converted electrical signal 110 having frequency components predominately near the predetermined frequency of the local oscillator 108.
• This electrical signal 110 may be conditioned to remove undesired frequency components by a bandpass filter 112.
• the filter 112 output i.e., the conditioned IF band electrical signal 112 may then be input to a second stage.
• a discrete Fourier transformer 148 operatively coupled to the commutator 130, generates a plurality of digitized channel signals 166, 168, 170, and 172 by discrete Fourier transforming the plurality of digitized signal portions.
• the discrete Fourier transformer 148 preferably includes signal conditioners 132, 134, 136, and 138 for pre-conditioning the plurality of digitized signal portions prior to the plurality of digitized signal portions being discrete Fourier transformed into the plurality of digitized channel signals 166, 168, 170, and 172.
• the signal conditioners 132, 134, 136, and 138 may be modified cascaded integrator comb (CIC) filters or polyphase filters as described above.
• the pre-conditioning performed by the signal conditioners 132, 134, 136, and 138 preferably consists of filtering each digitized signal portion with respect to a signal characteristic (e.g., optimum selectivity, controlled phase response, or controlled amplitude response).
• a signal characteristic e.g., optimum selectivity, controlled phase response, or controlled amplitude response
• Each information signal preferably conforms with a signal coding and channelization standard such as frequency division multiple access, time division multiple access, or frequency hopping code division multiple access.
• the inverse discrete Fourier transformer 324 preferably includes signal conditioners 334, 336, 338, and 340 for post-conditioning the plurality of digitized signals 342, 344, 346, and 348 after the plurality of input digitized information signals 300,
• the communication transmitting unit preferably includes a frequency selector mechanism 358, 362, 364, 368, 372, 374, and 378, operatively coupled to the analog-to-digital converter 354, for selecting a portion of a frequency band that the composite analog transmission signal 356 is to be transmitted.
• the composite analog transmission signal 356 preferably is provided to an input of a filter 358 that selects the portion 360 of the composite analog transmission signal 356 which is to be transmitted.
• This selected portion 360 of composite analog transmission signal 356 may be frequency translated to a different frequency band (e.g., an intermediate frequency (IF) band 366 or a radio frequency (RF) band 376) to facilitate transmission. This frequency translation may be accomplished through a one, two, or more stage process.
• IF intermediate frequency
• RF radio frequency
• the analog transmission signal 370 is up converted by a mixer 374 in conjunction with a local oscillator 372 and subsequently conditioned by a bandpass filter 378 to produce an RF band analog transmission signal.
• a transmitting mechanism 380 operatively coupled to the frequency selector mechanism 358, 362, 364, 368, 372, 374, and 378, transmits the RF band analog transmission signal over a frequency band.

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• Engineering & Computer Science (AREA)
• Computer Networks & Wireless Communication (AREA)
• Signal Processing (AREA)
• Noise Elimination (AREA)
• Radar Systems Or Details Thereof (AREA)
• Reduction Or Emphasis Of Bandwidth Of Signals (AREA)
• Measurement Of Velocity Or Position Using Acoustic Or Ultrasonic Waves (AREA)
• Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
• Mobile Radio Communication Systems (AREA)

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• 1993
  • 1993-03-08 US US08/027,981 patent/US5323391A/en not_active Expired - Lifetime
• 1994
  • 1994-02-03 SG SG1996003849A patent/SG46376A1/en unknown
  • 1994-02-03 WO PCT/US1994/001338 patent/WO1994021061A1/en not_active Application Discontinuation
  • 1994-02-03 JP JP6519982A patent/JPH07506711A/ja active Pending
  • 1994-02-03 EP EP94911379A patent/EP0643887A4/en not_active Withdrawn
  • 1994-02-03 KR KR1019940703975A patent/KR0153223B1/ko not_active Expired - Fee Related
  • 1994-02-03 CA CA002117920A patent/CA2117920C/en not_active Expired - Fee Related
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