WO1981003102A1 - Ballast pour la connexion d'une lampe a decharge - Google Patents

Ballast pour la connexion d'une lampe a decharge Download PDF

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Publication number
WO1981003102A1
WO1981003102A1 PCT/DE1981/000058 DE8100058W WO8103102A1 WO 1981003102 A1 WO1981003102 A1 WO 1981003102A1 DE 8100058 W DE8100058 W DE 8100058W WO 8103102 A1 WO8103102 A1 WO 8103102A1
Authority
WO
WIPO (PCT)
Prior art keywords
voltage
inverter
ballast
lamp
capacitor
Prior art date
Application number
PCT/DE1981/000058
Other languages
German (de)
English (en)
Inventor
R Priller
M Kerscher
A Nguyen
A Kroening
Original Assignee
Siemens Ag
M Kerscher
A Nguyen
A Kroening
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from DE3014419A external-priority patent/DE3014419C2/de
Priority claimed from DE19803029672 external-priority patent/DE3029672A1/de
Application filed by Siemens Ag, M Kerscher, A Nguyen, A Kroening filed Critical Siemens Ag
Priority to AT81900979T priority Critical patent/ATE8316T1/de
Publication of WO1981003102A1 publication Critical patent/WO1981003102A1/fr

Links

Classifications

    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/295—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices and specially adapted for lamps with preheating electrodes, e.g. for fluorescent lamps
    • H05B41/298—Arrangements for protecting lamps or circuits against abnormal operating conditions
    • H05B41/2981—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions
    • H05B41/2985—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions against abnormal lamp operating conditions

Definitions

  • the invention relates to a ballast for connecting a discharge lamp to an AC voltage network according to the preamble of claim 1.
  • Such a ballast is described in DE-OS 26 42 272: there, an essentially sinusoidal mains current is to be achieved by the two-point regulator connected upstream of the inverter.
  • a quantity derived from the rectified, unsmoothed AC line voltage is supplied to the two-point controller as a setpoint and the current of the discharge lamp as an actual value. This results in an approximately sinusoidal current load on the network.
  • the inverter is fed practically directly from the AC voltage network, the voltage at the discharge lamp is modulated with the frequency of the AC voltage, whereby the stability of the discharge in the lamp and also the luminous efficiency are impaired.
  • the invention has for its object to provide a ballast for operating a discharge lamp, with which on the one hand a stable discharge with high luminous efficacy can be achieved and which on the other hand ensures as sinusoidal a network load as possible with low self-losses and reasonable effort.
  • the Ent Charge lamp a relatively high-frequency supply voltage (about 40 kHz) with an essentially constant amplitude, so that a stable discharge and an optimal light output can be achieved.
  • the dimensioning of the components of the ballast is preferably selected so that on the one hand an ignition voltage sufficient even at the lowest temperatures Is available, on the other hand, the ballast provides the nominal power when the lamp is lit.
  • ballast is thus tailored to a very specific discharge lamp. Therefore, the voltage would reach impermissible values if the ballast was accidentally operated with a lamp of lower wattage.
  • a limiting device is provided with a zener diode and a transistor, by means of which the setpoint of the two-point regulator is reduced as soon as the voltage at the storage capacitor of the step-up converter reaches or exceeds a certain limit value. If the ballast is designed with several, in particular two, inverters for the operation of several, preferably two, lamps, then an OR circuit reduces the setpoint of the controller if only one of the inverters - e.g. because of non-igniting lamp - is switched off.
  • the ballast is preferably equipped with a monitoring part, which monitors the current through the Dros be the series resonant circuit of the inverter and by which the two-point controller and the inverter are switched off when the time integral of the current through the choke of the series resonant circuit a certain
  • the ballast is protected against overload, for example when a lamp with excessive power is connected.
  • the monitoring direction is designed in such a way that it also responds when the ballast is operated with the correct lamp, but it no longer ignites.
  • a response of the monitoring device during the ignition phase of a lamp with normal or extended ignition time is avoided by an integrating element with a defined discharge resistance.
  • the impedance of the capacitors at the average operating frequency of the inverter is dimensioned so that it is approximately equal to ten times the value of the resistance of an electrode.
  • each inverter is individually switched off at its output when the conditions described are present.
  • the common power supply of the inverters is only interrupted via an AND gate acting on the controller when all inverters are switched off.
  • the electrodes of the discharge lamp in series with the series resonant circuit of the inverter, on both sides of this capacitor. It is also advantageous to operate the inverter at a higher operating frequency for a sufficient time for preheating after switching on. As a result, the voltage at the series resonance circuit can be reduced to a value which is not sufficient to ignite the lamp even at high ambient temperatures. For this purpose, it is sufficient to short-circuit a small additional winding of the saturation transformer which determines the operating frequency during the desired preheating time.
  • the ballast has a starting device for the inverter, by which it is ensured that the inverter only starts to oscillate when there is sufficient voltage on the storage capacitor of the step-up converter: This prevents operation with reduced voltage which affects the life of the discharge lamp.
  • the reduction in losses that can be achieved in this way is based on the fact that a much greater voltage occurs at the respective emitting electrode than at the other electrode; this voltage is limited by the diode connected in parallel to its threshold value, which is considerably below the peak value of the voltage otherwise occurring at the emitting electrode.
  • Step-up converter flowing current on the storage capacitor occur at rates of change that result in interference voltages with frequencies in the megahertz range. Since the storage capacitor is generally an electrolytic capacitor and thus has a high resistance to these interference voltages, but these interference voltages cannot flow away via the charging choke, a correspondingly intense interference radiation occurs at the discharge lamp.
  • the storage capacitor and the parallel-connected inverter are connected to the main transistor of the step-up converter via a blocking choke, by means of which the commutation process is somewhat delayed; an inductance of the order of about 1 micro H is sufficient for this.
  • FIG. 1 shows a circuit diagram of the power supply unit, in which the inverter W connected to it and the control part X are shown as a block
  • FIG. 2 shows a circuit diagram of the inverter W
  • FIG. 3 shows a circuit diagram of the control part X, in addition in FIGS whose function particularly important circuit parts of other figures are shown with their terminals (1 to 16).
  • the storage capacitor C18 is connected on the one hand via a blocking inductor L1, the charging diode D27 and the charging inductor L4 on the other hand via a measuring resistor R33 to a main rectifier G1 in a two-way circuit, which is fed by an AC voltage network N and has an essentially at its terminals 1, 4 Unsmoothed voltage delivers: The lack of a Charging capacitor at this point leads to a more favorable course of the mains current.
  • the signal at the measuring resistor R33 between the main transistor V6 and the storage capacitor C18 on the one hand and the main rectifier G1 on the other hand is fed via a delay element (resistor R27 and capacitor C14) to the input 7 of the control part X, which consists of a monitoring part and a controller; the latter controls the main transistor V6 as a switch: when the main transistor V6 is switched on, the inductor L4 charges and delays the capacitor C14 until its voltage reaches the setpoint, which is fed to the controller of the control part X via the terminals 1 and 4 and the shape of an unsmoothed one rectified AC voltage:
  • the current drawn from the network thus has a sinusoidal profile on average.
  • the controller switches off the main transistor V6: the inductor L4 then transfers its energy, the charging diode D27 to the storage capacitor C18, the voltage of which in the meantime had dropped somewhat due to the load from the inverter W.
  • the signal at the measuring resistor R33 and delayed - the signal at the capacitor C14 decreases.
  • the delay element R27 / C14 is dimensioned so that the voltage at C14 reaches the lower switchover point of the controller when the current through the charging inductor L4 has become zero, that is, it has been completely magnetized back: When V6 is then switched on, this transi disturb therefore neither a residual current of the charging inductor L4 nor a reverse current of the diode D27, so that practically no switch-on losses occur.
  • the protective capacitor C17 is used to limit the voltage rise at the blocked main transistor V6 and is connected in parallel to the switching path of this transistor and the measuring resistor R33 via the decoupling diode D9 and a capacitor C8:
  • the protective capacitor C17 charges via L4, D9 and thereby delays the rise in the blocking voltage at the switching path of the main transistor V6, so that only a small power loss occurs.
  • the operating voltages for the control part X are supplied by the additional rectifier G2, which is connected to the AC voltage network N via capacitors C6, C7 and which feeds the partial capacitors C8 and C9 in series connection; the connection point of these two capacitors lies at the negative terminal 4 of the main rectifier G1, so that they supply a positive or a negative operating voltage.
  • the partial capacitor C8 is also connected to the main rectifier G1 via the decoupling diode D9, the protective capacitor C17 and the charging inductor L4, and thus receives an additional charge from the main rectifier when the main transistor V6 is blocked and C17 charges C17 is dimensioned as large at 300 pF, that this additional charge of C8 is sufficient for the subsequent control of the main transistor. Additional rectifiers and capacitors C6, C7 are therefore only dimensioned for the remaining power requirements of the controller and monitoring section.
  • the other partial capacitor C9 is connected via the protective inductor L9 and the protective capacitor C17 in parallel to the switching path of the main transistor V6 and the measuring resistor R33;
  • the discharge current of the protective capacitor C17 flows through this circuit and thus via C9 when the main transistor V6 is turned on, which in this case charges the partial capacitor C9.
  • both capacitors have approximately the same capacitance (approximately 50 ⁇ F).
  • the different voltages are set during operation by appropriate dimensioning of the protective choke L9: the latter is dimensioned about three times as large as would be necessary to delay the rise sufficiently.
  • the recharging of the partial capacitor C9 when the protective capacitor C17 is discharged via the main transistor V6 is namely significantly less than the charge of the partial capacitor C8 when the protective capacitor C17 is charged.
  • Such a coordination of the time constants is a prerequisite that a complete charge or discharge of C17 is guaranteed in the associated switching cycles of V6.
  • the energy stored in the protective choke L9 is finally discharged via the diode D9 and the two partial capacitors C8 and C9 in series connection. This discharge process is irrelevant for the voltage relation across the partial capacitors, since it affects both capacitors to the same extent.
  • the energy of the protective choke L9 is thus used to supply the electronics and does not burden the main transistor V6.
  • a swinging diode D42 is also connected in anti-parallel to the main transistor and the measuring resistor R33;
  • the "invisible" capacitances (the choke, the diodes, etc.) can discharge past the main transistor V6 via these.
  • the storage capacitor C18 is connected in parallel with the main transistor V6 via the charging diode D27 and a small blocking inductor L1 (approx. 1 ⁇ H): This reduces the voltage gradient at C18 and thus the ⁇ F interference voltage that would otherwise be emitted via the lamp L.
  • the inverter according to FIG 2 contains two transistors V7, V8 connected in series between its terminals 11, 8.
  • Parallel to the switching path of the transistor V7 is a series resonance circuit with the choke L7 and the capacitor C23 in series with a switching capacitor C22 and the primary winding L81 of a saturation transformer T8 with secondary windings L82, L83, L84 and L85.
  • the discharge lamp L is arranged in parallel with the capacitor C23, so that its electrodes l1, l2 are connected in series with the series resonant circuit.
  • the parallel circuit comprising a capacitor and a diode C26, D46 and C27, D47 is arranged in parallel with each electrode.
  • the diodes are polarized so that either the anodes or the cathodes of both diodes are located on the ends of the electrodes l1, l2 connected to the capacitor C23 of the series resonant circuit.
  • a control set St known per se is used, of which only the secondary windings L83, L84 of the saturation transformer T8 are shown: this results in an alternating charge of the reversing capacitor C22 via V8 from C18 and then a discharge via V7.
  • the operating frequency of the inverter determined by the saturation transformer is slightly above the resonance frequency of the series resonance circuit: This creates a gap between the reversal of V7 and V8.
  • a delayed start-up of the inverter is ensured with the help of a capacitor C19, which is only charged when the controller (V6) is working: If there is sufficient voltage at C19, V8 is controlled by a trigger diode D34 and C19 is discharged again (sufficient energy for cold start ).
  • the capacitor C19 is in turn connected in parallel via a capacitor C20 and a diode D32 to the switching path of the main transistor V6 and can therefore only be charged to a value sufficient for the inverter to oscillate when the controller is clocked properly.
  • the capacitor C20 is connected in parallel with the protective capacitor C17 via a resistor R40: the energy of C20 therefore serves, like that of C17, to charge the partial capacitors C8 and C9.
  • the saturation transformer T8 has two further secondary windings L82, L85; A current-dependent shutdown is effected via L82 and an increase in the operating frequency of the inverter is effected via L85 in the start-up phase (cf. FIG. 3).
  • the latter results in a lamp voltage which, even at high ambient temperatures, is not sufficient to ignite it, so that adequate lamp preheating is also ensured in this case.
  • the Short circuit of L85 is canceled at the end of the start-up phase: With * operating frequency, the lamp voltage then rises to a value that is sufficient for reliable ignition even at 0o C ambient temperature.
  • the heating power that is not required in continuous operation is disruptive.
  • this is significantly reduced by the diodes D46, D47: namely, the voltage at the emitting electrode is limited to the threshold value of the diode (approximately 1 volt) and thus the power consumption of the ignited lamp is significantly reduced.
  • the unlimited voltage of the emitting electrode is about six times greater than the threshold value of the diode, whereas the voltage of the non-emitting electrode is only about 2 volts anyway.
  • the control part X shown in FIG 3 consists of a controller (left of the dotted line) and a monitoring part; the controller is constructed as a two-point controller and has a comparator V13 at its heart, the output of which is connected via a resistor R25 to a terminal K to which a positive operating voltage can be switched. V13 controls a transistor V4, the collector of which is connected to the base of a further transistor V5, via which the control current of the main transistor V6 is then conducted.
  • V5 is also based on the tap of a voltage divider with resistors R31, R30, R24, R18 and R2, which is connected between the positive terminal K and the terminal 4 which is at zero potential and which supplies the setpoint for the comparator V13;
  • resistor R2 drops across the terminal 1 and the resistor R1 * a signal dependent on the rectified mains voltage is supplied.
  • the hysteresis of the two-position controller is determined by connecting the collector of transistor V4 to a tap on this voltage divider:
  • transistor V4 is turned on, taps the voltage divider at the negative input of the comparator via diode D22 to the negative terminal 3 and blocks the transistors V5 and V6 by negative potential at their bases.
  • the lower response limit value of the comparator V13 is thereby set to a small positive value via D23 and R30, which is practically independent of the course of the setpoint at 1.
  • the switch-on point of the main transistor V6 is independent of the course of the setpoint and can be set such that V6 only switches on when the current through the charging inductor L4 is zero.
  • the task of the monitoring part in FIG. 3 is to ensure adequate lamp preheating and to block the main transistor V6 with certainty under certain critical operating conditions.
  • the latter is achieved with the aid of a transistor V3, which interrupts the operating voltage of the regulator (except the end transistor V5).
  • Such blocking of the power supply is necessary as long as the partial capacitors C8 and C9 have not yet reached their operating voltage after switching on the device, since then no clear control of the main transistor V6 in switching operation is guaranteed;
  • the inverter is overloaded or idling, switch off and limit the voltage when operating with a lamp with insufficient power.
  • the transistor V3 is used to switch off the regulator, via which the positive operating voltage at terminal 2 can be switched to the regulator (terminal K).
  • This transistor receives its control current from a further transistor V2, the base of which is connected to a voltage divider R8, R9 connected between 2 and a diode D17 and the emitter of which is connected to a zener diode D13 which is connected between 2 and 4 via a resistor R3.
  • the Zener diode and voltage divider are dimensioned such that transistor V2 is only turned on when the voltage across the partial capacitor C8 has reached a minimum value required for operation. In this case, the potential the base of V2 is sufficiently larger than that at the emitter, which is determined by the Zener diode D13: this makes the transistors V2 and V3 conductive and the supply voltage for the controller is at K.
  • transistor V4 of the regulator is turned on via diode D17 and resistor R25 and thus V5 and V6 are blocked.
  • V3 remains on as long as there is negative potential at the output of the comparator V12 and transistor V17 is thereby blocked via diode D62.
  • This switching state characterizes normal operation, in which the potential at the output of comparator 12 is determined by the voltage at partial capacitor C9 (terminal 3), since in normal operation the voltage of zener diode D13 at the negative input is normally greater than that at the positive input.
  • the comparator V12 only switches over (positive potential at the output) when the voltage at its positive input becomes greater than the voltage at the Zener diode D13: Then V17 receives a control current via R60 and blocks the transistors V2, V3 and via this also V5 and the Main transistor V6.
  • transistor V10 which is blocked in normal operation, is turned on via R13, thereby short-circuiting winding L82 of transformer T8 via diode D41.
  • the transistors V7, V8 of the inverter no longer receive a control voltage and block.
  • the capacitor C11 is simultaneously discharged to such an extent that there is no immediate shutdown when the device is started up again.
  • the shutdown described is dependent on the voltage across the capacitors C11 and C10, which are discharged via R14 and charged via the diode D41 and resistor R41 with a voltage from the secondary winding L82 of the transformer T8 in the inverter
  • FIG 2 is supplied and which is proportional to the current through L7 (particularly high when the lamp is not ignited, the series resonance circuit is not damped).
  • V12 voltage divider. R12, D16 and R14
  • Zener diode D13 the negative input
  • Capacitors C10 and R14 are dimensioned in such a way that a quick succession of fewer start pulses - starting a new lamp - does not lead to switching off any more than a larger number of pulses with a greater distance (start of an old lamp).
  • capacitors C26 and C27 are provided in parallel to the electrodes 11, 12 and are dimensioned such that the current flowing when the lamp is missing is above the response limit value.
  • the reactance of the capacitors C26 and C27 is approximately ten times the resistance of an electrode at the average operating frequency of approximately 40 kHz.
  • the switching path of which is connected to resistor R2 is connected in parallel in the voltage divider at the input of the comparator V13 of the controller: Therefore, if the voltage at C18 exceeds a value specified by the dimensioning of the voltage divider R5, R6 and the Zener diode D12, V1 becomes conductive and the rest of the voltage supplied to the comparator via its negative input Setpoint dependent on the voltage at 1 more or less reduced. In this way, the voltage of the storage capacitor is limited to a predetermined value.
  • a switching device S is also controlled, of which only one transistor is shown, with the aid of which the secondary winding L85 of the saturation transformer T8 can be short-circuited; the switching path of this transistor lies between the DC terminals and the winding L85 between the AC terminals of a rectifier bridge (see FIG. 2).
  • the transistor V18 is turned on, for example, by a monostable multivibrator as soon as the inverter starts to work and a voltage occurs at C11.
  • the winding L85 is then short-circuited, so that the inverter oscillates at a higher frequency and the voltage on the lamp is thus reduced to a value which is insufficient for ignition.
  • the inverters are to be connected in parallel by connecting the terminals 4, 8, 11, 13 and 14 with the same names in FIG. 1.
  • the special circuit part to the right of the dash-dotted line in FIG. 3 is to be provided twice in this case, the terminals 4, 5 and 10 of this second circuit part having to be connected to the terminals of the same inverter with the same designation.
  • Terminals 15 and 16 of the second special circuit section are to be connected to terminals 15 'and 16' of the common monitoring section (on the left of the dash-dotted line):
  • this circuit ensures that each of the two inverters is switched off individually in accordance with the criteria explained above. After switching off one of the two inverters, however, the controller must be given a new setpoint in order to avoid an inadmissible increase in the voltage on the capacitor C18.
  • resistor R2 in the voltage divider at the input of comparator V13 is connected in parallel with a transistor V16, which is controlled by a comparator V15: The negative input of V15 is at R50 at a high positive potential, so that negative potential is at the output and V16 is therefore normally blocked is.
  • V15 switches over and controls V16 when one of the two inverters is switched off, because here the negative input of V15 via the activated transistor V10 is one of the two special circuit parts and via Diode D52 or diode D51 is set to zero potential (terminal 4).

Landscapes

  • Circuit Arrangements For Discharge Lamps (AREA)

Abstract

Les lampes a decharge souvent reliees au reseau de tension alternative par un onduleur et un redresseur, l'onduleur travaillant avec une frequence d'au moins 10kHz. Il est ainsi possible de maintenir l'inductance necessaire pour la limitation de courant faible et en meme temps d'obtenir un rendement d'eclairage eleve. Toutefois, avec ce systeme, l'allure du courant preleve du reseau s'ecarte tres fortement de la forme sinusoidale. Afin d'eviter cet inconvenient on a deja commande le prelevement de courant a l'aide d'un regulateur a deux positions recevant la demi-onde du reseau comme valeur de consigne et le courant de lampe comme valeur effective. Avec ce ballast on evite la modulation de la tension de lampe avec la tension de reseau qui en resulte.
PCT/DE1981/000058 1980-04-15 1981-04-14 Ballast pour la connexion d'une lampe a decharge WO1981003102A1 (fr)

Priority Applications (1)

Application Number Priority Date Filing Date Title
AT81900979T ATE8316T1 (de) 1980-04-15 1981-04-14 Vorschaltgeraet fuer eine entladungslampe.

Applications Claiming Priority (3)

Application Number Priority Date Filing Date Title
DE3014419A DE3014419C2 (de) 1980-04-15 1980-04-15 Vorschaltgerät für eine Entladungslampe
DE3014419 1980-04-15
DE19803029672 DE3029672A1 (de) 1980-08-05 1980-08-05 Vorschaltgeraet zum anschluss einer entladungslampe

Publications (1)

Publication Number Publication Date
WO1981003102A1 true WO1981003102A1 (fr) 1981-10-29

Family

ID=25784967

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/DE1981/000058 WO1981003102A1 (fr) 1980-04-15 1981-04-14 Ballast pour la connexion d'une lampe a decharge

Country Status (4)

Country Link
EP (1) EP0050131B1 (fr)
FI (1) FI69540C (fr)
IT (1) IT1137447B (fr)
WO (1) WO1981003102A1 (fr)

Cited By (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0092654A2 (fr) * 1981-04-14 1983-11-02 Siemens Aktiengesellschaft Appareil ballast
EP0116302A2 (fr) * 1983-01-19 1984-08-22 Siemens Aktiengesellschaft Convertisseur
EP0126556A1 (fr) * 1983-05-05 1984-11-28 Dubank Electronics (Pty) Limited Procédé pour mettre en oeuvre et démarrer une lampe à décharge, alimentation de puissance et ballast électronique y afférent
EP0085073B1 (fr) * 1981-07-31 1986-10-22 Siemens Aktiengesellschaft Convertisseur
WO1986006921A1 (fr) * 1985-05-14 1986-11-20 Zumtobel Aktiengesellschaft Circuit protecteur pour un circuit d'onduleur

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
DE3503778C2 (de) * 1985-02-05 1986-11-20 Trilux-Lenze Gmbh + Co Kg, 5760 Arnsberg Leuchtstofflampen-Vorschaltgerät

Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
FR2206541A1 (fr) * 1972-11-13 1974-06-07 Gen Electric
DE2461440A1 (de) * 1974-01-02 1975-07-10 Gen Electric Vorschaltanordnung zum elektronischen erzielen eines grossen leistungsfaktors
US3969652A (en) * 1974-01-04 1976-07-13 General Electric Company Electronic ballast for gaseous discharge lamps

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
FR2206541A1 (fr) * 1972-11-13 1974-06-07 Gen Electric
DE2461440A1 (de) * 1974-01-02 1975-07-10 Gen Electric Vorschaltanordnung zum elektronischen erzielen eines grossen leistungsfaktors
US3969652A (en) * 1974-01-04 1976-07-13 General Electric Company Electronic ballast for gaseous discharge lamps

Cited By (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0092654A2 (fr) * 1981-04-14 1983-11-02 Siemens Aktiengesellschaft Appareil ballast
EP0092654A3 (fr) * 1981-04-14 1984-04-18 Siemens Aktiengesellschaft Appareil ballast
EP0085073B1 (fr) * 1981-07-31 1986-10-22 Siemens Aktiengesellschaft Convertisseur
EP0116302A2 (fr) * 1983-01-19 1984-08-22 Siemens Aktiengesellschaft Convertisseur
EP0116302A3 (en) * 1983-01-19 1986-02-12 Siemens Aktiengesellschaft Converter
EP0126556A1 (fr) * 1983-05-05 1984-11-28 Dubank Electronics (Pty) Limited Procédé pour mettre en oeuvre et démarrer une lampe à décharge, alimentation de puissance et ballast électronique y afférent
WO1986006921A1 (fr) * 1985-05-14 1986-11-20 Zumtobel Aktiengesellschaft Circuit protecteur pour un circuit d'onduleur

Also Published As

Publication number Publication date
FI69540C (fi) 1986-02-10
IT8121036A0 (it) 1981-04-10
EP0050131B1 (fr) 1984-07-04
FI811179L (fi) 1981-10-16
FI69540B (fi) 1985-10-31
EP0050131A1 (fr) 1982-04-28
IT1137447B (it) 1986-09-10

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