US4524423A - Digital signal separation filters - Google Patents

Digital signal separation filters Download PDF

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Publication number
US4524423A
US4524423A US06/319,061 US31906181A US4524423A US 4524423 A US4524423 A US 4524423A US 31906181 A US31906181 A US 31906181A US 4524423 A US4524423 A US 4524423A
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United States
Prior art keywords
signal
signals
output
coupled
tap
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Expired - Lifetime
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US06/319,061
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English (en)
Inventor
Alfonse Acampora
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RCA Licensing Corp
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RCA Corp
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Assigned to RCA CORPORATION, A CORP. OF DE reassignment RCA CORPORATION, A CORP. OF DE ASSIGNMENT OF ASSIGNORS INTEREST. Assignors: ACAMPORA, ALFONSE
Priority to US06/319,061 priority Critical patent/US4524423A/en
Priority to PT75758A priority patent/PT75758B/pt
Priority to GB08230798A priority patent/GB2110496B/en
Priority to FI823708A priority patent/FI77130C/fi
Priority to ES516967A priority patent/ES516967A0/es
Priority to SE8206172A priority patent/SE453237B/sv
Priority to AU89892/82A priority patent/AU558853B2/en
Priority to CS827812A priority patent/CS258461B2/cs
Priority to SU823509653A priority patent/SU1313362A3/ru
Priority to BE0/209401A priority patent/BE894913A/fr
Priority to JP57194502A priority patent/JPS5887909A/ja
Priority to ZA828094A priority patent/ZA828094B/xx
Priority to CA000414902A priority patent/CA1173916A/en
Priority to FR8218610A priority patent/FR2516322B1/fr
Priority to KR8204996A priority patent/KR910004310B1/ko
Priority to DK494182A priority patent/DK162679C/da
Priority to NZ202398A priority patent/NZ202398A/en
Priority to DE3240906A priority patent/DE3240906C2/de
Priority to PL1982238883A priority patent/PL138112B1/pl
Priority to IT24106/82A priority patent/IT1205273B/it
Priority to NL8204299A priority patent/NL8204299A/nl
Priority to DD82244674A priority patent/DD206871A5/de
Priority to AT0406882A priority patent/AT389966B/de
Publication of US4524423A publication Critical patent/US4524423A/en
Application granted granted Critical
Assigned to RCA LICENSING CORPORATION, TWO INDEPENDENCE WAY, PRINCETON, NJ 08540, A CORP. OF DE reassignment RCA LICENSING CORPORATION, TWO INDEPENDENCE WAY, PRINCETON, NJ 08540, A CORP. OF DE ASSIGNMENT OF ASSIGNORS INTEREST. Assignors: RCA CORPORATION, A CORP. OF DE
Priority to HK735/89A priority patent/HK73589A/xx
Anticipated expiration legal-status Critical
Expired - Lifetime legal-status Critical Current

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
    • H04N9/00—Details of colour television systems
    • H04N9/64—Circuits for processing colour signals
    • H04N9/646—Circuits for processing colour signals for image enhancement, e.g. vertical detail restoration, cross-colour elimination, contour correction, chrominance trapping filters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H17/00—Networks using digital techniques
    • H03H17/02—Frequency selective networks
    • H03H17/06—Non-recursive filters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H7/00—Multiple-port networks comprising only passive electrical elements as network components

Definitions

  • This invention relates to digital signal separation filters, and, in particular, to digital filter networks which produce both a bandpassed digital signal and a lowpassed digital signal.
  • the chrominance signal extends down to about 1.5 MHz below the color subcarrier frequency of 3.58 MHz, and the vertical detail information is contained in the lower 1.0 MHz of the signal produced at the chrominance output of the comb filter.
  • the luminance and chrominance signal components of the output signal of the chrominance channel may be separated between the upper frequency of the vertical detail signal, 1.0 MHz, and the lower frequency of the chrominance signal of approximately 2.1 MHz.
  • a digital television receiver in which the luminance and chrominance signals are separated by a digital comb filter, it is likewise desirable to separate the vertical detail information from the chrominance information for recombination with the combed luminance signal when the video signal is comb filtered over the full band of video frequencies.
  • a digital finite impulse response (FIR) filter in which a plurality of weighted signal taps are symmetrically located in time about a weighted center tap. Weighted signals from the symmetrically located taps are summed at a first point in the filter, which sum is then combined with weighted signals from the center tap in one sense, that is, either additively or subtractively, to produce signals at a first output. The summed signals at the first point are also combined with weighted signals from the center tap in an opposite sense to produce signals at a second output. The two outputs will exhibit bandpass and lowpass filter response characteristics, with the outputs at which the respective responses are produced being determined by the respective senses of signal combination.
  • FIR digital finite impulse response
  • an output tap weighted FIR filter is constructed in accordance with the principles of the present invention to provide both a bandpassed output signal and a lowpassed output signal.
  • an input tap weighted FIR filter is arranged in accordance with the principles of the present invention to provide the filtered output signals.
  • FIG. 1 illustrates, in block diagram form, the baseband digital signal processing section of a television receiver, constructed in accordance with the principles of the present invention
  • FIG. 2 illustrates, in block diagram form, an output tapped FIR filter constructed in accordance with the principles of the present invention
  • FIG. 3 illustrates, in block diagram form, a more detailed representation of the FIR filter of FIG. 2;
  • FIG. 4 illustrates, in block diagram form, an input tap weighted FIR filter constructed in accordance with the principles of the present invention.
  • FIGS. 5, 6 and 7 illustrate response characteristic curves used to explain the operation of the embodiments of FIGS. 1 through 4.
  • Video signals are supplied from a source of video signals 10 which may comprise, for instance, a video detector in a television receiver.
  • the video signals are applied to an analog to digital converter 12, which converts the analog video signals to digital signals in the form of successive eight-bit words, for instance.
  • the digital signals are applied to the input of a digital comb filter 14, which separates the signals into separate luminance (Y) and chrominance (C) components.
  • the digital comb filter 14 may be constructed to operate as described in the article "Digital Television Image Enhancement" by John P. Rossi, 84 SMPTE at 545-51 (1974).
  • the separated Y signal is applied by way of a delay element 16 to an input of an adder 30.
  • the delay ⁇ of the delay element 16 is chosen to substantially match the delay encountered by a vertical detail signal as it is processed by a combination bandpass and lowpass filter network 20, which is coupled between the C output of the comb filter 14 and a second input of adder 30.
  • the adder 30 combines the vertical detail signal with the combed luminance signal to produce a restored luminance signal.
  • the restored Y signal is applied to one input of an adder 32, where a vertical peaking signal component is added to the restored Y signal.
  • the resulting peaked Y signal is applied to luminance signal processing circuitry 40, which may be controlled to modify the brightness and contrast of the luminance signal.
  • the processed luminance signal Y' at the output of processing circuitry 40 is applied to an input of a matrix 60.
  • the bandpass/lowpass filter network 20 constructed in accordance with the principles of the present invention, is coupled to the chrominance output C of the comb filter 14.
  • the bandpass/lowpass filter 20 produces a bandpassed chrominance signal at one output which is applied to the input of chrominance signal processing circuitry 50.
  • the bandpass/lowpass filter 20 also produces a lowpass filtered vertical detail signal at a second output, which is applied to the adder 30, and to the input of a nonlinear detail signal processor 34.
  • the nonlinear processor 34 exhibits a nonlinear transfer function as shown in FIG. 1 and described on pages 12-15 of the article "A CCD Comb Filter for Color TV Receiver Picture Enhancement" by D. H.
  • the nonlinear processor 34 operates to core low amplitude signals, to peak intermediate amplitude signals, and to pare, or attenuate, high amplitude signals.
  • the nonlinear processor 34 may comprise, for example, a random access memory (RAM), with the transfer function applied to the vertical detail information being a function of data stored in the RAM under control of a processor (not shown). New data may be stored in the RAM by processor control of a read/write control line 38 and RAM address lines 36 during inactive video intervals, such as the vertical blanking interval.
  • the processed vertical detail signal is applied to the adder 32 as a peaking signal.
  • the chrominance signal processing circuitry 50 may comprise a chroma peaker and a color mixture signal demodulator, as described in U.S. Pat. No. 4,415,918 entitled "DIGITAL COLOR TELEVISION SIGNAL DEMODULATOR", filed Aug. 31, 1981.
  • the chrominance signal processing circuitry 50 produces demodulated color mixture signals, such as color difference signals (B-Y) and (R-Y) or I and Q signals.
  • the color mixture signals are applied to the matrix 60.
  • the matrix 60 combines the color mixture and luminance signals to produce red, green and blue color signals, which are then converted to analog form by a digital to analog converter 54 for application to a television kinescope (not shown).
  • the bandpass/lowpass filter network 20 serves to separate the vertical detail information, contained in the low frequency portion of the signal produced by the chrominance comb filter, from the chrominance information contained in the high frequency portion of the signal produced by the chrominance comb filter.
  • the filter network 20 produces bandpass filtered chrominance signals, free of luminance signal components, for application to the chrominance signal processing circuitry.
  • the filter network 20 also produces, at a separate output, lowpass filtered vertical detail information for the luminance channel, free of chrominance signal residue which could create "dot crawl" on edges of the restored luminance signal.
  • FIG. 2 A bandpass/lowpass filter network, suitable for use in the arrangement of FIG. 1 and constructed in accordance with the principles of the present invention, is shown in FIG. 2.
  • the network of FIG. 2 comprises a digital FIR filter, including a tapped shift register 100, weighting function circuits 102-118, and an adder tree arrangement 120-140.
  • the broad arrows shown in the drawing figures represent parallel lines of digital information which couple digital words of a plurality of bits from one element to another.
  • combed chrominance signals in the form of, for example, eight-bit words, are applied to the first stage of shift register 100.
  • Each stage of the shift register 100 is capable of temporarily storing and transferring a word of the chrominance signal under control of a clock signal.
  • each stage of the shift register 100 numbered one through twenty-one can simultaneously hold eight bits.
  • the shift register 100 exhibits a delay from the input of the first stage to the output of the last stage which is a function of the number of stages and the frequency of the clock signal which shifts the signals through the register.
  • the arrangement of FIG. 2 therefore comprises a 21-order FIR filter with taps being coupled to the outputs of stages one, five, nine, eleven, thirteen, seventeen and twenty-one.
  • the weighting function circuits 102-118 are coupled to the output taps of shift register 100 and multiply the tapped signals by the fractional coefficients shown in the FIGURE.
  • the FIR filter exhibits an impulse response which is relatively concentrated and symmetrical about the tapped center stage eleven. Signals from stage eleven are weighted by one-half in this example by weighting function circuit 102 and the tap-weighted signals are then applied to an input of an adder 130, and to an input of a subtractor 140. Signals from stages nine and thirteen, both located two stages away from the center stage eleven, are weighted by the coefficient +(5/16) by weighting function circuits 104 and 114, and applied to inputs of adder 120.
  • adder 122 and 124 are coupled to inputs of an adder 126, which combines the applied signals and has an output coupled to an input of an adder 128.
  • Adder 128 combines signals produced by adder 126 with signals provided by adder 120, and has its output coupled to an input of adder 130.
  • Adder 130 combines the tap-weighted signals summed in adder 128 with the center tap-weighted signal, and exhibits a lowpass filter response characteristic at its output. Lowpass filtered vertical detail signals are thus produced at the output of adder 130.
  • the combined tap-weighted signals at the output of adder 128 are also applied to an input of subtractor 140, where they are subtractively combined with the weighted center tap signals.
  • Subtractor 140 thereby exhibits a bandpass filter response characteristic at its output, which is a complement of the response characteristic at the output of adder 130.
  • Bandpass filtered chrominance signals are produced at the output of subtractor 140.
  • a single filter is used to provide both lowpass filtering and bandpass filtering.
  • FIG. 3 A more detailed embodiment of the FIR filter of FIG. 2 is illustratively shown in FIG. 3. Since the weighting function coefficients of FIG. 2 all have denominators which are powers of two, the tapped signals may be weighted by a shift-and-add technique, as shown in FIG. 3, which obviates the need for coefficient multipliers. For example, because signals from shift register stages nine and thirteen are both weighted by the same coefficient value of (5/16), these two signals may be added in adder 120 before weighting, as shown in FIG. 3. If, as in this example, the tapped signals are each eight bits in length, the output of adder 120 will be a nine-bit word.
  • the nine-bit output of adder 120 is divided by sixteen, as indicated by clock 154, and by four, as indicated by block 156, in the coupling of adder 120 to the inputs of an adder 158.
  • the nine-bit output of adder 120 is divided by sixteen by coupling only the five most significant bits of the output to the low-order bit inputs of one input of adder 158, and is divided by four by coupling the seven most significant bits to the lower order bit inputs of a second input of adder 158.
  • Adder 158 adds these two words to produce an eight-bit output signal, which is the sum of (1/16) plus (1/4), or (5/16) of the values of the tapped signals. This is the desired weighting coefficient for signals tapped from stages nine and thirteen.
  • adder 122 In a similar manner, signals tapped from shift register stages five and seventeen are summed in adder 122, which produces a nine-bit output signal.
  • the output of adder 122 is divided by sixty-four and by sixteen in its coupling to the two inputs of an adder 160, as indicated by blocks 162 and 164.
  • Adder 160 produces a six-bit output signal which is weighted by (5/64) with respect to the tapped signals.
  • This output signal is inverted by an inverting circuit 170 and applied to an input of adder 126, together with a logical "1" carry-in bit.
  • the signal inversion and the carry-in bit perform a two's complementing of the output of adder 160, which effectively provides the minus sign for the weighting coefficient.
  • Adder 128 The output of adder 128 is coupled to one input of adder 130.
  • the seven most significant bits of the signal tapped from center stage eleven are coupled to the second input of adder 130, as connoted by block 152.
  • Adder 130 thereby exhibits a lowpass filter characteristic at its output, at which the vertical detail information of the input signal is produced.
  • the seven most significant bits of the signal tapped from center stage eleven are coupled to one input of an adder 180.
  • the output of adder 128 is coupled to a second input of adder 180 by an inverting circuit 172, together with a logical "1" carry-in bit.
  • the inversion of the output signal of adder 128 along with the carry-in bit provide a two's complementing of the output signal of adder 128. This causes the output signal of adder 128 to be subtracted from the weighted center tap signal in adder 180, causing adder 180 to exhibit a bandpass filter response characteristic at its output. Bandpass filtered chrominance information is thereby produced at its output.
  • FIG. 5 illustrates the response characteristic 200 exhibited at the output of adder 128, when the shift register 100 is clocked by a 14.32 MHz signal. This response characteristic 200 is seen to have substantially equal amplitude variations about a midpoint value of 0.00.
  • Response curve 210 thus defines a lowpass filter response from zero Hz. to a six dB point at about 1.8 MHz, and a highpass filter above approximately 5.2 MHz, with a stopband interposed in the intermediate frequency range.
  • the highpass portion of the response contains essentially no signal content in the television receiver.
  • the lowpass filter portion then defines a passband for lowpass filtered vertical detail infomation signals at the output of adder 130.
  • response characteristic 200 of FIG. 5 is essentially inverted about the median 0.00 value.
  • Response characteristic 220 is seen to define a bandpass filter characteristic with a passband between approximately 1.8 MHz and 5.2 MHz. Since the chrominance passband terminates at approximately 4.1 MHz, adder 180 (or subtractor 140) will pass the chrominance signal of a television receiver in the passband of approximately 1.8 to 4.1 MHz.
  • the bandpass/lowpass filter network of the present invention may be configured as an input tapped FIR filter, as shown in the arrangement of FIG. 4.
  • a twenty-stage shift register 302 is used in the FIR filter, with adders 320-328 inserted between four-stage segements of the shift register.
  • Weighted input signals from the combed chrominance signal applied to the filter input 300 are applied to the input of the first stage of the shift register 302 and to the intervening adders.
  • the shift register stages are clocked by a common clock signal.
  • the combed chrominance input signal is applied to the input of the first stage and to an input of an adder 330 by weighting function circuits 304 and 316, which weight the applied signals by +(1/64).
  • Adder 330 has a second input coupled to the output of the last shift register stage twenty.
  • the input signal is applied to an input of an adder 320, which is coupled between shift register stages four and five, by a weighting function circuit 306.
  • the input signal is applied to an input of an adder 328, which is coupled between shift register stages sixteen and seventeen, by a weighting function circuit 314.
  • Weighting function circuits 306 and 314 weight the input signals by a factor of -(5/64).
  • the input signal is weighted by a factor of +(5/16) by weighting function circuits 308 and 312, which are coupled to inputs of adders 322 and 326.
  • Adder 322 is coupled between shift register stages eight and nine, and adder 326 is coupled between shift register stages twelve and thirteen.
  • the impulse response of the FIR filter of FIG. 4 is concentrated around the junction of shift register stages ten and eleven, which is located equidistantly between the first and last stages. Weighted signal components are accumulated in the adders as they pass through the shift register and adders, with the response characteristic of FIG. 5 being exhibited at the output of adder 330. Signals at the output of adder 330 are applied to an input of an adder 340 for combination with signals provided by a shift register 360.
  • the shift register 360 receives input signals which have been weighted by one-half by weighting function circuit 310, and delays these signals by the same amount of delay time as exhibited by the FIR filter from its impulse response center at the output of stage ten to the output of adder 330. Signals at the output of shift register 360 thus correspond to the center tap weighted signals of the embodiments of FIGS. 2 and 3. Therefore, adder 340 will exhibit the lowpass filter response characteristic of FIG. 6 at its output, at which the vertical detail information signals are provided.
  • Signals developed at the output of adder 330 are two's complemented for subtractive combination with signals provided by a shift register 360 in an adder 350.
  • the output of adder 330 is coupled to an input of adder 350 by an inverting circuit 352, with the adder 350 also receiving a logical "1" carry-in bit.
  • Adder 350 thus exhibits the bandpass filter response characteristic of FIG. 7 at its output by reason of the sutractive combination of signals supplied by shift register 360 and adder 330.
  • the FIR filter embodiments of the present invention have complementary forms, which may be obtained by selective reversal of the signs of the weighting coefficients, and/or reversal of the senses of the signal combining elements which produce the output signals. For example, if the signs of the weighting function coefficients of circuits 104, 106, 108 and 114, 116, 118 of FIG. 2 are all reversed, adder 130 will exhibit the bandpass response characteristic and subtractor 140 will exhibit the lowpass response characteristic. If, in addition, the senses of adder 130 and subtractor 140 are changed so that adder 130 becomes a subtractor and subtractor 140 becomes an adder, the new adder 140 will exhibit a bandpass response and the new subtractor 130 will exhibit a lowpass response.
  • signals passed by the new subtractor 130 will now exhibit a phase reversal with respect to the input signals of the filter.
  • subtractor 140 will exhibit a bandpass response with a phase reversal of the passed signals with respect to the filter input signal.
  • the bandpass/lowpass filter networks of the present invention have application in television receivers in which the luminance and chrominance information is not separated by comb filtering. Luminance and chrominance signals can then be separated directly by the bandpass lowpass filter.
  • the values of the weighting function coefficients or the clock frequency can be adjusted to relocate the crossover frequencies (transition bands) of the output response characteristics at a higher frequency. In the NTSC television system, this crossover frequency would be approximately 3.2 MHz.
  • the lowpass filter output in such an arrangement would pass signals up to approximately 3.2 MHz, and the bandpass filter output would provide signals of frequencies from 3.2 MHz to the upper limit of the video frequency range.

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  • Engineering & Computer Science (AREA)
  • Physics & Mathematics (AREA)
  • Mathematical Physics (AREA)
  • Multimedia (AREA)
  • Signal Processing (AREA)
  • Computer Hardware Design (AREA)
  • Acoustics & Sound (AREA)
  • Processing Of Color Television Signals (AREA)
  • Filters That Use Time-Delay Elements (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
  • Picture Signal Circuits (AREA)
  • Complex Calculations (AREA)
  • Electrophonic Musical Instruments (AREA)
US06/319,061 1981-11-06 1981-11-06 Digital signal separation filters Expired - Lifetime US4524423A (en)

Priority Applications (24)

Application Number Priority Date Filing Date Title
US06/319,061 US4524423A (en) 1981-11-06 1981-11-06 Digital signal separation filters
PT75758A PT75758B (en) 1981-11-06 1982-10-27 Dual output digital filters
GB08230798A GB2110496B (en) 1981-11-06 1982-10-28 Dual output digital filter and television receiver including such a filter
FI823708A FI77130C (fi) 1981-11-06 1982-10-29 Digitalfilter.
ES516967A ES516967A0 (es) 1981-11-06 1982-10-29 Perfeccionamientos introducidos en un filtro digital sensible a una senal de entrada para producir senales de salida primera y segunda.
SE8206172A SE453237B (sv) 1981-11-06 1982-10-29 Digitalfilter
AU89892/82A AU558853B2 (en) 1981-11-06 1982-10-29 Digital filter
CS827812A CS258461B2 (en) 1981-11-06 1982-11-03 Digital filter
CA000414902A CA1173916A (en) 1981-11-06 1982-11-04 Digital signal separation filters
BE0/209401A BE894913A (fr) 1981-11-06 1982-11-04 Filtres numeriques a deux sorties
JP57194502A JPS5887909A (ja) 1981-11-06 1982-11-04 デジタル・フイルタ
ZA828094A ZA828094B (en) 1981-11-06 1982-11-04 Dual output digital filters
SU823509653A SU1313362A3 (ru) 1981-11-06 1982-11-04 Цифровой фильтр
IT24106/82A IT1205273B (it) 1981-11-06 1982-11-05 Filtri digitali a doppia uscita
NL8204299A NL8204299A (nl) 1981-11-06 1982-11-05 Digitaal filter.
DK494182A DK162679C (da) 1981-11-06 1982-11-05 Digitalt filterkredsloeb
NZ202398A NZ202398A (en) 1981-11-06 1982-11-05 Digital chroma filter for television receiver:delay line shift register with weighted taps
DE3240906A DE3240906C2 (de) 1981-11-06 1982-11-05 Digitalfilter
PL1982238883A PL138112B1 (en) 1981-11-06 1982-11-05 Digital filter having two outputs
FR8218610A FR2516322B1 (fr) 1981-11-06 1982-11-05 Filtres numeriques a
KR8204996A KR910004310B1 (ko) 1981-11-06 1982-11-05 디지털 필터
DD82244674A DD206871A5 (de) 1981-11-06 1982-11-08 Digitale filteranordnung mit zwei ausgaengen
AT0406882A AT389966B (de) 1981-11-06 1982-11-08 Digitales filter
HK735/89A HK73589A (en) 1981-11-06 1989-09-14 Dual output digital filter and television receiver including such a filter

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Application Number Priority Date Filing Date Title
US06/319,061 US4524423A (en) 1981-11-06 1981-11-06 Digital signal separation filters

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US4524423A true US4524423A (en) 1985-06-18

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US06/319,061 Expired - Lifetime US4524423A (en) 1981-11-06 1981-11-06 Digital signal separation filters

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US (1) US4524423A (ru)
JP (1) JPS5887909A (ru)
KR (1) KR910004310B1 (ru)
AT (1) AT389966B (ru)
AU (1) AU558853B2 (ru)
BE (1) BE894913A (ru)
CA (1) CA1173916A (ru)
CS (1) CS258461B2 (ru)
DD (1) DD206871A5 (ru)
DE (1) DE3240906C2 (ru)
DK (1) DK162679C (ru)
ES (1) ES516967A0 (ru)
FI (1) FI77130C (ru)
FR (1) FR2516322B1 (ru)
GB (1) GB2110496B (ru)
HK (1) HK73589A (ru)
IT (1) IT1205273B (ru)
NL (1) NL8204299A (ru)
NZ (1) NZ202398A (ru)
PL (1) PL138112B1 (ru)
PT (1) PT75758B (ru)
SE (1) SE453237B (ru)
SU (1) SU1313362A3 (ru)
ZA (1) ZA828094B (ru)

Cited By (19)

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Publication number Priority date Publication date Assignee Title
US4615026A (en) * 1984-01-20 1986-09-30 Rca Corporation Digital FIR filters with enhanced tap weight resolution
US4626895A (en) * 1984-08-09 1986-12-02 Rca Corporation Sampled data video signal chrominance/luminance separation system
US4644389A (en) * 1984-02-29 1987-02-17 Kabushiki Kaisha Toshiba Digital television signal processing circuit
US4700345A (en) * 1983-06-03 1987-10-13 Independent Broadcasting Authority Downsampling and prefilter implementation in television systems
US4701874A (en) * 1983-04-06 1987-10-20 Nec Corporation Digital signal processing apparatus
US4782458A (en) * 1986-12-18 1988-11-01 North American Philips Corporation Architecture for power of two coefficient FIR filter
US4786963A (en) * 1987-06-26 1988-11-22 Rca Licensing Corporation Adaptive Y/C separation apparatus for TV signals
US4809209A (en) * 1985-08-26 1989-02-28 Rockwell International Corporation Mybrid charge-transfer-device filter structure
US4816830A (en) * 1987-09-14 1989-03-28 Cooper James C Waveform shaping apparatus and method
US4829367A (en) * 1987-08-28 1989-05-09 Institut National De La Recherche Scientifique Apparatus and method for encoding and decoding a NTSC color video signal
WO1991001526A1 (en) * 1989-07-25 1991-02-07 At&E Corporation Digital filter and method of design
US5130942A (en) * 1988-02-24 1992-07-14 Canon Kabushiki Kaisha Digital filter with front stage division
US5150413A (en) * 1984-03-23 1992-09-22 Ricoh Company, Ltd. Extraction of phonemic information
US5260888A (en) * 1992-05-28 1993-11-09 Eastman Kodak Company Shift and add digital signal processor
US5438532A (en) * 1993-01-20 1995-08-01 Sanyo Electric Co., Ltd. Digital filter for use in synthesizing filter or a separation filter
US6298366B1 (en) * 1998-02-04 2001-10-02 Texas Instruments Incorporated Reconfigurable multiply-accumulate hardware co-processor unit
US20050174492A1 (en) * 2004-02-05 2005-08-11 Brad Delanghe Method and system for data compression for storage of 3D comb filter data
US7123652B1 (en) * 1999-02-24 2006-10-17 Thomson Licensing S.A. Sampled data digital filtering system
US9992573B1 (en) 2013-10-29 2018-06-05 Meyer Sound Laboratories, Incorporated Phase inversion filter for correcting low frequency phase distortion in a loudspeaker system

Families Citing this family (8)

* Cited by examiner, † Cited by third party
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GB2110044A (en) * 1981-11-06 1983-06-08 Rca Corp Digital signal separation network and television receiver including such a network
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JPS60112309A (ja) * 1983-11-24 1985-06-18 Hitachi Ltd 信号処理用フィルタ
JPS60119116A (ja) * 1983-11-30 1985-06-26 Fujitsu Ltd ２次元積和演算装置
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JPH0732352B2 (ja) * 1985-11-20 1995-04-10 株式会社東芝 デジタルフイルタ
JP5428481B2 (ja) * 2009-04-15 2014-02-26 株式会社Ｊｖｃケンウッド 帯域分割フィルターおよびプログラム

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US4701874A (en) * 1983-04-06 1987-10-20 Nec Corporation Digital signal processing apparatus
US4700345A (en) * 1983-06-03 1987-10-13 Independent Broadcasting Authority Downsampling and prefilter implementation in television systems
US4615026A (en) * 1984-01-20 1986-09-30 Rca Corporation Digital FIR filters with enhanced tap weight resolution
US4644389A (en) * 1984-02-29 1987-02-17 Kabushiki Kaisha Toshiba Digital television signal processing circuit
US5150413A (en) * 1984-03-23 1992-09-22 Ricoh Company, Ltd. Extraction of phonemic information
US4626895A (en) * 1984-08-09 1986-12-02 Rca Corporation Sampled data video signal chrominance/luminance separation system
US4809209A (en) * 1985-08-26 1989-02-28 Rockwell International Corporation Mybrid charge-transfer-device filter structure
US4782458A (en) * 1986-12-18 1988-11-01 North American Philips Corporation Architecture for power of two coefficient FIR filter
US4786963A (en) * 1987-06-26 1988-11-22 Rca Licensing Corporation Adaptive Y/C separation apparatus for TV signals
US4829367A (en) * 1987-08-28 1989-05-09 Institut National De La Recherche Scientifique Apparatus and method for encoding and decoding a NTSC color video signal
US4816830A (en) * 1987-09-14 1989-03-28 Cooper James C Waveform shaping apparatus and method
US5130942A (en) * 1988-02-24 1992-07-14 Canon Kabushiki Kaisha Digital filter with front stage division
WO1991001526A1 (en) * 1989-07-25 1991-02-07 At&E Corporation Digital filter and method of design
US5404322A (en) * 1989-07-25 1995-04-04 Seiko Corp. Digital filter and method of design
US5260888A (en) * 1992-05-28 1993-11-09 Eastman Kodak Company Shift and add digital signal processor
US5438532A (en) * 1993-01-20 1995-08-01 Sanyo Electric Co., Ltd. Digital filter for use in synthesizing filter or a separation filter
US6298366B1 (en) * 1998-02-04 2001-10-02 Texas Instruments Incorporated Reconfigurable multiply-accumulate hardware co-processor unit
US7123652B1 (en) * 1999-02-24 2006-10-17 Thomson Licensing S.A. Sampled data digital filtering system
US20050174492A1 (en) * 2004-02-05 2005-08-11 Brad Delanghe Method and system for data compression for storage of 3D comb filter data
US7492415B2 (en) * 2004-02-05 2009-02-17 Broadcom Corporation Method and system for data compression for storage of 3D comb filter data
US9992573B1 (en) 2013-10-29 2018-06-05 Meyer Sound Laboratories, Incorporated Phase inversion filter for correcting low frequency phase distortion in a loudspeaker system

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DE3240906C2 (de) 1994-02-03
SE8206172L (sv) 1983-05-07
FI823708L (fi) 1983-05-07
PT75758B (en) 1985-12-09
IT1205273B (it) 1989-03-15
DK162679B (da) 1991-11-25
GB2110496A (en) 1983-06-15
KR840002795A (ko) 1984-07-16
JPH0342527B2 (ru) 1991-06-27
BE894913A (fr) 1983-03-01
ATA406882A (de) 1989-07-15
KR910004310B1 (ko) 1991-06-25
AT389966B (de) 1990-02-26
CA1173916A (en) 1984-09-04
ES8308661A1 (es) 1983-09-16
ES516967A0 (es) 1983-09-16
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JPS5887909A (ja) 1983-05-25
IT8224106A0 (it) 1982-11-05
PL238883A1 (en) 1983-05-23
SE8206172D0 (sv) 1982-10-29
NL8204299A (nl) 1983-06-01
SE453237B (sv) 1988-01-18
ZA828094B (en) 1983-09-28
SU1313362A3 (ru) 1987-05-23
FI77130B (fi) 1988-09-30
FR2516322B1 (fr) 1987-10-23
NZ202398A (en) 1986-01-24
DE3240906A1 (de) 1983-05-19
CS258461B2 (en) 1988-08-16
AU8989282A (en) 1983-05-12
GB2110496B (en) 1985-10-23
HK73589A (en) 1989-09-22
PT75758A (en) 1982-11-01
FI823708A0 (fi) 1982-10-29
FR2516322A1 (fr) 1983-05-13
FI77130C (fi) 1989-01-10
CS781282A2 (en) 1988-01-15
AU558853B2 (en) 1987-02-12

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