TWI568166B - A High Efficiency LLC Resonant Converter with Secondary Side Synchronous Rectifier Blind Control - Google Patents

Description

High-efficiency LLC resonant converter for blind-time regulation of secondary side synchronous rectifier

The present invention relates to LLC resonant converters, and more particularly to a high efficiency LLC resonant converter for blind time regulation of a secondary side synchronous rectifier.

Today's switching power supplies have been widely used in different electrical equipment. In order to meet the requirements of high efficiency, light and short, high power density (High Power Density), for DC-DC converters, the switching frequency can be increased to achieve high power density, but to reduce hard switching (Hard Switching) Switching losses to improve the overall efficiency of the converter, usually using Soft Switching technology, has the advantage of reducing the switching loss caused by the product of the voltage across the switch and the current when switching the power switch, achieving zero voltage switching (Zero Voltage Switching, ZVS) or Zero Current Switching (ZCS), which improves the efficiency of the power converter.

The Series Resonant Converter (SRC) has the characteristics of high switching frequency and zero voltage switching. When the series resonant converter operates in different operating ranges, it will have different circuit characteristics and derive the LLC resonant converter. LLC resonant converter has the advantages of simple circuit structure, wide input voltage, high power density and high efficiency. Therefore, LLC resonant converter has been widely used in power supplies such as LCD TVs and personal computers. On the other hand, because the digital power supply can be controlled by the digital control interface, the power management and monitoring can be accurately performed. The biggest difference between the digital power supply and the analog power supply is the feedback design. The analog power supply is used. The traditional analog circuit is controlled by feedback, and the processor needs to be additionally monitored. The digital power supply is programmable and not susceptible to temperature, and the function integration is higher than the analogy. Therefore, the digital power supply is regarded as the power supply design. An important development trend is the increasing demand for digital power supplies in the market.

In the past, there have been many literatures discussing the basic architecture of LLC resonant converters and analyzing their operating principles. In terms of efficiency, in order to improve the efficiency of the overall power converter, it is proposed in the literature that synchronous rectification technology is applied to LLC resonant converters, and different The driving method improves the efficiency of the LLC resonant converter. It has also been proposed in the literature to describe synchronous rectification by steady-state analysis, which can reduce the loss of output rectification and control the primary and secondary side switches with a voltage clamp drive circuit. Finally, the synchronous rectification LLC resonant converter is implemented and operated with an asynchronous LLC resonant converter. The efficiency comparison shows that the synchronous rectification efficiency is indeed better than the non-synchronization. With the advancement of semiconductor technology and microprocessor technology, digital power supply can be controlled and designed through digital control interface. Many literatures propose different digital control strategies to drive LLC resonant converters, and some control methods for LLC resonant converters. Explain that another paper discusses the impact of the microprocessor design part of the microprocessor on efficiency. LLC resonant converter is also widely used in computer power supply, LCD power supply and vehicle charger. It has been designed and implemented as a digital control LLC resonant converter as a charger. It also introduces the mode analysis and control strategy of LLC resonant converter. .

The main object of the present invention is to provide a high-efficiency LLC resonant converter for blind-time regulation of a secondary-side synchronous rectifier, which can improve the power conversion efficiency by adjusting the blind time interval of the secondary side.

In order to achieve the above object, a high-efficiency LLC resonant converter for blind-time regulation of a secondary side synchronous rectifier is proposed, which has:

a first half bridge switching circuit having two input ends coupled to the positive and negative terminals of an input voltage, two control terminals coupled to a first driving signal and a second driving signal, and an output terminal When the first driving signal is coupled to the positive terminal and the second driving signal exhibits an active potential, the first driving signal is coupled to the negative terminal;

a capacitor-inductor series circuit having one end coupled to the output end of the first half bridge switch circuit;

a transformer having a main coil and a primary coil, one end of the main coil being coupled to the other end of the capacitor-inductor series circuit, the other end of the main coil being coupled to the negative terminal of the input voltage, The secondary coil has a first output end, a second output end, and a center tap contact;

a second half-bridge switching circuit having two input ends coupled to the first output end and the second output end, and two control ends coupled to a third driving signal and a fourth driving signal, respectively The output end is coupled to the second output end when the third driving signal is coupled to the first output terminal when the third driving signal exhibits an active potential; and the fourth driving signal exhibits an active potential;

An output capacitor coupled between the output end of the second half bridge switch circuit and the center tap contact;

a load resistor coupled between the output end of the second half bridge switch circuit and the center tap contact;

a feedback circuit for generating a feedback signal according to one of the load resistors; and

a control unit configured to perform a driving signal generating program to generate the first driving signal, the second driving signal, the third driving signal, and the fourth driving signal according to the feedback signal, wherein the first driving There is a fixed blind time interval between the signal and the second driving signal, the third driving signal has a delayed on-time and an early cut-off time with respect to the first driving signal, and the fourth driving signal is opposite to the second driving signal. The delay on-time and the advance-off time are available, and the drive signal generation program includes a proportional-integral-derivative operation to adjust the delay on-time and the advance-off time.

In an embodiment, the feedback circuit includes a voltage dividing circuit and an optical coupling circuit.

In one embodiment, the drive signal generation program includes an analog to digital conversion operation.

In an embodiment, the drive signal generating program further includes a filtering operation.

In one embodiment, the control unit includes a pulse width modulation module to provide the first driving signal, the second driving signal, the third driving signal, and the fourth driving signal.

Referring to FIG. 1, an embodiment of a high efficiency LLC resonant converter for blind time regulation of a secondary side synchronous rectifier of the present invention is illustrated. As shown in FIG. 1 , the LLC resonant converter has a first half bridge switching circuit 100 , a capacitor-inductor series circuit 110 , a transformer 120 , a second half bridge switching circuit 130 , an output capacitor 140 , and a load resistor . 150, a feedback circuit 160, and a control unit 170.

The first half bridge switching circuit 100 has two input terminals A, B coupled to the positive and negative terminals of an input voltage V _IN , and two control terminals respectively coupled to a first driving signal S ₁ and a second driving signal S _{2 .} The output terminal C is coupled to the negative terminal when the first driving signal S ₁ is coupled to the positive terminal and the second driving signal S ₂ exhibits an active potential.

One end of the capacitor-inductor series circuit 110 is coupled to the output terminal C of the first half bridge switch circuit 100.

The transformer 120 has a main coil and a primary coil. One end of the main coil is coupled to the other end of the capacitor-inductor series circuit 110. The other end of the main coil is coupled to the negative terminal of the input voltage V _IN . The secondary coil has a first output terminal D, a second output terminal E, and a center tap contact F.

The second half-bridge switching circuit 130 has two input ends for coupling with the first output terminal D and the second output terminal E, and two control terminals respectively for a third driving signal S ₃ and a fourth driving signal S _{4 .} And the output terminal O is coupled to the first output terminal D when the third driving signal S ₃ exhibits an active potential, and the fourth driving signal S ₄ exhibits an active potential and the second output terminal E coupled.

The output capacitor 140 is coupled between the output terminal O of the second half bridge switch circuit 130 and the center tap contact F.

The load resistor 150 is coupled between the output terminal O of the second half bridge switch circuit and the center tap contact F.

The feedback circuit 160 includes a voltage dividing circuit 161 and an optical coupling circuit 162 for generating a feedback signal V _FB according to one of the load resistors 150 across the voltage V _O .

The control unit 170 stores a firmware program for executing a driving signal generating program according to the feedback signal V _FB to generate the first driving signal S ₁ , the second driving signal S ₂ , and the third driving signal S _{3 .} And the fourth driving signal S ₄ , wherein the first driving signal S ₁ and the second driving signal S ₂ have a fixed blind time interval, and the third driving signal S _{3 is} opposite to the first driving signal. S ₁ has a delay on-time and an early cut-off time, the fourth driving signal S ₄ with respect to the second driving signal S ₂ has the delay turn-on time and the early cut-off time, and the drive signal generator comprises a proportional - integral a differential operation to adjust the delayed on-time and the advance-off time. That is, the high potential period of the third driving signal S ₃ is adaptively adjusted within the high potential period of the first driving signal S _{1 and} the high potential period of the fourth driving signal S ₄ is in the The second drive signal S ₂ is adaptively adjusted within the high potential period.

The control unit 170 includes an analog-to-digital conversion unit 171, a filter operation unit 172, a proportional-integral-derivative operation unit 173, a pulse width modulation operation unit 174, and a drive unit 175.

The analog to digital conversion unit 171 is configured to perform an analog-to-digital conversion operation on the feedback signal V _FB ; the filtering operation unit 172 is configured to perform a filtering operation on the output of the analog to digital conversion unit 171; proportional-integral-derivative operation The unit 173 is configured to adjust the delayed on-time and the advance-off time to provide the first control signal S ₁ , the second control signal S ₂ , and the third control signal S via the pulse width modulation operation unit 174 and the driving unit 175 . ₃ and the fourth control signal S ₄ .

Accordingly, the present invention can greatly improve power conversion efficiency in applications of low voltage and high current.

The principle of the invention will be described in detail below.

LLC resonant converter architecture

Figure 2 shows the architecture of a LLC resonant converter with a diode on the secondary side. In the circuit architecture, the resonant tank consists of a resonant inductor L _r , a resonant capacitor C _r and a magnetizing inductor L _m , where the resonant inductor and the resonant capacitor The combination produces a higher resonant frequency, and the combination of the magnetizing inductance, resonant inductor and resonant capacitor produces a lower resonant frequency.

Fundamental wave approximation frequency response analysis

In order to explore the LLC series resonant converter circuit and simplify the analysis, the following uses the basic harmonic approximation (FHA) and converts the nonlinear circuit of Fig. 2 into the linear bifurcation model of Fig. 3 for analysis to understand its circuit. Frequency response.

The equations of each parameter in the circuit diagram and its derivation process are as follows:

The voltage input to the resonant tank:

Fourier series analysis

(1)

among them , ,

If ,then

(2)

among them Switch the frequency for the system.

The fundamental wave instantaneous value, effective value and average value of the voltage input to the resonant tank:

(3)

(4)

(5)

The fundamental wave instantaneous value, effective value and average value of the current input to the resonant tank:

(6)

(7)

(8)

among them The phase difference between the voltage and current input to the resonant network.

Resonant tank output voltage:

(9)

among them The voltage and current phase angle difference for the resonant network output.

The fundamental wave instantaneous value, effective value and average value of the output voltage of the resonant tank:

(10)

(11)

(12)

The fundamental wave instantaneous value, effective value and average value of the output current of the resonant tank:

(13)

(14)

(15)

among them For output power, For the output load.

4 is an equivalent circuit diagram of the LLC resonant tank of FIG. 2, including an equivalent resistance reflected from the secondary side to the primary side. Assuming that the secondary winding voltage does not contain harmonic components, the AC equivalent resistance can be obtained. for:

(16)

(17)

Transfer function Input impedance :

(18)

(19)

Substituting the above equation, the voltage gain of the circuit and the input impedance of the resonant tank can be obtained.

(20)

(twenty one)

The parameters are defined as follows:

First resonant frequency:

Characteristic impedance:

Resonant inductance ratio:

Normalized frequency:

Quality factor:

Figure 5 is a graph showing the voltage gain and normalized frequency response of the LLC resonant circuit at different Q values. It can be observed from the figure that the LLC resonant circuit has two resonant frequencies, which are as follows:

among them

As shown in FIG. 5, when the circuit operating frequency is in the region 1 or the region 2, the switch has a zero voltage switching characteristic when turned on; and the region 3 is a region separated by the first resonant frequency and the second resonant frequency, if the operation is performed. In this zone, the switch has a zero current switching characteristic when it is turned off. The circuit characteristics of the operation of the area 1, the area 2, and the area 3 will be separately described below.

When the circuit switching frequency is greater than the first resonant frequency At this time, the converter operates in the region 1, in which the energy is transmitted in the transformer, so that the output voltage is mapped back to the primary side to clamp the magnetizing inductance L _m through the transformer, so L _m cannot participate in the resonance in the operating region, and the resonant frequency The resonant inductor L _r is determined by the resonant capacitor C _r ; and the half-bridge series resonant converter operates in a mode similar to a series resonant converter. When the switching frequency changes, the impedance of the resonant tank also changes. The equivalent load mapped back to the primary side is connected in series with the resonant inductor L _r and the resonant capacitor C _r , so the voltage from the output is mapped back to the equivalent load on the primary side of the transformer. The input voltage of the resonant tank is in a divided voltage relationship, so when the converter operates in the region 1 and the switching frequency is equal to the first resonant frequency, the maximum voltage gain that the resonant tank can provide is one. In this interval, the input current of the resonant tank is behind the input voltage, and the input impedance is inductive.

When the circuit switching frequency is between the first resonant frequency and the second resonant frequency At this time, the converter operates in the region 2 , in which the resonant inductor current is equal to the magnetizing inductor current to decouple the transformer before the power switch is turned off, and the secondary side rectifying diode also has no current flowing. The zero current cutoff is reached, and the output voltage is no longer clamped to the magnetizing inductance L _m , so that L _m participates in the resonance, and the resonant frequency will be determined by the resonant capacitor C _r , the resonant inductor L _r , and the magnetizing inductance L _m . In this region, the voltage gain is greater than 1, and the voltage gain can be greatly changed with a small frequency variation.

When the switching frequency is lower than the second resonant frequency The converter operates in zone 3, in which the input current of the resonant tank leads the input voltage and the input impedance is capacitive.

In general, when the half-bridge resonant converter operates in the area 1 or the area 2, the switching switch of the upper and lower bridges of the circuit has the advantage of zero voltage switching, and this characteristic has an advantage in the application architecture of inputting high voltage and low current. In this case, the condition that the half-bridge resonant converter operates in the region 1 is called SRC, and the operation in the region 2 is called LLC-SRC. Next, the circuit operation principle of each stage of the SRC and the LLC-SRC will be introduced.

Circuit operation mode analysis

3.1 SRC circuit operation mode analysis (Zone 1)

SRC is the state in which the LLC resonant converter operates in the first interval. Figure 6 shows the main current and voltage signals of the LLC resonant converter of Figure 2, where I _Lr is the resonant inductor current, I _Lm is the magnetizing inductor current, and V _Cr is the resonance. The capacitor voltage and V _Lm are the magnetizing inductance voltage. FIG. 7 is an operation timing chart of the SRC, in which a total of ten stages can be divided in one switching period. Since the t ₀ to t ₅ circuit action is similar to t ₅ to t ₁₀ , this case will only analyze the t ₀ to t ₅ circuit actions. The following will analyze and explain each mode:

SRC-Mode 1 ( ):

Wherein, the upper bridge switch (driven by S ₁ ) is turned on at t ₀ , the lower bridge switch (driven by S ₂ ) is turned off, and the resonant inductor current I _{Lr is} changed from the D _B1 path to the upper bridge switch due to the continuous flow characteristic of the dimension. itself flow to V _in, this time the energy via a transformer, the secondary side rectifier diode D ₁ is transmitted to the load terminal and the output capacitor is charged, the voltage across the magnetizing inductance L _m is mapped back by the output voltage of a primary side immobilized by L _m A dynamic short circuit is formed, so the magnetizing inductance voltage V _Lm = nV _out and the exciting inductor current I _Lm rise linearly. This mode ends when the resonant inductor current I _Lr rises to zero. 8 and 9 are conduction paths and equivalent circuit diagrams of SRC-mode 1, respectively.

The resonant inductor current I _Lr (s) and the resonant capacitor voltage V _Cr (s) in the figure can be respectively obtained from the equivalent circuit of FIG. 9, and the relationship is derived as follows:

(twenty two)

(twenty three)

The inverse pull conversion of equations (22) and (23) can be obtained:

(twenty four)

(25)

among them

Characteristic impedance , resonant frequency.

SRC-Mode 2 ( ):

Mode 2 starts at t ₁ , the upper bridge switch remains in the on state, the lower bridge switch remains in the off state, the resonant inductor current I _Lr begins to commutate, and the energy is transmitted to the secondary side rectifying diode via the resonant tank and the transformer. D ₁ and then to the load terminal, the voltage across the magnetizing inductance L _m is the output voltage due to a back side of the mapping L _m immobilized by a short circuit to form a dynamic, then the magnetizing inductance voltage V _Lm = nV _out the magnetizing inductor current I _Lm and still maintain negative Linear rise state. When the magnetizing inductor current I _Lm rises to zero, it proceeds to the next mode. 10 and 11 are conduction paths and equivalent circuit diagrams of SRC-mode 2, respectively.

From the equivalent circuit of Fig. 11, the resonant inductor current I _Lr (s) and the resonant capacitor voltage V _Cr (s) in the figure can be respectively obtained, and the relationship is derived as follows:

(26)

(27)

The inverse pull conversion of equations (26) and (27) can be obtained:

(28)

(29)

among them

Characteristic impedance , resonant frequency.

SRC-Mode 3 ( ):

Mode 3 starts at t ₂ , the upper bridge switch remains on, the lower bridge switch remains in the off state, and the resonant current begins to fall after reaching the maximum value. The energy is transmitted to the secondary side rectifying diode via the resonant tank and the transformer and then sent to At the load end, the voltage on the magnetizing inductance L _m is dynamically short-circuited by being mapped back to the primary side by the output voltage. At this time, the magnetizing inductance voltage V _Lm = nV _out and the exciting inductor current I _Lm maintains a linear rising state. When the upper bridge switch is turned off at t ₃ , it enters the next mode. 12 and 13 are conduction paths and equivalent circuit diagrams of SRC-mode 3, respectively.

The resonant inductor current I _Lr (s) and the resonant capacitor voltage V _Cr (s) in the figure can be respectively obtained from the equivalent circuit of Fig. 13, and the relationship is derived as follows:

(30)

(31)

The inverse pull conversion of equations (30) and (31) can be obtained:

(32)

(33)

among them

Characteristic impedance , resonant frequency.

SRC-mode 4 ( )

In mode 4, the upper bridge switch is turned off at t ₃ , the lower bridge switch remains in the off state, and the resonant inductor current I _Lr is a continuous flow characteristic. When both the upper and lower bridge switches are turned off, the power supply V _in will be on. a parasitic capacitance _C oss1 bridge switches charging the parasitic capacitance _C oss2 discharge side switch. Referred to as blind when the time interval t ₃ to t ₅ period, the low-side switch are in an off state (Dead time), it is used to avoid switching on the bridge has not yet completely cut off low-side switch has been turned on causing the power supply V _in a short-circuit Case. Mode 4 is terminated when the parasitic capacitance C _{oss1 of the} upper bridge switch is charged to V _in and the parasitic capacitance C _{oss2 of the} lower bridge switch is discharged to be equal to the turn-on voltage of the body diode D _B2 . 14 and 15 are conduction paths and equivalent circuit diagrams of SRC-mode 4, respectively.

The resonant inductor current I _Lr (s) and the resonant capacitor voltage V _Cr (s) in the figure can be respectively obtained from the equivalent circuit of Fig. 15, and the relationship is derived as follows

(34)

(35)

The inverse Laplace transform of equations (34) and (35) can be obtained:

(36)

(37)

among them

Characteristic impedance ,Resonant frequency . Total parasitic capacitance, .

SRC-mode 5 ( t ₄ < t < t ₅ ):

In mode 5, the upper and lower bridge switches are still in the off state. When the parasitic capacitance C _{oss1 of the} upper bridge switch is charged to V _{in ,} it is no longer charged, and the parasitic capacitance C _{oss2 of the} lower bridge switch is discharged to be electrically connected to the body diode D _B2 . When the voltages are equal, the current path will be changed to D _B2 . At this time, the resonant inductor current I _Lr will drop rapidly. When the lower bridge switch is turned on at t ₅ , the mode 5 is ended and the next mode is entered. 16 and 17 are conduction paths and equivalent circuit diagrams of SRC-mode 5, respectively.

The resonant inductor current I _Lr (s) and the resonant capacitor voltage V _Cr (s) in the figure can be respectively obtained from the equivalent circuit of Fig. 17, and the relationship is derived as follows

(38)

(39)

The inverse Laplace transform of equations (38) and (39) can be obtained:

(40)

(41)

Characteristic impedance ,Resonant frequency .

3.2 LLC-SRC Circuit Operation Mode Analysis (Zone 2)

The LLC-SRC is a state in which the LLC resonant converter operates in the region 2, and its mode can be analyzed and discussed in two stages. In the first stage, the magnetizing inductor current I _{Lm is} smaller than the resonant inductor current I _Lr , the resonant component is the resonant capacitor C _r , and the resonant inductor L _r ; in the second phase, the exciting inductor current I _Lm is equal to the resonant inductor current I _Lr , and the resonant component is a resonant capacitor C _r , resonant inductor L _r , and magnetizing inductance L _m . Figure 18 shows the timing diagram of the LLC resonant converter operating in zone 2. One switching cycle can be divided into twelve phases. Since the t ₀ to t ₆ circuit action is similar to t ₆ to t ₁₂ , this case only analyzes t. ₀ to t ₆ circuit action, the following will analyze and explain for each mode:

LLC-SRC-Mode 1 ( ):

In mode 1, the upper bridge switch is turned on at t ₀ and the lower bridge switch is turned off. The resonant inductor current I _{Lr is} changed from the D _B1 path to the upper bridge switch itself to V _in due to the continuous flow characteristic. via a transformer, the secondary side rectifier diode D ₁ is transmitted to the load terminal and the output capacitor is charged, the voltage across the magnetizing inductance L _m is mapped back to the primary side of the output voltage immobilized by L _m which forms the dynamic short-circuited, the voltage V _Lm magnetizing inductance = nV _out and the magnetizing inductor current I _Lm rises linearly. When the resonant inductor current I _Lr rises to zero, mode 1 will end and proceed to the next mode. 19 and 20 are conduction paths and equivalent circuit diagrams of the LLC-SRC-mode 1, respectively.

The resonant inductor current I _Lr (s) and the resonant capacitor voltage V _Cr (s) in the figure can be respectively obtained from the equivalent circuit of FIG. 20, and the relationship is derived as follows:

(42)

(43)

The inverse pull conversion of equations (42) and (43) can be obtained:

(44)

(45)

among them

Characteristic impedance ,Resonant frequency.

LLC-SRC-Mode 2 ( ):

Mode 2 starts at t ₁ , the upper bridge switch remains in the on state, the lower bridge switch remains in the off state, the resonant inductor current I _Lr begins to commutate, and the energy is transmitted to the secondary side rectifying diode via the resonant tank and the transformer. D ₁ and then to the load terminal, the voltage across the magnetizing inductance L _m is the output voltage due to a back side of the mapping L _m immobilized by a short circuit to form a dynamic, then the magnetizing inductance voltage V _Lm = nV _out the magnetizing inductor current I _Lm and still maintain linearity rise status. When the resonant inductor current I _Lm rises to zero, mode 2 will end and proceed to the next mode. 21 and 22 are conduction paths and equivalent circuit diagrams of the LLC-SRC-mode 2, respectively.

From the equivalent circuit of Fig. 22, the resonant inductor current I _Lr (s) and the resonant capacitor voltage V _Cr (s) in the figure can be respectively obtained, and the relationship is derived as follows:

(46)

(47)

The inverse pull conversion of equations (46) and (47) can be obtained:

(48)

(49)

Characteristic impedance ,Resonant frequency.

LLC-SRC-Mode 3 ( ):

Mode 3 starts at t ₂ , the upper bridge switch remains on, the lower bridge switch remains in the off state, and the resonant inductor current I _Lr reaches a maximum value and then begins to fall. The energy is transmitted to the secondary side rectified diode via the resonant tank and the transformer. The body is sent to the load end, and the voltage on the magnetizing inductance L _m is dynamically short-circuited by the output voltage mapping back to the primary side. At this time, the magnetizing inductance voltage V _Lm = nV _out and the exciting inductor current I _Lm maintains a linear rising state. When the resonant inductor current I _Lr is equal to the magnetizing inductor current I _Lm , mode 3 will end and proceed to the next mode. 23 and 24 are conduction paths and equivalent circuit diagrams of LLC-SRC-mode 3, respectively.

From the equivalent circuit of Fig. 24, the resonant inductor current I _Lr (s) and the resonant capacitor voltage V _Cr (s) in the figure can be respectively obtained, and the relationship is derived as follows:

(50)

(51)

The inverse pull conversion of equations (50) and (51) can be obtained:

(52)

(53)

Characteristic impedance ,Resonant frequency.

LLC-SRC-Mode 4 ( t ₃ < t < t ₄ ):

In mode 4, the upper bridge switch remains on and the lower bridge remains in the off state. Since the resonant inductor current I _Lr is equal to the magnetizing inductor current I _Lm at t ₃ , the current on the primary side of the transformer is zero, the transformer is decoupled, and energy cannot be transferred to the secondary side, so the rectifier diode is turned off. The output voltage is not mapped back to the primary side clamped magnetizing inductance L _m via the transformer. At this time, the magnetizing inductance L _m will participate in the resonance, and the output energy is provided by the output filter capacitor. When the upper bridge switch is turned off, mode 4 will be terminated and the next mode will be entered. 25 and 26 are conduction paths and equivalent circuit diagrams of LLC-SRC-mode 4, respectively.

From the equivalent circuit of Fig. 26, the resonant inductor current I _Lr (s) and the resonant capacitor voltage V _Cr (s) in the figure can be respectively obtained, and the relationship is derived as follows:

(54)

(55)

The inverse Laplace transform of equation (54) and equation (55) can be obtained:

(56)

(57)

Characteristic impedance ;Resonant frequency .

LLC-SRC-Mode 5 ( t ₄ < t < t ₅ ):

In mode 5, the upper bridge switch is turned off at t ₄ , and the lower bridge switch remains in the off state. Since the resonant inductor current I _{Lr is} still equal to the exciting inductor current I _Lm , the transformer is decoupled and the magnetizing inductance participates in resonance. Energy is provided by the output filter capacitor. When the parasitic capacitance C _{oss1 of the} upper bridge switch is charged to V _in and the parasitic capacitance C _{oss2 of the} lower bridge switch is discharged to be equal to the turn-on voltage of the body diode D _B2 , the mode 5 is ended and the next mode is entered. 27 and 28 are conduction paths and equivalent circuit diagrams of the LLC-SRC-mode 5, respectively.

From the equivalent circuit of Fig. 28, the resonant inductor current I _Lr (s) and the resonant capacitor voltage V _Cr (s) in the figure can be respectively obtained, and the relationship is derived as follows:

(58)

(59)

Inverted Laplace transform of equation (58) and equation (59) can be obtained:

(60)

(61)

Characteristic impedance ;Resonant frequency , total parasitic capacitance .

(6) LLC-SRC-mode 6 ( t ₅ < t < t ₆ ):

In mode 6, both the upper and lower bridge switches are in the off state. When the power supply V _in charges the parasitic capacitance C _oss1 of the upper bridge switch to V _in is no longer charged, and the parasitic capacitance C _{oss2 of the} lower bridge switch is discharged to be equal to the on-voltage of the body diode D _B2 , the current path is changed to D The path flow of _B2 , at this time, the resonant inductor current I _{Lr is} still equal to the exciting inductor current I _Lm , the transformer is still decoupled, and the output energy is provided by the output capacitor C _o . When the lower bridge switch is turned on, mode 6 will be terminated and the next mode will be entered. 29 and 30 are conduction paths and equivalent circuit diagrams of LLC-SRC-mode 6, respectively.

The resonant inductor current I _Lr (s) and the resonant capacitor voltage V _Cr (s) in the figure can be respectively obtained from the equivalent circuit of FIG. 30, and the relationship is derived as follows:

(62)

(63)

Inverted Laplace transform of equation (62) and equation (63) can be obtained:

(64)

(65)

Characteristic impedance ;Resonant frequency .

LLC Converter Synchronous Rectification Technology

When the LLC resonant converter operates in the zero voltage switching interval, the power loss generated by its power switch is very small. In this mode of operation, the maximum power loss is mainly the output rectifying diode, which is also the main reason for the low efficiency. The half-bridge series resonant converter generally uses a Schottky diode as the secondary side rectifying diode, but in the application of low voltage and high current, it will be due to the forward voltage drop of the Schottky diode. _f produces a large conduction loss (eg output voltage 24 V / output current 10 A / forward voltage V _F = 0.85 V / on-resistance R _{ds (on)} = 17 mΩ, loss using a rectifying diode , the loss of using MOSFET is At the same time, this loss will be transferred into heat and reduce the life of the Schottky diode. In order to reduce conduction losses and reduce heat, synchronous rectification techniques are often used to increase converter efficiency.

4.1 Discussion on synchronous rectification drive mode

Synchronous rectification drive methods can be divided into two types:

(1) Self-Driven: The self-excited drive mode is the drive signal required by the circuit itself. Self-excited drive is divided into voltage self-excitation and current self-excitation. The voltage self-excited drive mostly generates signals from the secondary winding of the transformer or an additional winding to the synchronous rectifier switch to achieve the purpose of driving the switch. Although the self-excited voltage has the advantages of simple structure and low cost, it has the disadvantage of being limited by the architecture; the current self-excited driving is to connect a current sensor in front of the synchronous rectification switch and sense the current flowing through it. The drive circuit generates a drive signal switching synchronous rectification switch. The advantage of the current self-excitation type is that there is no architectural limitation.

(2) Its external drive (External-Driven): its exciting drive mode is to sense the current signal flowing through the synchronous rectification switch and send the correct drive signal through the external drive circuit to drive the synchronous switch. Its excitation drive has the advantage of stable drive signal and is not easy to be distorted, but additional drive circuit is required, which increases circuit cost and complexity. On the other hand, digital control can give precise control signals, and has the advantages of high reliability and temperature resistance, which causes component parameters to drift. In order to achieve stable driving signal and high reliability control, the driving method of digital control technology is adopted in this case, and the control signal is directly generated by the control core as the gate signal of the secondary side synchronous rectification switch. In addition, in order to avoid the interference between the large power of the power stage and the small power of the control stage, the large electrical signal of the power stage needs to be electrically isolated from the small electrical signal of the control stage.

4.2 Synchronous rectification control signal considerations

When the LLC resonant circuit with synchronous rectification function operates in region 2, if the synchronous rectification switch is not turned off before the resonant inductor current is equal to the excitation inductor current, a malfunction will occur because no current flows into the primary side of the transformer, and the transformer is formed. Decoupling, there is no energy transmitted to the secondary side. Since the MOSFET is a symmetrical element, the current flowing through it during conduction is bidirectional. At this time, the current on the secondary side is reversed from the secondary side back to the primary side through the MOSFET, so that the secondary side of the original cut-off is connected, not only Disturbs the original resonant action and, in severe cases, damages the MOSFET. In contrast, the previously used Schottky diodes, because of the unidirectional current flow, can block the secondary current recharge problem.

When operating in zone 1, there is a current flow in the primary side of the transformer at all times, and it is in the state of transmitting energy throughout the working cycle. Therefore, there is no problem of current recirculation when adding synchronous rectification. In addition, the difference between the MOSFET and the rectifying diode is that when the synchronous rectification switch is turned off, the current on the primary side of the transformer still flows from the striking end, so the current flowing from the secondary side of the striking end flows through the body diode of the MOSFET. Therefore, it should be noted that the body diode of the MOSFET used must be in the same direction as the rectifier diode to be replaced.

The operation interval of this case is designed in area 2. There is a problem of current recirculation due to operation in this area, and the secondary side control signal must consider the blind time interval. Figure 31 is a schematic diagram of the synchronous rectification switch signal used in the present invention. As shown in FIG. 31, a delay on-time and an advance-off time are added between the control signals V _gs3 and V _gs4 and the control signals V _gs1 and V _gs2 .

Hardware circuit specification and component design

The following is a detailed description of the design of the primary side power switch, resonant inductor, resonant capacitor, main transformer lap design and core selection, and secondary side synchronous rectification switch and output capacitor selection of the synchronous rectified LLC resonant converter of the present invention. Consideration.

5.1 circuit specification

First define the parameters used in the later formula, as listed below: Transformer turns ratio: ,Resonant frequency: , series resonant inductor: , resonant capacitor: , resonant inductance ratio: . The circuit specifications of one embodiment of the LLC resonant converter to be developed are as follows:

Input voltage range :380~432 V

Input rated voltage :390 V

Maximum output power :432 W

Output voltage range :48 V

Output maximum current :9 A

First resonant frequency :100 kHz

Second resonant frequency :30 kHz

Circuit switching frequency :85 kHz ~ 185 kHz

5.2 Circuit component parameter design and selection

5.2.1 Resonant tank parameter design

Step 1: Determine the transformer turns ratio

According to the frequency response analysis of the aforementioned LLC resonant converter, the voltage conversion ratio is equal to half of the transfer function of the AC resonant module at the switching frequency, so the required transformer turns ratio can be calculated, and the expression is as follows:

(66)

(67)

will , Substitute (67) available (匝), wherein the forward voltage drop of the secondary side rectifying diode is too small, and the present invention adopts the turns ratio in this embodiment. .

Step 2: Determine the voltage gain range of the resonant tank

Through the following mathematical formulas, the demand gain can be obtained, and then the voltage gain range of the circuit from no load to full load can be known.

(68)

will , , Substitute (68) available

(69)

will , , Substitute (69) is available

Step 3: Calculate the equivalent load resistance :

Using the derivation result of equation (17), the AC load resistance equivalent to the primary side of the transformer can be calculated, and the calculation formula is as follows:

(70)

will , , Substitute (70) available .

Step 4: Inductance ratio ( ) design:

Higher peak gain can be increased by Value to get, but too big The value will cause transformer coupling degradation and efficiency degradation. The following will calculate the inductor ratio of the no-load maximum operating frequency to the maximum input voltage.

(71)

(72)

among them The maximum operating frequency for normalization.

Step 5: Make sure ZVS Calculation and selection:

Maintains ZVS with full load and minimum input voltage Calculation

(73)

The LLC resonant converter input voltage range is determined by the peak voltage gain, and below the peak gain point will enter the zero current switching interval. To ensure operation at zero voltage switching, the input impedance at the maximum peak gain point has zero phase. Characteristic, Q value has some margin range (5% to 10%) to choose from. The invention in this embodiment has a 10% margin as an option:

(74)

Guarantee the quality factor of ZVS under no load conditions Calculation

(75)

Therefore, in order to ensure that ZVS can be achieved over the entire resonant converter operating range, one must be selected below with The maximum quality factor. The invention is selected in this embodiment .

Step6: Calculate the minimum operating frequency under full load and minimum input voltage :

(76)

(77)

Minimum operating frequency employed by the present invention in this embodiment .

Step 7: Resonant slot design

The characteristic impedance Z _o and the resonant inductance ratio of the resonant converter defined above The parameter values and the theoretical values of the relevant components in the resonant tank are obtained as follows:

(78)

(79)

In this embodiment, the resonant capacitor C _r = 100 nF and the resonant impedance are selected. . If a larger resonant capacitor value is selected, the voltage stress on the resonant capacitor is smaller, but the impedance of the resonant tank is lowered, which will result in a larger short-circuit current and a higher switching frequency, which will limit the change of the output current. . Therefore, the present invention selects the lowest dielectric loss angle (Dielectric Loss Angle) in this embodiment, and is applicable to a polypropylene film capacitor (Polypropylene) in which a high voltage pulse wave circuit operates. Then find the values of the resonant inductor and the magnetizing inductance

(80)

(81)

The resonant inductor used in this embodiment is , magnetizing inductance .

5.2.2 Primary side power switch selection

The primary side power switch of the LLC resonant converter is dominated by a gold oxide half field effect transistor (MOSFET), which has a faster switching speed. In addition to considering that the withstand voltage must be greater than the input DC voltage and the current resistance must be greater than the primary side current, the only consideration is the input capacitance of the power switch. Parasitic capacitance And on resistance . In general, the smaller the on-resistance on the power switch, the smaller the conduction loss, but the larger the parasitic capacitance. Or input capacitance . If a power switch with a large parasitic capacitance is selected, the characteristics of zero voltage switching are not easily achieved. After combining the above points and considering the overall loss, the following parameter values should be considered in order to sort out the power switch:

The rated withstand voltage of the switch is the input DC voltage.

The rated current resistance of the switch is related to the magnitude of the input current.

The on-resistance of the switching element.

Open the parasitic capacitance of the component ( , ).

In this embodiment, the primary side power switch selects the MOSFET of the model IPP65R110CFD produced by Infineon, which has a withstand voltage of 650V, a current with a current resistance of 99.6 A, and an on-resistance. Is 0.099 Ω , parasitic capacitance It is 160 pF.

5.2.3 Secondary side power switch selection

The synchronous rectified LLC half-bridge resonant converter uses two power switching MOSFETs on the secondary side. The selection of this power switch must also consider its withstand voltage and current resistance. In the current-resistant part, the output current will flow through the MOSFET, so the MOSFET must withstand the maximum current at full load. for

(82)

In the withstand voltage part, the MOSFET must withstand the maximum withstand voltage for

(83)

From equations (82) and (83), it can be inferred that the secondary MOSFET withstand voltage and current resistance of the present invention in this embodiment are respectively

The secondary side MOSFET selected by the present invention in this embodiment is selected from the model AOT2918L, which has a withstand voltage of 100 V and a current with a current resistance of 90 A. For 5.6 The withstand voltage and current resistance both satisfy the specifications of the secondary side MOSFET of the present invention in this embodiment.

5.3 digital controller design

In this embodiment, the dsPIC33FJ16GS502 microprocessor introduced by Microchip is used as a digital control core, which sends analog signals by feedback voltage sampling to the microprocessor, first through an analog digital converter (Analog Digital Control, ADC). After converting the analog data into digital data, it is filtered by a digital filter, and then the filtered result is sent to a digital proportional-integral-derivative (Proportional, Integral, Differential, PID) compensator operation, and then the compensation result is The PWM module outputs an appropriate PWM signal to drive the power switch and complete a closed loop control, thereby achieving digital control of the LLC resonant converter with synchronous rectification.

5.3.1 firmware design

The program used in this embodiment of the present invention is mainly divided into two parts, a main program and an ADC interrupt subroutine, as shown in FIG. 32(a) and FIG. 32(b). The main program shown in Fig. 32(a) first declares the required Global Variable and Local Variable, and uses the variable name, register, and output for the required variables. The module (ADC, TIMER, PWM) enable and interrupt vector are initialized, and then enter the infinite loop to wait for the interrupt flag to occur.

The ADC interrupt subroutine flow chart is shown in Figure 32(b). The ADC interrupt subroutine is roughly divided into two sections: sampling filtering and incremental PID calculation compensation. When the interrupt is triggered, it will enter the ADC interrupt subroutine, perform ADC conversion, FIR filtering, and PID feedback compensation to determine whether the output value is less than the lowest frequency or greater than the highest frequency. If yes, set the minimum frequency or the highest frequency respectively. For the purpose of inverter control, the final program will clear the ADC interrupt flag and return to the main program to enter the infinite loop to wait for the next ADC interrupt to occur.

5.3.2 Digital PID Controller

The PID controller consists of three parts: proportional (P), integral (I), and differential (D). The output can be expressed as

(84)

Where u(t): the output of the controller, e(t): the amount of error in the input, K _P , K _I , K _D : proportional gain, integral gain, and differential gain. The digital control system is a kind of sampling control, which can only calculate the output control amount according to the error amount of the sampling point and the input command. Therefore, the discretization method is needed to calculate the digital sampling value. The formula (84) can be represented as a discrete form as shown in the formula (85).

(85)

Among them, e(n): the current error amount of the system, e(n-1): the previous error amount of the system, T: the sampling period.

If the microprocessor implements the digital PID control of equation (85), because it contains integral terms, the memory memory width of the microprocessor is 16 bits, and the overflow may occur during the operation to make the result wrong, in order to reduce the micro processing. The calculation amount of the device and the improvement of the performance, so the case uses an incremental PID controller. The expression of the incremental PID can be derived via equation (85). The sample value of the n-1th sampling time is known from the equation (85).

(86)

Control increment The expression is as follows:

(87)

Subtracting equation (85) from equation (86) into equation (87)

(88)

First hypothesis , as well as , Equation (87) can be rewritten as Equation (88).

(88)

The incremental PID program flow is shown in Figure 33, which subtracts the output command value from the sampled value to obtain an error amount e(n), and then the previous error amount e(n-1) and the first two. The error amount e(n-2) of the second is substituted into the equation (88) to obtain three ERROR parameters ER_KP_V_0, ER_V_0, ER_KD_V_0, and sequentially multiply K _p , K _I , K _D and add them to obtain one. Output variation , PID output result PID _out is equal to Plus the amount of the previous time period, and then compared with the lower limit on the period, if the output result is smaller than or greater than the upper limit Period _min Period _max, the output are equal to the lower limit or the upper limit value, and finally outputs the result to the PWM generator.

Experimental result

The experimental results are divided into two parts. The first part presents the measured waveform result of the digitally controlled synchronous rectification LLC resonant converter proposed by the present invention, and the second part is delayed and turned on and off by the secondary side synchronous switching signal than the primary side switching signal. The implementation method compares efficiency trends. Finally, the actual measured efficiency data is presented.

Figure 34 is a waveform diagram of the switch control signal measured by the embodiment of the present invention. In the figure, V _gs1 and V _gs2 are driving signals of the primary side switch, respectively, and V _gs3 and V _gs4 are driving signals of the secondary side synchronous rectifying switch. Fig. 35 is a waveform diagram showing driving signals of the primary side switch measured in the embodiment of the present invention. It can be seen from the measurement data in FIG. 35 that the primary side signal is set to a 300 ns blind time interval to prevent the switching elements from being simultaneously turned on to cause a short circuit of the power supply or damage of the circuit components. Figure 36 is a diagram showing the zero voltage switching waveform of the power switch measured in this embodiment at light load (1A) and full load (9A).

37(a) to 37(c) show that the LLC resonant converter of this embodiment of the present invention is set to a constant voltage mode, in which the input voltage is fixed at 380 V and the output voltage is fixed at 48 V, respectively, in light load (1A), Gate voltage V _GS measured by (5A) and heavy load (9A), voltage across the power component V _DS , transformer primary voltage V _P , resonant current i _Lr , switching frequency, resonant tank current and The waveform of the primary side voltage of the transformer, in which, at light load, the switching frequency is the highest (the value is 125.5kH _Z ) in order to maintain a stable output voltage; and in Figure 37(b) and Figure 37(c) The resonant current waveform can be inferred that the LLC resonant converter has entered the operating range of zone 2.

Then, the primary side switching signal and its blind time interval are fixed and the blind time interval of the secondary side switching signal is adjusted to observe the change of the circuit efficiency, and the secondary side switching signal is compared with the primary side switching signal to delay conduction and early cutoff. The method realizes the circuit action and compares the experiment with different input voltage and blind time interval. Figure 38(a)~38(c) are the efficiency versus load curves for the fixed-side switching signal and its blind time interval and adjusting the blind-time interval of the secondary-side switching signal at the input voltages of 380V, 390V, and 400V. .

From Fig. 38(a)~38(c), it is found that regardless of the input voltage, the blind time interval of the secondary side switching signal is 50 ns, which has higher efficiency under different load conditions. In terms of efficiency trends, the smaller the blind time interval, the higher the efficiency.

in conclusion

In terms of power conversion efficiency, the driving signal of the synchronous rectification switch must increase its conduction time as much as possible to reduce the on-time of the body diode and reduce the power loss of the synchronous rectification switch. It can be seen from the experimental efficiency graph that the smaller the blind time interval of the secondary side synchronous rectification switch, the higher the power conversion efficiency.

The disclosure of the present invention is a preferred embodiment. Any change or modification of the present invention originating from the technical idea of the present invention and being easily inferred by those skilled in the art will not deviate from the scope of patent rights of the present invention.

In summary, this case, regardless of its purpose, means and efficacy, is showing its technical characteristics that are different from the conventional ones, and its first invention is practical and practical, and it is also in compliance with the patent requirements of the invention. I will be granted a patent at an early date.

100‧‧‧First half bridge switching circuit
110‧‧‧Capacitor-inductor series circuit
120‧‧‧Transformers
130‧‧‧Second half bridge switching circuit
140‧‧‧ output capacitor
150‧‧‧Load resistor
160‧‧‧Return circuit
170‧‧‧Control unit
161‧‧‧voltage circuit
162‧‧‧Optical coupling circuit
171‧‧‧ analog to digital conversion unit
172‧‧‧Filtering unit
173‧‧‧Proportional-Integral-Derivative Unit
174‧‧‧ pulse width modulation unit
175‧‧‧ drive unit

1 illustrates an embodiment of a high efficiency LLC resonant converter for blind time regulation of a secondary side synchronous rectifier of the present invention. FIG. 2 illustrates the architecture of an LLC resonant converter employing a diode on the secondary side. 3 is a circuit diagram of converting the nonlinear circuit of FIG. 2 into a linear double-turn model. 4 is an equivalent circuit diagram of the LLC resonant tank of FIG. 2. Figure 5 is the LLC resonant circuit of Figure 2 in different The voltage gain under the value and the normalized frequency response plot. Figure 6 shows the main current and voltage signals of the LLC resonant converter of Figure 2. FIG. 7 is a timing chart showing the operation of the SRC. 8 and 9 respectively show the conduction path and equivalent circuit diagram of the SRC-mode 1. 10 and 11 respectively show the conduction path and equivalent circuit diagram of the SRC-mode 2. 12 and 13 respectively show the conduction path and equivalent circuit diagram of the SRC-mode 3. 14 and 15 respectively show the conduction path and equivalent circuit diagram of SRC-mode 4. 16 and 17 respectively show the conduction path and equivalent circuit diagram of the SRC-mode 5. FIG. 18 illustrates a timing diagram of the LLC resonant converter operating in region 2. 19 and 20 respectively show the conduction path and equivalent circuit diagram of the LLC-SRC-mode 1. 21 and 22 respectively show the conduction path and equivalent circuit diagram of the LLC-SRC-mode 2. 23 and 24 respectively show the conduction path and equivalent circuit diagram of the LLC-SRC-mode 3. 25 and 26 respectively show the conduction path and equivalent circuit diagram of the LLC-SRC-mode 4. 27 and 28 respectively show the conduction path and equivalent circuit diagram of the LLC-SRC-mode 5. 29 and 30 respectively show the conduction path and equivalent circuit diagram of the LLC-SRC-mode 6. Figure 31 is a schematic diagram of the synchronous rectification switch signal used in the present invention. 32(a) and 32(b) illustrate a main program and an ADC interrupt subroutine used in an embodiment of the present invention. Figure 33 is a flow chart of an incremental PID program. Figure 34 is a waveform diagram of the switch control signal measured in an embodiment of the present invention. Figure 35 is a waveform diagram of driving signals of the primary side switch measured in an embodiment of the present invention. Figure 36 is a diagram showing the zero voltage switching waveform of the power switch measured in one embodiment of the present invention at light load (1A) and full load (9A). 37(a) to 37(c) show the LLC resonant converter according to an embodiment of the present invention in a constant voltage mode, in which the input voltage is fixed at 380 V and the output voltage is fixed at 48 V, respectively, in light load (1A), Gate voltage V _GS measured by (5A) and heavy load (9A), voltage across the power component V _DS , transformer primary voltage V _P , resonant current i _Lr , switching frequency, resonant tank current and Waveform of the primary side voltage of the transformer. 38(a) to 38(c) respectively illustrate a blind time interval in which the primary side switching signal and its blind time interval are fixed and the secondary side switching signal is adjusted at an input voltage of 380V, 390V, and 400V according to an embodiment of the present invention. Efficiency vs. load graph.

100‧‧‧First half bridge switching circuit

110‧‧‧Capacitor-inductor series circuit

120‧‧‧Transformers

130‧‧‧Second half bridge switching circuit

140‧‧‧ output capacitor

150‧‧‧Load resistor

160‧‧‧Return circuit

170‧‧‧Control unit

161‧‧‧voltage circuit

162‧‧‧Optical coupling circuit

171‧‧‧ analog to digital conversion unit

172‧‧‧Filtering unit

173‧‧‧Proportional-Integral-Derivative Unit

174‧‧‧ pulse width modulation unit

175‧‧‧ drive unit

Claims (5)

A high-efficiency LLC resonant converter for blind-time regulation of a secondary side synchronous rectifier, comprising: a first half-bridge switching circuit having two inputs for coupling with positive and negative ends of an input voltage, and two control terminals for Each of the first driving signal and the second driving signal is coupled to the first driving signal and the second driving signal, and an output terminal is coupled to the positive terminal when the first driving signal exhibits an active potential, and the second driving signal exhibits an active potential The negative terminal is coupled; a capacitor-inductor series circuit having one end coupled to the output end of the first half bridge switch circuit; a transformer having a main coil and a primary coil, one end of the main coil The other end of the main circuit is coupled to the negative end of the input voltage, and the second coil has a first output end, a second output end, and a center tap contact; a second half bridge switch circuit having two input ends coupled to the first output end and the second output end, and two control ends respectively for a third drive signal and a fourth drive Signal coupling, and one input The second output end is coupled to the second output end when the third driving signal is coupled to the first output end and the fourth driving signal is coupled to the second output end; an output capacitor coupled to the first Between the output end of the two-half bridge switch circuit and the center tap contact; a load resistor coupled between the output end of the second half-bridge switch circuit and the center tap contact; a circuit for generating a feedback signal according to one of the load resistors; and a control unit for performing a driving signal generation program to generate the first driving signal and the second driving signal according to the feedback signal The third driving signal and the fourth driving signal, wherein the The operating frequency of a driving signal and the second driving signal is between a first resonant frequency and a second resonant frequency, wherein the first resonant frequency is a resonant frequency of a resonant inductor and a resonant capacitor, and the first The second resonant frequency is a resonant inductor, the resonant inductor and a resonant frequency of the resonant capacitor, and the first half-bridge switching circuit is driven in a zero voltage switching mode and a fixed connection between the first driving signal and the second driving signal The third driving signal has a delayed on-time and an advanced off-time with respect to the first driving signal, and the fourth driving signal has the delayed on-time and the advanced-off time with respect to the second driving signal. And the driving signal generating program includes a proportional-integral-differential operation to adjust the delayed on-time and the advance-off time to reduce the power loss of the second half-bridge switching circuit. The high-efficiency LLC resonant converter for blind-time regulation of a secondary side synchronous rectifier according to claim 1, wherein the feedback circuit comprises a voltage dividing circuit and an optical coupling circuit. A high-efficiency LLC resonant converter for blind-time regulation of a secondary-side synchronous rectifier according to claim 1, wherein the drive signal generating program includes an analog-to-digital conversion operation. A high-efficiency LLC resonant converter for blind-time regulation of a secondary side synchronous rectifier according to claim 3, wherein the drive signal generating program further comprises a filtering operation. The high efficiency LLC resonant converter of the secondary side synchronous rectifier blind time control according to claim 1, wherein the control unit comprises a pulse width modulation module to provide the first driving signal, the first a second drive signal, the third drive signal, and the fourth drive signal.