TWI683523B - LLC resonant converter capable of adjusting input voltage with load variation - Google Patents

Abstract

An LLC resonant converter capable of adjusting input voltage according to load fluctuations has a half-bridge switching circuit with two input terminals to couple with a positive and negative terminal of an input voltage, and two control terminals to respectively receive a first driving signal Coupled to a second drive signal, and an output terminal to be coupled to the positive terminal when the first drive signal exhibits an active potential and to the negative terminal when the second drive signal exhibits an active potential; A capacitor-inductor series circuit, one end of which is coupled to the output end of the half-bridge switching circuit; a transformer having a main coil and a primary coil, one end of the main coil is connected to the other of the capacitor-inductor series circuit One end is coupled, the other end of the main coil is coupled to the negative end of the input voltage, the secondary coil has a first output end, a second output end, and a center tap contact; a first The diode has a first anode and a first cathode, the first anode is coupled to the first output terminal, the first cathode is coupled to a voltage output terminal; a second diode has A second anode and a second cathode, the second anode is coupled to the second output terminal, the second cathode is coupled to the voltage output terminal; an output capacitor is coupled to the voltage output terminal and the Between the center tap contacts; a load resistor coupled between the voltage output terminal and the center tap contact; a feedback circuit for generating a feedback signal based on a voltage across the load resistor; and a The control unit is used for performing a proportional-integral-derivative operation according to the difference between the voltage value of the feedback signal and a preset voltage value to determine a PWM operating frequency, and generating the first driving signal and the PWM operating frequency according to the PWM operating frequency The second driving signal, and when the PWM operating frequency is lower than a preset resonant frequency, the input voltage is increased by a voltage difference, and when the PWM operating frequency is higher than a preset resonant frequency, the input voltage is reduced The voltage difference.

Description

LLC resonant converter capable of adjusting input voltage with load variation

The invention relates to a switching power supply, in particular to an LLC resonant converter capable of adjusting input voltage according to load fluctuation.

The 21st United Nations Climate Change Conference (COP21), held in Paris, France in 2015, adopted the historically inclusive and legally binding carbon reduction agreement-the Paris Agreement, and the participating countries agreed to control greenhouse gas emissions to achieve The global average temperature rise in 2100 does not exceed 2 ^o C, and we strive to control the target within 1.5 ^o C. It can be seen that the deterioration of the global environment, climate, and ecology caused by greenhouse gas emissions has been fully appreciated by the world, and issues such as green environmental protection, energy conservation, and carbon reduction have been truly valued by countries around the world. Electricity is the primary issue of whether humanity can continue to move towards civilization. Since environmental protection concepts and sustainable development have become a global consensus, how to use existing energy sources more efficiently and actively develop new alternative energy sources is currently the top priority in the engineering science and technology community. Business. Therefore, how to reduce electricity consumption and improve the efficiency of electricity conversion and use to reduce greenhouse gas emissions is a problem that we urgently need to solve.

With the advancement of power supply technology, in order to meet the market demand for light, thin, short and high power density (High Power Density), the switching power supply gradually replaces the traditional linear power supply. However, most of the switching power supplies use PWM (Pulse Width Modulation) technology to increase the switching frequency of the main switch to achieve the purpose of reducing the circuit. However, when controlling the power switch on or off, the voltage And the current is not zero, so-called hard switching (Hard Switching), will cause higher switching loss (Switching Loss), and as the frequency increases, switching loss (Switching Loss) and electromagnetic interference will also increase, so It is necessary to use soft switching technology.

Among converters with soft switching function, LLC resonant converters have gradually gained attention in recent years due to their advantages such as ZVS (Zero Voltage Switching) and better voltage regulation.

In order to reduce the switching frequency of LLC resonant converters, some literatures have proposed an asymmetric pulse-width modulation (APWM) for sustaining time, which can obtain higher gain than the same switching frequency. Peak gain placement (Peak Gain Placement) optimization design method is used to minimize the conduction loss within the range that meets the specification gain requirements; there are still literatures that propose adaptive link-voltage-variation , ALVV) method, this method can reduce the switching frequency range of the half-bridge LLC, so that the switching frequency within the full load range can fall near the resonance frequency.

In terms of improving the efficiency of power conversion, some literatures propose that synchronous rectification technology be used in LLC resonant converters, and improve the efficiency of LLC resonant converters with different driving methods; some literatures propose to describe synchronous rectification with steady-state analysis, which can reduce Output the loss of rectification and control the primary and secondary switches with a voltage clamp drive circuit. Finally, implement a LLC resonant converter with synchronous rectification and compare the efficiency with the non-synchronous LLC resonant converter. The result is that the efficiency of synchronous rectification is indeed excellent It is asynchronous.

However, the above documents do not solve the problem that when the operating frequency of the LLC resonant converter is far from the resonant frequency point, the conversion efficiency will also be reduced accordingly. Therefore, there is an urgent need in the art for a novel LLC resonant converter that can adjust the input voltage as the load changes.

An object of the present invention is to disclose an LLC resonant converter that can adjust the input voltage according to load changes. The DC resonant converter can be adjusted to keep the LLC resonant converter operating near the resonant frequency point to achieve the entire load output range. Purpose of high efficiency operation.

Another object of the present invention is to disclose an LLC resonant converter that can adjust the input voltage according to load changes. Under the condition of ZVS, the larger the magnetizing inductance, the higher the efficiency.

Another object of the present invention is to disclose an LLC resonant converter that can adjust the input voltage according to load changes. Within the controllable input voltage range of 380V ~ 400V, the conversion efficiency of light load (0.5A) and full load (10A) are 91.4% and 95.86%, the highest efficiency can reach 96.7%.

In order to achieve the foregoing purpose, an LLC resonant converter capable of adjusting the input voltage according to load fluctuations has been proposed, which has: a half-bridge switching circuit with two input terminals to couple with the positive and negative terminals of an input voltage, and two control terminals To be coupled to a first driving signal and a second driving signal, respectively, and an output terminal to be coupled to the positive terminal when the first driving signal exhibits an active potential and the second driving signal to exhibit an active potential Coupled to the negative terminal; a capacitor-inductor series circuit, one end of which is coupled to the output end of the half-bridge switching circuit; a transformer having a main coil and a primary coil, one end of the main coil is connected to The other end of the capacitor-inductor series circuit is coupled. The other end of the primary coil is coupled to the negative end of the input voltage. The secondary coil has a first output end, a second output end, and a Center tap contact; a first diode with a first anode and a first cathode, the first anode is coupled to the first output terminal, the first cathode is coupled to a voltage output terminal; A second diode having a second anode and a second cathode, the second anode is coupled to the second output terminal, the second cathode is coupled to the voltage output terminal; an output capacitor is coupled It is connected between the voltage output terminal and the center tap contact; a load resistor is coupled between the voltage output terminal and the center tap contact; a feedback circuit is used to cross the voltage according to one of the load resistance Generating a feedback signal; and a control unit for performing a proportional-integral-derivative operation based on the difference between the voltage value of the feedback signal and a predetermined voltage value to determine a PWM operating frequency, based on the PWM operating frequency Generating the first driving signal and the second driving signal, and increasing the input voltage by a voltage difference when the PWM operating frequency is lower than a preset resonant frequency, and when the PWM operating frequency is higher than a preset At the resonant frequency, the input voltage is reduced by the voltage difference.

In one embodiment, the feedback circuit includes a voltage divider circuit and an optical coupling circuit.

In one embodiment, the control unit includes an analog-to-digital converter to perform an analog-to-digital conversion operation on the feedback signal to generate a first input digital signal.

In an embodiment, the control unit includes a filter operation function module to perform a filter operation on the first input digital signal to generate a second input digital signal, and the control unit is based on the second input digital signal and The difference between the preset voltage values performs the proportional-integral-derivative operation.

In one embodiment, the control unit includes a pulse width modulation module to provide the first driving signal and the second driving signal.

In order to enable your review committee to further understand the structure, features and purpose of the present invention, the drawings and detailed description of the preferred embodiments are attached as follows.

Please refer to FIG. 1, which illustrates a block diagram of an embodiment of the LLC resonant converter of the present invention that can adjust the input voltage according to load fluctuations.

As shown in the figure, the LLC resonant converter capable of adjusting the input voltage according to load fluctuations has a half-bridge switching circuit 100, a capacitor-inductor series circuit 110, a transformer 120, a first diode 130, and a second diode Body 140, an output capacitor 150, a load resistor 160, a feedback circuit 170, and a control unit 180.

The half-bridge switching circuit 100 has two input terminals A, B and to the input voltage V _in a positive and a negative terminal coupled to a second control terminal ₂ are respectively coupled with a first driving signals S ₁ and a second driving signal S And an output terminal C is coupled to the positive terminal when the first driving signal S ₁ exhibits an active potential and is coupled to the negative terminal when the second driving signal S ₂ exhibits an active potential.

One end of the capacitor-inductor series circuit 110 is coupled to the output terminal C of the half-bridge switching circuit 100.

The transformer 120 having a primary coil and a secondary coil, one end of the primary winding of the capacitor - the other end of the series circuit of the inductor 110 is coupled to the other end of the main coil system of the input voltage V _in and the negative terminal of Coupling, the secondary coil has a first output terminal D, a second output terminal E, and a center tap contact F.

The first diode 130 has a first anode and a first cathode. The first anode is coupled to the first output terminal D, and the first cathode is coupled to a voltage output terminal O.

The second diode 140 has a second anode and a second cathode. The second anode is coupled to the second output terminal E, and the second cathode is coupled to the voltage output terminal O.

The output capacitor 150 is coupled between the voltage output terminal O and the center tap contact F.

The load resistor 160 is coupled between the voltage output terminal O and the center tap contact F.

The feedback circuit 170 includes a voltage dividing circuit 171 and an optical coupling circuit 172 for generating a feedback signal V _FB according to one of the load resistors 160 across the voltage V _O.

The control unit 180 is used to perform a proportional-integral and derivative (PID) operation based on the difference between the voltage value of the feedback signal V _FB and a predetermined voltage value to determine a PWM operating frequency, according to The PWM operating frequency generates the first driving signal S ₁ and the second driving signal S ₂ , and increases the input voltage by a voltage difference when the PWM operating frequency is lower than a preset resonant frequency, and at the PWM When the operating frequency is higher than a preset resonance frequency, the input voltage is reduced by the voltage difference.

The feedback circuit 170 further includes a voltage divider circuit 171 and an optical coupling circuit 172.

The control unit 180 further includes an analog-to-digital converter 181, a filter operation function module 182, and a pulse width modulation module 183.

The analog-to-digital converter 181 is used for performing an analog-to-digital conversion operation on the feedback signal V _FB to generate a first input digital signal; the filter operation function module 182 is used for the first input digital signal A filtering operation is performed to generate a second input digital signal, and the control unit 180 performs the proportional-integral-derivative operation according to the difference between the second input digital signal and the predetermined voltage value; a pulse wave The width modulation module 183 is used to provide the first driving signal S ₁ and the second driving signal S ₂ .

According to this, the present invention can adjust the DC input voltage to keep the LLC resonant converter operating near the resonance frequency point, so as to achieve the purpose of high efficiency operation in the entire load output range.

The principle of the present invention will be described below:

Please also refer to FIGS. 2a to 2e, wherein FIG. 2a shows a schematic diagram of the half-bridge LLC resonant converter, FIG. 2b shows an equivalent circuit diagram of the series resonant converter converted to the primary side, and FIG. 2c shows Frequency response curves at different quality factor Q values. FIG. 2d shows a schematic diagram of a linear two-port model that converts the non-linear circuit of FIG. 2a into an LLC series resonant circuit. FIG. 2e shows the equivalent of an LLC resonant tank. Circuit diagram.

As shown in FIG. 2a, S1 and S2 are primary-side upper and lower bridge switching elements, Lm is a magnetizing inductance, L _r is a resonance inductance, and C _r is a resonance capacitance. Wherein S1 and S2 switching elements are 50 percent duty cycle alternately conductive, the energy transferred to the secondary side via a resonant tank L _r, C _r consisting of, and must set a blind zone During the period the periphery of both the switch ON of ( Dead Time), the ZVS characteristic is completed in this interval, the output-end transformer adopts center-tapped full-wave rectification, which is suitable for the application of low voltage and large current.

The Lm of the half-bridge series resonant converter does not participate in resonance, and its resonant network is composed of the resonant inductance L _r and resonant capacitor C _r , and the equivalent impedance R _ac formed by the secondary side load reflected to the primary side, as shown in Figure 2b The voltage transfer function is shown in equation (1).

(1)

(1)

，ω _s為開關切換角頻率，品質因數

。 Among them, the resonance frequency

, Ω _s is the switching angular frequency, quality factor

.

並正規化其頻率響應曲線圖f _norm= ω _s/ω _r，如圖2c所示。 Use equation (1) to plot the voltage gain at different Q values

And normalize its frequency response curve graph f _norm = ω _s /ω _r , as shown in Figure 2c.

Then, the first harmonic approximation (FHA) is used to convert the non-linear circuit of FIG. 2a to the linear two-port model of FIG. 2d to analyze the frequency response of the LLC resonant converter circuit.

The equivalent circuit diagram of the LLC resonant tank is shown in Figure 2e, and the equivalent resistance of the secondary side is mapped to the primary side. Assuming that the secondary winding voltage does not contain harmonic components, the AC equivalent resistance R _o,ac can be obtained as shown in equation (2) and equation (3).

(2)

(2)

(3)

(3)

Where P _out is the output power and R _out is the output load.

The input/output transfer function G(s) and the input impedance Z _in (s) are shown in equation (4) and equation (5), respectively.

(4)

(4)

(5)

(5)

After substituting the above equation, the voltage gain of the circuit and the input impedance of the resonant tank can be obtained as shown in equation (6) and equation (7), respectively.

(6)

(6)

(7)

(7)

，特性阻抗

，諧振電感比

，正規化頻率

，品質因數

。 The definition of each parameter is as follows: the first resonance frequency

, Characteristic impedance

, Resonant inductance ratio

, Normalized frequency

,Quality factor

.

Please refer to FIG. 3, which shows the voltage gain and normalized frequency response of the LLC resonant circuit at different Q values.

。 As shown in the figure, the LLC resonant circuit has two resonant frequencies, the resonant frequencies of which are shown in equation (8) and equation (9), respectively, where

.

(8)

(8)

(9)

(9)

The operation of the LLC resonant converter can be divided into three zones, zone-1, zone-2 and zone-3, which are distinguished by the first resonance frequency _fr1 and the second resonance frequency _fr2 . Among them, zone-1 and zone-2 are ZVS zones, and zone-3 are ZCS (Zero Current Switching, zero current switching) zones.

When the switching frequency is greater than the first resonant frequency (f _sw > f _r1 ), the converter operates in region -1 and the circuit gain is less than 1, because the magnetizing inductance L _m is clamped by the voltage reflected by the transformer to the primary side and does not participate in resonance, The resonance frequency is determined by the resonance inductance L _r and the resonance capacitance C _r . The input current of the resonant tank lags the input voltage, so the input impedance is inductive. In this interval, the converter operates like a series resonant circuit (SRC).

When the switching frequency is between the first resonant frequency and the second resonant frequency (f _r2 ＞ f _sw ＞ f _r1 ), the converter operates in the region-2, the voltage gain is greater than 1, and in this interval L _m participates in resonance and resonance The frequency is determined by C _r , L _r and L _eq .

When the switching frequency is less than the second resonance frequency (f _sw <f _r2 ), the converter operates in region-3. Operating in this interval, the input current of the resonant tank leads the input voltage, and the input impedance is capacitive. Because it belongs to the ZCS (Zero Current Switching) zone, it is not the ZVS zone designed by the present invention and is not to be discussed.

If the half-bridge resonant converter is operated in the area-1 or the area-2, the upper and lower bridge switches of the circuit have the characteristics of ZVS, which has advantages for the high-voltage low-current architecture. The present invention uses the LLC half-bridge resonant converter The working area is designed in this area.

LLC The best operating point of resonant converter :

In the design process of resonant converters, high efficiency and output voltage stability have always been the two goals of the design. The following discusses the optimal operating point of LLC half-bridge resonant converters and the impact of magnetizing inductance on efficiency.

The following discusses the operation of the LLC resonant converter at the zone-1, zone-2 and resonance frequency points:

1. Area-1

Please refer to FIG. 4, which shows a typical waveform of the LLC resonant converter operating in the region-1.

As shown in the figure, when the switch S _{1 is} turned on, there is ZVS switching. When t ₁ < t < t ₂ , the switch S ₁ cuts off the switch with a high cut-off current, which results in a high switching loss. In addition, the rectified diode current on the secondary side will instantly drop to zero, so di / dt is very large, resulting in high reverse recovery loss.

2. Area-2

Please refer to FIG. 5, which shows a typical waveform of the LLC resonant converter operating in the region-2.

As shown in the figure, when the switch S _{1 is} turned on, there is ZVS switching. When t ₁ < t < t ₃ , the resonance current i _Lr and the excitation current i _Lm are equal, because all the current flows into the excitation inductance L _{m to} decouple the transformer, the transformer is regarded as an open circuit, L _m participates in the circuit resonance, making the secondary side Both the rectifier diodes D ₁ and D ₂ are cut off, i _Lr circulates in the resonant network and cannot provide energy to the load. At this time, the circulating current will cause additional conduction loss. On the other hand, as the switching frequency moves away from the resonance frequency, the rectifier diode current on the secondary side drops to zero, so di/dt is very small, eliminating the reverse recovery loss.

Third, the resonance frequency point

Please refer to FIG. 6, which shows the waveform of the LLC resonant converter operating at the first resonant frequency ( _{fr 1} ).

As shown in the figure, when the switch S _{1 is} turned on, there is ZVS switching. At t = t ₁ , the resonance current i _Lr and the excitation current i _Lm are equal, the switch S _{1 is} cut off, the cut-off current is much smaller than the cut-off current of the area -1, the secondary side rectifier diode current drops to zero, so di/ The dt is very small, eliminating the reverse recovery loss.

At the resonance point, the LLC resonant converter has the smallest cycle loss in the resonant tank, which corresponds to the lowest conduction loss. At this time, the conduction loss of the resonance frequency is much smaller than that of the region-2. At the same time, the switching loss of the resonance frequency is much smaller than the switching loss of region-1.

It can be concluded that the LLC resonant converter operating at the resonant frequency can achieve the minimum loss and the maximum conversion efficiency. Table 1 is a comparison table of the losses at three different operating points of zone-1, zone-21 and the resonance frequency point.

Table 1 Area-1 Area-2 Resonance frequency point Primary side conduction ZVS ZVS ZVS Primary cut-off loss high low low di/dt high low low Circulation loss in high low Switching loss high low low

Magnetoelectric The effect of L _m on efficiency:

As shown in Figure 6, the switch on the primary side can reach ZVS when turned on, so the switching loss is the cut-off loss of the LLC resonant converter. The switching loss is mainly estimated according to the magnitude of the cut-off current and the length of its cut-off time, and the magnitude of the current can be determined by the magnetizing inductance L _m , as shown in equation (10).

(10)

(10)

When the switch on the primary side is turned on, the conduction loss is mainly estimated according to the size of the resonant tank current i _Lr , as shown in equation (11).

(11)

(11)

Where I _{RMS_P} is the RMS value of the resonant tank current, ω _o is the resonant angular velocity, y is the starting angle of i _Lr , i _{Lm_peak} is the peak value of the exciting current, and T _s is the switching period.

Based on the circuit characteristics of the LLC resonant converter, at the beginning of each switching cycle, i _Lr and i _{Lm_peak} are equal, and equation (12) can be derived.

(12)

(12)

Since the difference between i _Lr and i _{Lm_peak} is the current through the secondary side, the relationship shown in equation (13) can be derived.

(13)

(13)

The current RMS value of the resonant tank can be obtained as shown in equation (14).

(14)

(14)

From equation (14), it can be seen that the larger the magnetizing inductance L _{m is} , the better the conduction loss and the switching loss are reduced. However, ZVS is required to achieve the above, so the ZVS condition needs to be calculated, and the amount of charge on the magnetizing inductance L _m must be equal to The charge amount of its equivalent parasitic capacitance is shown in equation (15).

(15)

(15)

Where C _{oss 1} = C _{oss 2} = C _oss , T _s is the circuit operation period, t _ZVS is the minimum blind time required to reach ZVS, and equation (15) is obtained after collating equation (15).

(16)

(16)

The blind time t _{dead of the} actual circuit must be greater than t _ZVS , then the maximum magnetizing inductance L _{m_max} can be obtained from equation (16) as shown in equation (17).

(17)

(17)

In addition, the value of K (= L _m / L _r ) affects the operating frequency range. Please refer to FIG. 7, which shows the operating frequency variation range under different K values.

As shown in the figure, the larger the value of K, the greater the range of operating frequency variation, which is not conducive to efficiency. And at light loads, the operating frequency is close to infinity, making it difficult to control the output voltage. Accordingly, the present invention can vary with load fluctuation by adjusting the input voltage V _in operation so that the switching frequency is fixed to the vicinity of the resonance frequency to thereby improve efficiency and stabilize the output voltage, maximum power point tracking method is similar to the fixed-order perturbation and observation method step , Because its principle is simple and easy to implement.

Please refer to FIG. 8, which illustrates a schematic diagram of a control mechanism that can adjust the input voltage according to the load variation provided by the present invention.

As shown in the figure, if the current frequency of the system is greater than the resonant frequency (such as part (i)), you can know that the current gain value is less than the resonant frequency point gain. At this time, the input voltage control command is reduced by a change of ΔV, proportional-integral- In order to stabilize the output voltage, the differential controller will reduce the switching frequency and increase the gain to be close to the resonance frequency. Conversely, if the current frequency of the system is less than the resonant frequency (such as part (ii)), it can be known that the gain value at this stage is greater than the gain of the resonant frequency point. At this time, the input voltage control command is added with a variation ΔV, proportional-integral-derivative control In order to stabilize the output voltage, the converter will increase the switching frequency and reduce the gain close to the resonance frequency.

數 Position controller design :

In order to achieve the purpose of digital control, the present invention uses Microchip's dsPIC series microcontroller dsPIC33FJ16GS502 as the controller core to realize a digital LLC resonant converter. The digital controller can simplify the hardware circuit and improve the shortcomings of the passive component being changed by the environment, and can also provide more peripheral functions for the system, so that the design can have more flexibility.

Please refer to FIG. 9, which shows a schematic diagram of a digitally controlled LLC resonant converter.

As shown in the figure, the quantized output voltage data is filtered by a digital filter, and then the filtered result is sent to a digital proportional-integral-derivative compensator for calculation, and finally the PWM module sends a signal to complete closed-loop control.

Firmware program planning: Please refer to Figures 10a~10c together, where Figure 10a shows a flowchart of the overall firmware program of the present invention using the dsPIC33FJ16GS502 microcontroller, and Figure 10b shows the ADC interrupt program of the present invention using the dsPIC33FJ16GS502 microcontroller Fig. 10c is a flow chart of the subprogram of the present invention which can adjust the input voltage as the load changes using the dsPIC33FJ16GS502 microcontroller.

The present invention uses a dsPIC33FJ16GS502 microcontroller to implement a digital filter and an incremental proportional-integral-derivative controller.

As shown in Figure 10a, the firmware program can be divided into a main program, an ADC interrupt subprogram, and an input voltage subprogram that can be adjusted with load changes. At the beginning of the program, first declare the global variables and regional variables, set the variable name, the initial value setting of the register, the input and output port settings, the module (PWM, ADC, TIMER, etc.) enable and the interrupt vector setting, and then enter the infinite loop waiting An interrupt vector flag occurs.

As shown in Figure 10b, once the ADC interrupt is triggered, it will enter the ADC interrupt subroutine, open the counter, perform ADC conversion, FIR filtering, and proportional-integral-derivative feedback compensation to achieve frequency conversion control. The counter value will decrease by one and the program will end Clear the ADC interrupt flag, end the program and enter an infinite loop to wait for the next ADC interrupt.

As shown in Figure 10c, when the counter is triggered by zero, enter the subroutine that can adjust the input voltage according to the load change, calculate the input voltage control command through the current frequency, reset the counter, end the program and wait for the next ADC interrupt.

The control mechanism is shown in Figure 8. The current operating frequency of the system is compared with the resonant frequency. If the current frequency of the system is greater than the resonant frequency, the current gain value is less than the gain of the resonant frequency point. At this time, the input voltage control command is reduced by one The amount of change ΔV, proportional-integral-derivative controller in order to stabilize the output voltage, will reduce the switching frequency, increase the gain to close to the resonance frequency. Conversely, if the current frequency of the system is less than the resonant frequency, it can be known that the gain value at this stage is greater than the gain of the resonant frequency point. At this time, the input voltage control command is added with a variation ΔV. The proportional-integral-derivative controller is to stabilize the output voltage. It will increase the switching frequency and decrease the gain close to the resonance frequency.

digit Proportional-integral-derivative controller :

Please also refer to FIGS. 11a to 11b, wherein FIG. 11a shows a block diagram of proportional-integral-derivative system control, and FIG. 11b shows a flowchart of an incremental proportional-integral-derivative program.

The proportional-integral-derivative controller is a common application tool in industrial control applications. The principle of the proportional-integral-derivative controller is to linearly combine the error amount using the three parts of proportional, integral, and derivative into a control quantity, and then control the If the command value is x ( t ), the actual output is y ( t ), and the error value is e ( t ), and the controlled body has a single input u ( t ), then The output is obtained as shown in equation (18).

(18)

(18)

The digital control system processes the input and output signals at discrete intervals of a fixed interval, and calculates the output control amount according to the error amount between the sampling point and the input command, which is different from the continuous proportional-integral (equation-integral) of equation (18) and derivative, PID) control algorithm, the digital proportional-integral-derivative controller needs to use the discretization method to calculate the digital sample value, use Euler integration method and difference method to approximate integration and differentiation, and express equation (18) as discrete The form is shown in equation (19).

(19)

(19)

Among them, K _p , K _I , K _D are proportional, integral, and differential constants respectively, T is the sampling period, e ( n ) is the current error of the system, e ( n -1) is the previous error of the system, n is Sampling signal.

If the digital proportional-integral-derivative control shown in equation (19) is implemented by a microprocessor, the output is related to the entire past state due to the integral term, which is the accumulation of the entire past. Therefore, the problem of integral saturation needs to be considered. When the system continues to have an error in a fixed direction, the integration term will continue to accumulate until the maximum value that the accumulator can represent. At this time, the integration amount has long exceeded the upper and lower limits of the operating frequency of the switch. Therefore, the range of values that can be expressed by considering the memory width of the microprocessor is limited, and in order to reduce the amount of operation of the microprocessor and improve the operation efficiency, the present invention uses incremental proportional-integral and derivative , PID) control, the output of the incremental proportional-integral-derivative is only related to the error of the current, the previous and the previous two, so there will be no problem of integral saturation. The incremental proportional-integral-derivative is like the equation (20 ) As shown.

(20)

(20)

Where A = e ( n ) -e ( n -1), B = e ( n ), C = e ( n )-2 e ( n -1) + e ( n -2).

As shown in Figure 11b, the output command value is subtracted from the sampled value to obtain an error amount e ( n ), and then the previous error amount e ( n- 1) and the previous two error amounts e ( n- 2) After substituting equation (20) together for calculation, A , B, and C can be obtained, and then multiplied by K _P , K _I, and K _D in sequence, and then added, to obtain an output variation Du , proportional-integral- The differential output result PID _OUT is equal to Du plus the previous cycle amount, and then compared with the upper and lower limits of the period. If the output result is less than the lower limit Period _min or greater than the upper limit Period _max , the output result is equal to the lower limit or upper limit respectively, and finally The result is output to the PWM generator.

Experimental results and comparison of the present invention and conventional technology:

The circuit specifications of the LLC resonant converter developed by the present invention are shown in Table 2.

Table 2 parameter specification Input rated voltage V _in 380～400 V Maximum output power 率P _out 480 W Output voltage V _out 48 V Maximum output 流I _{out, max} 10 A First resonant frequency 率f _{r 1} 96 kHz _{Electricity 路切} Frequency conversion率f _sw 50 kHz ~ 200 kHz

The specifications of the main component parameters of the LLC resonant converter designed according to the specifications are shown in Table 3.

table 3 element Specifications S ₁ , S ₂ MOSFET: IPP60R099P6 650V/109A, R _{sd (on)} 99mW, C _oss 140pF D ₁ , D ₂ Schottky diode: STPS20150CT 150V/20A Transformer Core: TDK PQ35/35, Turns ratio n = 4, N _p = 28 (AWG19´1), N _s = 7 (0.32mm´16) L _r , L _m , C _r 28mH, 640mH, 100nF k _l , K 0.04375, 22.857

The experimental results prove the feasibility and correctness of the proposed system and control method. The verification items include the ZVS effect of the LLC resonant converter, digital frequency conversion control, and input voltage regulation. Finally, the actual measured experimental waveforms and data are added Explain the analysis.

Please refer to FIGS. 12a-12b together, wherein FIG. 12a shows a waveform diagram of zero voltage switching light load (0.5 A), and FIG. 12b shows a waveform diagram of zero voltage switching heavy load (10 A).

It can be seen from the measured waveform that the designed converter can achieve ZVS switching at both light load (0.5 A) and heavy load (10 A), where V _{GS 1} is 10V/div and V _{DS 1} is 200V/div.

Please refer to Figures 13a~13c together, where Figure 13a shows the output voltage 48 V resonance frequency 96 kHz at light load (0.5 A) waveform, and Figure 13b shows the output voltage 48 V resonance frequency 96 kHz at medium load (5 A) Waveform, Figure 13c shows the waveform of the output voltage 48 V resonance frequency 96 kHz under heavy load (10 A).

As shown in the figure, the waveforms of light load (0.5 A), medium load (5 A) and heavy load (10 A) under different loads are measured. The gate voltage V _GS and the cross voltage V _DS across the power components are measured. 3. Resonant current i _Lr , where V _{GS 1} is 10V/div, V _{DS 1} is 200V/div, i _Lr is 5A/div, 10ms/div, it can be seen that i _{Lr is} close to a sine wave, confirming the operation of LLC resonant converter At the resonance frequency.

Please refer to FIGS. 14a to 14b together, wherein FIG. 14a shows the efficiency curve at different resonance inductance values (640 μH, 380 μH and 250 μH) operating at the resonance point, and FIG. 14b shows the present invention can be used with the load Comparison graph of the efficiency curve of the variable input voltage and fixed voltage operation.

As shown in Figure 14a, the effect of the magnetizing inductance on the efficiency is tested. In this test, three different magnetizing inductances (640 μH, 380 μH, and 250 μH) are used to compare the effect of the magnetizing inductance on the efficiency. Under conditions, the greater the magnetizing inductance, the higher the efficiency.

The invention realizes the design of an LLC resonant converter that can adjust the input voltage according to the load change in a digital control mode. The output voltage of the LLC resonant converter is 48V and the current can reach up to 10A.

As shown in Fig. 14b, this test uses 640 μH magnetizing inductance L _m to compare the input voltage of 380V, 390V, 400V and the input voltage of the present invention that can be adjusted with load changes. It can be seen from the figure that the efficiency of the present invention operating at the resonance frequency point in the range of light load to full load is better than the operation efficiency of the other three fixed input voltages, especially at light load (0.5A), the maximum efficiency is improved by 1.9%, full load (10A) increased by 1.01%. When the load changes, the present invention can adjust the DC input voltage to keep the LLC resonant converter operating near the resonant frequency point, so as to achieve the purpose of high efficiency operation over the entire load output range. The experimental results show that in the controllable input voltage range of 380V ~ 400V, the conversion efficiency of the light load (0.5A) and full load (10A) of the present invention is 91.4% and 95.86%, and the highest efficiency can reach 96.7%.

The present invention also actually develops a 480W prototype circuit to verify the correctness of theoretical analysis, the feasibility and performance improvement of the proposed method, as shown in FIG. 15.

With the design disclosed above, the present invention has the following advantages:

1. The LLC resonant converter disclosed in the present invention can adjust the input voltage according to the load variation. The DC resonant converter can be adjusted to keep the LLC resonant converter operating near the resonant frequency point to achieve a high output range throughout the load Purpose of efficiency operation.

2. The LLC resonant converter disclosed in the present invention can adjust the input voltage according to load changes. Under the condition of ZVS, the larger the magnetizing inductance, the higher the efficiency.

3. The invention discloses an LLC resonant converter that can adjust the input voltage according to load changes. The conversion efficiency of light load (0.5A) and full load (10A) is 91.4 within a controllable input voltage range of 380V ~ 400V. % And 95.86%, the highest efficiency can reach 96.7%.

The disclosure of the present invention is a preferred embodiment, and any part of the changes or modifications that are derived from the technical idea of the present invention and easily inferred by those skilled in the art do not deviate from the scope of the patent right of the present invention.

In summary, regardless of the purpose, means and effectiveness of the present invention, the present invention is showing that it is very different from the conventional technical characteristics, and its first invention is in practical use, and it is also in compliance with the patent requirements of the invention. Pray for a patent as soon as possible to benefit the society and feel virtuous.

Half-bridge switching circuit 100 capacitor-inductor series circuit 110 transformer 120 first diode 130 second diode 140 output capacitor 150 load resistance 160 feedback circuit 170 voltage divider circuit 171 optocoupler circuit 172 control unit 180 analog to digital Converter 181 Filter operation function module 182 Pulse width modulation module 183

FIG. 1 is a block diagram of an embodiment of the LLC resonant converter capable of adjusting input voltage according to load variation according to the present invention. FIG. 2a is a schematic diagram of a half-bridge LLC resonant converter. FIG. 2b shows an equivalent circuit diagram of the series resonant converter switching to the primary side. Figure 2c shows the frequency response curve under different Q values. FIG. 2d is a schematic diagram of a linear dual-port model for converting the nonlinear circuit of FIG. 2a into an LLC series resonant circuit. FIG. 2e shows an equivalent circuit diagram of the LLC resonant tank. FIG. 3 shows the voltage gain and normalized frequency response of the LLC resonant circuit at different Q values. Figure 4 shows a typical waveform of the LLC resonant converter operating in Region-1. Figure 5 shows a typical waveform of the LLC resonant converter operating in Region-2. FIG. 6 shows the waveform of the LLC resonant converter at the first resonance frequency. Fig. 7 shows the operating frequency variation range under different K values. FIG. 8 is a schematic diagram of a control mechanism that can adjust the input voltage according to the load variation provided by the present invention. 9 is a schematic diagram of a digitally controlled LLC resonant converter. FIG. 10a shows a flowchart of the overall firmware program of the present invention using the dsPIC33FJ16GS502 microcontroller. FIG. 10b shows a flowchart of the ADC interrupt routine of the present invention using the dsPIC33FJ16GS502 microcontroller. FIG. 10c shows a flow chart of the subprogram of the present invention which can adjust the input voltage as the load changes using the dsPIC33FJ16GS502 microcontroller. FIG. 11a shows a block diagram of proportional-integral-derivative system control. Figure 11b shows a flowchart of the incremental proportional-integral-derivative formula. Figure 12a shows a waveform diagram of zero voltage switching light load (0.5 A). FIG. 12b shows a waveform diagram of zero voltage switching heavy load (10 A). Figure 13a shows the waveform of the output voltage 48 V resonance frequency 96 kHz at light load (0.5 A). Figure 13b shows the waveform of the output voltage 48 V resonance frequency 96 kHz at medium load (5 A). Figure 13c shows the waveform of the output voltage 48 V resonance frequency 96 kHz under heavy load (10 A). Fig. 14a shows the efficiency curve of operation at different resonance inductance values (640 μH, 380 μH and 250 μH) at the resonance point. FIG. 14b is a comparison diagram of the efficiency curve of the present invention that can adjust the input voltage according to the load variation and the fixed voltage operation. 15 shows a physical photograph of an embodiment of the LLC resonant converter of the present invention.

Half-bridge switching circuit 100 capacitor-inductor series circuit 110 transformer 120 first diode 130 second diode 140 output capacitor 150 load resistance 160 feedback circuit 170 voltage divider circuit 171 optocoupler circuit 172 control unit 180 analog to digital Converter 181 Filter operation function module 182 Pulse width modulation module 183

Claims (5)

An LLC resonant converter capable of adjusting input voltage according to load fluctuations has: a half-bridge switching circuit with two input terminals to be coupled to the positive and negative terminals of an input voltage, and two control terminals to respectively receive a first drive signal Coupled to a second drive signal, and an output terminal to be coupled to the positive terminal when the first drive signal exhibits an active potential and to the negative terminal when the second drive signal exhibits an active potential; a A capacitor-inductor series circuit, one end of which is coupled to the output end of the half-bridge switching circuit; a transformer having a main coil and a primary coil, one end of the main coil is connected to the other of the capacitor-inductor series circuit One end is coupled, the other end of the main coil is coupled to the negative end of the input voltage, the secondary coil has a first output end, a second output end, and a center tap contact; a first The diode has a first anode and a first cathode, the first anode is coupled to the first output terminal, the first cathode is coupled to a voltage output terminal; a second diode has A second anode and a second cathode, the second anode is coupled to the second output terminal, the second cathode is coupled to the voltage output terminal; an output capacitor is coupled to the voltage output terminal and the Between the center tap contacts; a load resistor coupled between the voltage output terminal and the center tap contact; a feedback circuit for generating a feedback signal based on a voltage across the load resistor; and a The control unit is used for performing a proportional-integral-derivative operation according to the difference between the voltage value of the feedback signal and a preset voltage value to determine a PWM operating frequency, and generating the first driving signal and the PWM operating frequency according to the PWM operating frequency The second driving signal, and when the PWM operating frequency is lower than a preset resonant frequency, the input voltage is increased by a voltage difference, and when the PWM operating frequency is higher than a preset resonant frequency, the input voltage is reduced The voltage difference. As described in item 1 of the patent application scope, an LLC resonant converter capable of adjusting the input voltage according to load changes, wherein the feedback circuit includes a voltage divider circuit and an optical coupling circuit. As described in Item 1 of the patent application scope, an LLC resonant converter capable of adjusting the input voltage with load changes, wherein the control unit includes an analog-to-digital converter to perform an analog-to-digital conversion operation on the feedback signal to generate a The first input digital signal. As described in item 3 of the patent application scope, an LLC resonant converter capable of adjusting the input voltage according to load changes, wherein the control unit includes a filter operation function module to perform a filter operation on the first input digital signal to generate a first Two input digital signals, and the control unit performs the proportional-integral-derivative operation according to the difference between the second input digital signal and the preset voltage value. As described in Item 1 of the patent application scope, an LLC resonant converter capable of adjusting the input voltage according to load changes, wherein the control unit includes a pulse width modulation module to provide the first driving signal and the second driving signal.

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