CN107302512B - 用于控制组合无线信号的系统和方法 - Google Patents

用于控制组合无线信号的系统和方法 Download PDF

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CN107302512B
CN107302512B CN201710404796.0A CN201710404796A CN107302512B CN 107302512 B CN107302512 B CN 107302512B CN 201710404796 A CN201710404796 A CN 201710404796A CN 107302512 B CN107302512 B CN 107302512B
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J.D.特里
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2614—Peak power aspects
    • H04L27/2615—Reduction thereof using coding
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2614—Peak power aspects
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J11/00—Orthogonal multiplex systems, e.g. using WALSH codes
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00—Arrangements affording multiple use of the transmission path
    • H04L5/0001—Arrangements for dividing the transmission path
    • H04L5/0003—Two-dimensional division
    • H04L5/0005—Time-frequency
    • H04L5/0007—Time-frequency the frequencies being orthogonal, e.g. OFDM(A), DMT
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2614—Peak power aspects
    • H04L27/2621—Reduction thereof using phase offsets between subcarriers

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Abstract

一种用于控制组合波形的方法，该组合波形表示至少两个具有正交频率复用信号分量的信号的组合，该方法包括：接收定义该至少两个信号的信息；将定义每个信号的信息变换为具有正交频率复用信号分量的表示，使得至少一个信号具有该相同信息的至少两个替换表示，以及使用该至少两个替换表示以至少两种不同方式来组合该变换后的信息以定义各自不同的组合；针对至少一个标准分析该各自不同的组合；以及输出相应的组合波形或者定义该波形的信息，其表示来自基于该分析选择的至少两个信号中的每一个的变换后的信息的所选择的组合。

Description

用于控制组合无线信号的系统和方法
本申请是申请日为2010年09月24日、申请号为201080038040.5、发明名称为“用于控制组合无线信号的系统和方法”的发明专利申请的分案申请。
技术领域
本发明涉及射频信号的无线通信的领域。更具体地说，本发明涉及控制组合信号以例如降低其峰值对平均功率比或接收机处的推断差错。
背景技术
移动无线通信的常见信号格式是正交频域复用或OFDM、以及紧密相关的格式，例如正交频域多址(OFDMA)。对于在OFDM信道上传送的信号，其在频域中特征为一束窄的相邻子信道，而在时域中特征为每个具有时间T、每个由保护间隔ΔT(参见图1)分隔开的相对缓慢的一系列OFDM码元。在每个码元之前的保护间隔内是循环前缀(CP)，其包括在时间上被循环移位的码元周期内的相同信号。该CP被设计为在存在多路径(即，从陆地上的大物体(例如高建筑物、山等)反射的射频信号)时降低接收信号对精确时间同步的敏感性。如果以微小时间延迟(小于△T)接收给定码元，则该给定码元仍将被无差错地接收。除了与OFDM“有效载荷”相关的数据码元外，典型地还有“前导码”信号，其建立定时和其它标准。前导码可具有其自己的CP(未在图1中示出)。
在OFDM中，选择子载波频率以使子载波彼此正交，意味着在子信道之间的串扰被消除并且不需要子载波间保护频带。这大大简化了发射机和接收机二者的设计；与传统FDM不同，不需要用于每个子信道的单独滤波器。正交性要求子载波间距是△f＝k/(TU)赫兹，其中TU秒是有用码元持续时间(接收机侧窗口大小)，并且k是正整数，典型地等于1。因此，在N个子载波的情况下，总通带带宽将是B≈N·△f(Hz)。正交性也允许高频谱效率，其中总码元速率接近奈奎斯特速率。几乎可使用整个可用频带。OFDM一般具有几乎“白色”的频谱，给予它对于其他同信道用户的良好电磁干扰性能。
当组合两个OFDM信号时，结果通常是非正交的信号。虽然被限制到单个OFDM信号的频带的接收机一般不被信道外信号所影响，当这样的信号通过普通的功率放大器时，由于模拟系统组件的固有非线性，存在着相互作用。
OFDM需要在接收机和发射机之间的非常准确的频率同步；在频率偏差的情况下，子载波将不再正交，引起载波间干扰(ICI)，即在子载波之间的串扰。频率偏移典型地由失配的发射机和接收机振荡器引起，或者由因为移动导致的多普勒移位(shift)而引起。尽管多普勒移位自己可被接收机补偿，但是当其与多路径组合时，情况将恶化，因为反射将出现在各种频率偏移处，这更难纠正。
正交性允许在接收机侧使用快速傅里叶变换(FFT)算法，在发送机侧使用逆FFT(IFFT)实现高效的调制器和解调器。尽管FFT算法比较高效，但是其具有适度的计算复杂性，这可能是限制因素。
OFDM的一个关键原则是：由于低码元速率调制方案(即，其中码元与信道时间特性相比相对较长)遭受多路径传播引起的码元间干扰较少，因此并行传送多个低速率流代替单个高速流是有利的。因为每个码元的持续时间长，因此在OFDM码元之间插入保护间隔是可行的，从而消除了码元间干扰。保护间隔还消除了对脉冲整形滤波器的需要，并且它降低了对时间同步问题的敏感性。
在保护间隔期间传送的循环前缀由被复制进保护间隔中的OFDM码元的末端组成，并且在OFDM码元之前传送保护间隔。保护间隔由OFDM码元末端的副本组成的原因是使得接收机在其使用FFT执行OFDM解调时对多路径的每一个将在整数个正弦周期上进行积分。
如果子信道足够窄带，即如果子信道的数量足够大，则频率选择性信道状态的影响，例如多路径传播引起的衰落，可被认为是在OFDM子信道上恒定(平坦)的。与传统单载波调制相比，这使得在OFDM中在接收机处均衡简单得多。均衡器只须将每个检测的子载波(每个傅里叶系数)乘以恒定的复数或不常改变的值。因此，接收机一般能够容忍信号的这种修改，而不需要传送显式信息。
总是结合信道编码(前向纠错)使用OFDM，并且OFDM几乎始终使用频率和/或时间交织。频率(子载波)交织增大了对像衰落那样的频率选择性信道状态的抵抗能力。例如，当信道带宽的一部分衰落时，频率交织保证将由于在带宽的衰落部分中的那些子载波导致的比特差错散布在比特流中而不是被集中。类似地，时间交织保证在比特流中原来靠近在一起的比特在时间上被远远分开地传送，因此减少当以高速传播时将发生的严重衰落。因此，类似于均衡本身，接收机通常容忍一定程度的这种类型的修改，而不增加作为结果的差错率。
OFDM信号是通过逆(快速)傅里叶变换(IFFT)从数字基带数据生成的，这在计算上较为复杂，并且如下面将讨论的，对于包括全部范围的码元的集合生成具有相对高的峰值与平均功率比(PAPR)的作为结果的信号。该高PAPR继而通常导致对于功率放大器(PA)的增加的购置(acquisition)成本和运行成本，以及通常导致与针对具有较低PAPR的信号设计的系统相比更大的非线性失真。除了其他的以外，该非线性还导致限幅失真和互调(IM)失真，它们具有消耗功率、引起带外干扰、以及可能引起带内干扰并且在接收机处比特差错率(BER)相应增加的影响。
在传统类型OFDM发射机中，信号发生器对输入信息比特序列执行纠错编码、交织以及码元映射以产生传输码元。该传输码元在串并(S/P)转换器中进行串并转换并且被转换为多个并行信号序列。该S/P转换后的信号在IFFF单元进行快速傅里叶逆变换。该信号进一步在并串(P/S)转换转换器中进行并串转换，并且被转换为一信号序列。然后，由保护间隔(GI)添加单元添加保护间隔。然后，该格式化信号被上变频到射频，在功率放大器处被放大，并最终通过无线电天线作为OFDM信号被发送。
另一方面，在传统类型OFDM接收机中，射频信号被下变频到基带或中频，并且在保护间隔去除单元处从接收的信号去除保护间隔。然后，该接收信号在S/P转换器进行串并转换，在快速傅里叶变换(FFT)单元进行快速傅里叶变换，以及在P/S转换器进行并串转换。然后，输出该解码比特序列。
传统地，每个OFDM信道具有它自己的传送链，其在功率放大器(PA)和天线元件中结束。不过，在某些情况下，人们可能希望使用相同的PA和天线传送两个或更多单独的OFDM信道，如图2所示。这可允许在有限数量的基站塔上具有额外通信带宽的系统。给出对于额外用户和额外数据速率的驱动力，这是非常需要的。使用如图2所示的二级上变频过程可在中频组合这两个信道。虽然图2示出了真实的基带信号的放大，但是通常有具有同相和正交上变频(未示出)的复二相信号。图2也未示出数字和模拟信号之间的分界线。基带信号通常是数字的，而RF传送信号通常是模拟的，在这些级之间某处具有数模变换。
考虑两个相似的信道，每个具有平均功率P0和最大瞬时功率P1。这对应于峰值与平均功率比PAPR＝P1/P0，其通常用dB而被表示为PAPR[dB]＝10log(P1/P0)。对于组合信号，平均功率是2P0(增加了3dB)，但最大瞬时功率可能高达4P1，增加了6dB。因此，组合信号的PAPR可增加多达3dB。如果来自两个信道的信号碰巧具有同相峰值，则将出现该最大功率。这可能是罕见的瞬时事件，但通常所有传送组件的线性动态范围必须为这种可能性而进行设计。非线性将产生互调产物，这将使信号劣化以及使信号扩展到频谱的不希望的区域中。这进而可能需要滤波，并且在任何情况下将可能减小系统的功率效率。
需要增加线性动态范围以处理该较高PAPR的组件包括例如数模转换器，其必须具有更多数量的有效比特以处理更大的动态范围。但更重要的是功率放大器(PA)，因为PA一般是发射机里最大的并且功率消耗最多的组件。尽管有时可以保留具有仅在一小部分时间内使用的额外动态范围的组件，但是这浪费并且无效率，并且在可能时应避免。具有更大动态范围的放大器通常比具有较低动态范围的放大器更贵，并且对于可比较的输入和输出常常具有更高的静止电流消耗和更低的效率。
峰值与平均功率比(PAPR)的问题是OFDM和相关波形中的公知的常见问题，因为OFDM和相关波形由多个很近地间隔开的子信道构成。存在许多经典的策略来减小PAPR，它们在综述文章中得到了陈述，所述综述文章例如为“Directions and Recent Advances inPAPR Reduction Methods”，Hanna Bogucka，Proc.2006IEEE International Symposiumon Signal Processing and Information Technology，第821-827页，通过引用而将其合并于此。这些PAPR减小策略包括限幅和滤波、编码、音调(tone)保留、音调注入、活动星座扩展和多种信号表示技术，例如部分传送序列(PTS)、选择性映射(SLM)和交织。这些技术能实现显著的PAPR减小，但代价是传送信号功率增加、比特差错率(BER)增加、数据速率受损、计算复杂性增加等。此外，这些技术中很多都需要与信号本身一起传送额外的附带信息(有关信号变换)，以使接收信号被正确地解码。这样的附带信息降低了该技术的通用性，特别是对于想要简单的移动接收机从多种基站发射机接收信号的技术而言。在兼容的程度上，可结合下面讨论的技术使用在Bogucka中公开的以及在本领域中以其他方式公知的技术。
在OFDM传输方案中解决PAPR(峰值与平均功率比)问题的各种努力包括频域交织方法、限幅滤波方法(参见例如X.Li和L.J.Cimini，“Effects of Clipping and Filteringon the Performance of OFDM”，IEEE Commun.Lett.，第2卷第5期第131-133页，1998年5月)、部分传送序列(PTS)方法(参见例如L.J Cimini和N.R.Sollenberger，“Peak-to-Average Power Ratio Reduction of an OFDM Signal Using Partial TransmitSequences”，IEEE Commun.Lett.，第4卷第3期第86-88页，2000年3月)、循环移位序列(CSS)方法(参见例如G.Hill和M.Faulkner，“Cyclic Shifting and Time Inversion ofPartial Transmit Sequences to Reduce the Peak-to-Average Ratio in OFDM”，PIMRC2000，第2卷第1256-1259页，2000年9月)。此外，为了在使用非线性传输放大器时改善OFDM传输中的接收特性，提出了一种使用最小限幅功率损失方案(MCPLS)的PTS方法以使得由传输放大器限幅的功率损失最小(参见例如Xia Lei，Youxi Tang，Shaoqian Li，“A MinimumClipping Power Loss Scheme for Mitigating the Clipping Noise in OFDM”，GLOBECOM 2003，IEEE，第1卷第6-9页，2003年12月)。MCPLS也适用于周期性移位序列(CSS)方法。
在部分传送序列(PTS)方案中，预先为各自子载波确定的相位旋转值的适当集合是从多个集合中选择的，并且选择的相位旋转集合用于在信号调制之前旋转每个子载波的相位从而降低峰值与平均功率比(参见例如S.H.Muller和J.B.Huber，“A Novel PeakPower Reduction Scheme for OFDM”，Proc.of PIMRC'97，第1090-1094页，1997年；以及G.R.Hill，Faulkner和J.Singh，“Deducing the Peak-to-Average Power Ratio in OFDMby Cyclically Shifting Partial Transmit Sequences”，Electronics Letters，Vol.36,No.6,16.sup.th March,2000(第36卷第6期，16.sup.th，2000年3月16日)。
需要一种用于以不使接收信号劣化或不需要传送附带信息的方式降低组合OFDM信号的PAPR的实际方法和相关装置。
下列专利(其每一个通过引用而被明确地合并于此)涉及峰值功率比考虑：
7,535,950 Blind selected mapping techniques for crest factorreduction of forward link CDMA signals
7,499,496 Transmitter and transmission controlling method
7,496,028 Apparatus and method for minimizing PAPR in an OFDMcommunication system
7,467,338 Apparatus and method for generating an error signal
7,463,698 Transmitter and transmission control method
7,443,904 Scrambling system and method for peak power reduction inMC-CDMA,and recording medium for storing corresponding program
7,376,202 OFDM peak-to-average power ratio reduction by combinedsymbol rotation and inversion with limited complexity
7,376,074 Apparatus and method for transmitting and receiving sideinformation of a partial transmit sequence in an OFDM communication system
7,349,817 Apparatus and method for reducing peak-to-average powerratio in a broadband wireless communication system
7,345,990 Method and apparatus for performing digital communications
7,342,978 Method and apparatus for PAPR reduction of an OFDM signal
7,340,006 Apparatus and method for reducing PAPR in OFDMcommunication system
7,321,629 Method and apparatus for protecting and transmitting theside information related to peak-to-average power ratio reduction in amulticarrier system
7,315,580 Method and apparatus for high-order PAPR reduction of anOFDM signal
7,292,639 Method and apparatus for peak to average power ratioreduction for orthogonal frequency division multiplex systems
7,002,904 Method and apparatus for reducing peak power in partialtransmit sequence OFDM
6,925,128 Method and apparatus for reducing a peak-to-average powerratio in an orthogonal frequency division multiplex signal7,535,950 Blindselected mapping techniques for crest factor reduction of forward link CDMAsignals
7,499,496 Transmitter and transmission controlling method
7,496,028 Apparatus and method for minimizing PAPR in an OFDMcommunication system
7,467,338 Apparatus and method for generating an error signal
7,443,904 Scrambling system and method for peak power reduction inMC-CDMA,and recording medium for storing corresponding program
7,376,074 Apparatus and method for transmitting and receiving sideinformation of a partial transmit sequence in an OFDM communication system
7,349,817 Apparatus and method for reducing peak-to-average powerratio in a broadband wireless communication system
7,345,990 Method and apparatus for performing digital communications
7,342,978 Method and apparatus for PAPR reduction of an OFDM signal
7,340,006 Apparatus and method for reducing PAPR in OFDMcommunication system
7,339,884 STBC MMO-OFDM peak-to-average power ratio reduction bycross-antenna rotation and inversion
7,321,629 Method and apparatus for protecting and transmitting theside information related to peak-to-average power ratio reduction in amulticarrier system
7,315,580 Method and apparatus for high-order PAPR reduction of anOFDM signal
7,301,891 Apparatus and method for reducing peak-to-average powerratio in an orthogonal frequency division multiplexing system
7,292,639 Method and apparatus for peak to average power ratioreduction for orthogonal frequency division multiplex systems
7,002,904 Method and apparatus for reducing peak power in partialtransmit sequence OFDM
6,925,128 Method and apparatus for reducing a peak-to-average powerratio in an orthogonal frequency division multiplex signal
5,302,914 Method and apparatus for reducing the peak-to-average powerin multi-carrier RF communication systems
20090147870 Method for solving High PAPR problem of MCM communicationsystem using unitary transform
20090097579 Apparatus and method for reducing PAPR In An OFDM system
20090086848 Apparatus and method for reducing peak-to-average powerratio in a wireless communication system
20090074093 Apparatus for transmitting data using carriers and methodthereof
20090060070 Apparatus and Method for Peak Suppression in WirelessCommunication Systems
20090052577 Peak to average power ratio reduction
20090052561 Methods and apparatus for generating and communicatingwireless signals having pilot signals with variable pilot signal parameters
20090034407 Receiver-site restoration of clipped signal peaks
20090011722 Methods and apparatus for wirelessly communicatingsignals that include embedded synchronization/pilot sequences
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发明内容
当具有不同载波频率的多个无线电信号被组合以便传输时，由于峰值同相组合的可能性，该组合信号通常具有增加的峰值与平均功率比(PAPR)，需要以低平均效率工作的更大的射频功率放大器(PA)。正交频域复用(OFDM)信道的数字组合的PAPR可通过以下操作而被降低：在存储缓冲器中存储给定码元周期的时域OFDM信号，并且执行至少一个OFDM信号的循环时间移位，以便选择与该组合多信道信号的降低的PAPR相对应的时间移位。这可应用于基带上的信号或上变频后的信号。仿真表明可获得PAPR的若干分贝的降低而没有降低系统性能。不需要向接收机传送附带信息。
本系统和方法的优选实施例试图通过在存储缓冲器中存储给定码元周期的时域OFDM信号，并且执行至少一个OFDM信号的循环时间移位，以选择与该组合多信道信号的期望的PAPR对应的时间移位，来控制PAPR。多数情况下，希望将PAPR降到最小，但是这不是对该技术的限制，并且选择的时间移位可基于其他标准。
应注意的是每个OFDM信号可根据已知方案进行预处理，因此每一个自己可能已经被处理以降低固有PAPR，尽管最好将信号的任何预处理与组合信号的处理相协调以获得最佳成本和益处。例如，在每个具有高PAPR的两个单独信号将被组合的情况下，如果峰值异相相加从而抵消，则可获得所得到的具有降低的PAPR的信号。因此，修改输入的OFDM信号的最初的未经协调的努力可能有有限的益处。
应注意的是本系统试图组合独立格式化的OFDM，其一般针对不同的接收机或者接收机的集合，并且这些集合通常彼此不进行协调。例如，在蜂窝收发器系统中，基站可服务于成百上千部蜂窝电话，每部电话监控单个OFDM广播信道，并且基站为多个OFDM信道服务。特别注意，OFDM子载波的每个组是正交的，但是分离的OFDM信号及其子载波一般彼此不正交。OFDM信号可以在邻近的或者被移动了的信道中，因此在单个码元周期期间可发生OFDM信号之间的相对相位变化。因此，必须在整个码元周期上考虑PAPR。
事实上，根据本方法的另一个实施例，被分析以进行最优化的不是信号的PAPR，而是接收机处的推断差错。如果复合信号的PAPR仅对码元周期的一小部分较高使得在那个时间PA使信号失真或限幅，而在大多数其它时间组合信号完全处于规范内，则结果可能是可导致低差错概率的可接受的传输。事实上，在某些情况下，差错概率可能比对于具有更低绝对峰值的信号的差错概率更低。因此，通过采用本身可包括对于针对特定接收机的特定通信信道缺陷以及多普勒频移(其可例如通过分析返回路径特性来确定)的余量或在可能的变化的范围上的余量的型号的接收机作为发射机信号处理路径的一部分，可获得比通过简单最小化PAPR更好的性能。
另一个选择是以一种方式在全部或部分周期期间修改OFDM信号，该方式背离例如为IEEE-802OFDM标准、WiFi、WiMax、DAB、DVB、蜂窝通信、LTE信号等等的标准协议，但其基本上不增大标准或特定接收机的预测BER。例如，如果PAPR对码元周期的一小部分较高，使得如果在该码元周期的一部分期间去除或修改了一个或多个子载波，则PAPR将是可接受的，并且接收机处的信号将有足够的信息以使用标准接收机来解码而不显著增加BER，因此发射机能执行这样的修改而不需要传送解调所必需的、标识所述修改的附带信息。另一个可能的背离是例如在接收机的容差内对信号进行频移(这略微违反正交性标准)以便在等效于频移的多普勒移位的范围内操作。
考虑如图2组合的两个OFDM信号。为简单起见，将信号1(S1)称为参考信号，将信号2(S2)称为修改信号。在每个OFDM码元周期期间，每个信号的基带数字数据比特将被存储在存储器中。假定前导码已经被剥离，但是循环前缀CP仍然存在。如用于本发明的一个实施例的图3所示，参考信号S1的比特被存储在先入先出(FIFO)移位寄存器(SR)中。修改信号S2的相应比特被存储在循环移位寄存器(CSR)中，循环移位寄存器(CSR)被配置为使得可以在程序的控制下旋转所包含的数据。两个信号的数据首先上变频到中频(IF)，然后被组合(相加)，同时以增大超过数字数据速率的采样频率保持数字格式。组合IF信号然后进行PAPR测试，以确定峰值功率水平是否是可接受的，或在其他实施例中确定是否满足其他标准。这可对应于例如9dB的PAPR。如果通过测试，则读出该组合OFDM码元的数据比特，以随后将其重新组装为完整的OFDM帧并上变频为全RF，以便进一步在PA中放大和传输。根据另一个实施例，组合数据的组合OFDM表示本身是上变频的来源。
更一般地，一旦确定用于达到希望标准的参数变换(相对时间移位)，就随后根据该参数或作为结果的表示来形成(formulate)最终信号，其可以是基带信号或其转换形式的数字数据比特；在后一种情况下，系统可对数据执行一系列变换(其中一些变换是冗余的或失败的)，从而寻找可接受的变换或最佳的变换；一旦发现可接受的变换或最佳的变换，则可不必再次重复该系列变换。同样，恢复到初始数字数据并且重复该确定的一系列变换的选项允许在寄存器中形成稍微不同的表示，例如被简化或预失真以允许在组合测试中考虑模拟部件的性能问题的表示。
更一般地，该技术提供了：给要被组合的每个信号提供一个或多个可接受参数的范围(其可递增地、根据算法(algorithmically)地、随机地或以其他方式改变)、以及针对与一个或多个标准的相符性而测试和/或分析的可能组合的至少一部分，随后使用从更大的一组可用参数中选择的一个或多个参数实现OFDM信号的组合。该参数变化和测试可利用诸如超导逻辑的高速数字电路以串行方式执行，或者在必要时利用具有并行化的较慢逻辑执行，尽管在合适和/或必要时可使用其他技术，包括但不限于光计算机、可编程逻辑阵列、大规模并行计算机(例如，图形处理器，例如nVidia
Figure BDA0001310714980000231
GPU、ATI Radeon R66、R700)等等。例如在大量使用专门化的高速处理器的大量复杂计算的情况下，例如在作为发射机优化的一部分而将大量独立接收机建模的情况下，使用超导数字电路可能是有利的。
在优选实施例中，在对参数范围的测试的任何状态处，如果测试未通过，则反馈控制信号给寄存器，例如CSR，其旋转修改信号S2的数据比特。然后像以前一样将移位后的数据与来自S1的初始存储的数据组合，并重新测试PAPR。这一直重复到PAPR测试通过为止。图4说明了类似的步骤序列，其中明确地示出剥离了前导码并且将其重新附加在末端。应注意的是，在某些情况下，可并行地施加测试，因此不需要严格迭代测试。这进而允许使用低速测试逻辑，尽管复杂度较高。同样，在每个相对时间移位处，还可以考虑次要(secondary)参数。
例如，对于最佳组合的次要考虑可以是带内(非可滤波)互调失真。因此，在每个基本参数变化处，可计算预测的带内互调失真，其被表示为例如功率和/或推断BER。例如可在可接受的PAPR范围内通过施加阈值或优化简单线性组合“成本函数”而将该考虑与PAPR结合。
尽管在该移位和相加过程(SAA)中可以存在一些延迟，但是用于包括所有迭代的全部判决算法的时间必须不超过扩展码元时间T+△T。我们已经在图3和4中描述了串行判决过程。如上面讨论的，在某些情况下，可能优选的是使用具有不同移位的多个CSR和多个并行PAPR测试来并行执行该过程的各部分，以更快地完成该过程。这在图5中说明，其提出了并行存储器(在这里显示为RAM)，每一个具有适当的时间移位，其中选择最小PAPR以发送到RF子系统。在电路速度和复杂性之间的最优权衡将确定优选的配置。
在某些情形下，对最佳组合信号的搜索需要巨大的计算资源。实际上，启发法(heuristics)可用于限制搜索同时仍获得可接受的结果。在PAPR优化的情况下，通常目标是对有限、低概率“最坏情况”码元组合进行测试。如果可得到原始数字数据，则可采用查找表来测试不良组合，所述不良组合然后可根据预定的修改来解决。然而，对于复码元的多路组合，该查找表可能是不可行的。另一方面，可在各个OFDM波形每个中搜索峰值，所述峰值例如为平均值以上6dB，并且仅分析信号的这些部分以确定是否存在与其他OFDM信号的峰值的时间对准；如果峰值在时间上不同步，则推定不可接受的峰值将不会导致最后的组合信号。该方法做出应当在统计上可接受的推定，即以下推定：OFDM波形的仅仅本身为相对峰值的部分将对OFDM信号的组合中的大峰值做出贡献。该方法避免了依序参数变化的连续测试，并且相当简单地避免了二进制阈值条件的最坏情况叠加。
重要的是注意：修改信号的循环移位的码元数据准确地表示与未移位数据相同的码元集合。此外，由于OFDM信号的标准特性，移位码元集合可在没有专门的附带信息并且没有信号完整性的劣化的情况下被传送和接收。因此，具有降低的PAPR的组合OFDM信道应当展现出与初始的未移位版本基本相同的性能。在下面的详细说明部分描述了对此进行确认的一组详细仿真。
虽然这些图集中在对于两个OFDM信道的组合降低PAPR的情况，但该方法不限于两个信道。可以通过循环时间移位然后进行PAPR测试的相似的方法来优化三个或更多信道。
附图说明
图1示出在频域和时域中的正交频域复用信道的典型行为。
图2示出使用双重上变频方法的发射机中的两个OFDM信道的组合。
图3提供了简单框图，其示出可以如何组合两个OFDM信道，其中一个OFDM信道的数据比特可被循环移位，以降低峰值与平均功率比(PAPR)。
图4示出两个OFDM信道的结构，其中一个信道的数据被循环移位以降低PAPR。
图5提供了示出来自OFDM信道的数据的多个移位副本的存储器存储的框图，其中选择对应于最小化PAPR的一个副本。
图6示出在发射机中包含了移位相加算法的仿真通信系统的框图。
图7示出用于图6所示的仿真的发射机中包含的功率放大器的传递函数。
图8绘制出在使用和不使用移位相加算法的情况下，作为信号噪声比(SNR)的函数的、使用正交相移键控(QPSK)的OFDM信号的仿真的比特差错率(BER)。
图9绘制出在使用和不使用移位相加算法的情况下，作为SNR的函数的、使用16正交幅度调制信号(16-QAM)的BER。
具体实施方式
OFDM信道包括很多子信道，每个子信道是窄带信号(图1)。OFDM信道本身具有时变的包络，并且可以表现出相当大的PAPR，典型地为9-10dB。但是，如果两个独立的相似OFDM信道被组合，则作为结果的信号将表现出12-13dB的PAPR，有3dB的增益。这大得不可接受，因为其将需要具有四倍容量的功率放大器来传送平均仅2倍大的组合信号。
因此，优选实施例提供了一种PAPR降低方法，其将两个OFDM信道组合的信号的PAPR从12-13dB降回到原始分量的9-10dB。优选地，完成该～3dB的PAPR降低而不使信号劣化，并且不需要传送接收机将需要用于恢复OFDM码元的任何特殊的附带信息。此外，该算法足够简单，使得它可用任何硬件技术来实现，只要其足够快。
降低PAPR的传统方法集中在组合子信道以及生成不具有过度的PAPR的单个OFDM信道。本技术在某些方面可以被视为部分传送序列(PTM)和选择映射(SLM)的组合。
在传统PTS中，N个码元的输入数据块被分为分离的子块。每个子块中的子载波通过用于该子块的相位因子进行加权。选择该相位因子使得组合信号的PAPR最小化。
在SLM技术中，发射机产生一组显著不同的候选数据块(其全部表示与原始数据块相同的信息)，并且选择对传输最有利的(最低PAPR而没有信号劣化)。
本混合方法针对求和后的载波调制信号组合PTS和SLM的元素。搜索过采样的OFDM波形的不同循环时间移位，并且选择具有最低PAPR的时间移位。一个OFDM信号被用作参考，而一个或多个其他载波调制信号用于以类似于PTS的方式生成时间移位。通过循环前缀长度和过采样速率来确定搜索窗口。
当移位(shift)的可能组合的相位空间显著增大时，可不必探查全部这样的组合。通常，非常高的PAPR值相对较少地发生，使得以高PAPR状态开始的大多数时间移位将趋向于导致PAPR的降低。可依序地或并行地、或者以这两者的某种组合来实现多个信道中的移位。因此，例如，具有在可接受范围内的PAPR的任何组合是可接受的，任何不可接受的PAPR状态出现1％的时间，用于寻找可接受的PAPR的搜索空间一般将小于可能状态的2％。另一方面，如果采用了其他可接受性标准，则更大搜索空间可能是必要的或合适的。例如，假定对于传送更高PAPR信号具有更高的成本，例如功率成本或干扰成本，则形式(formal)最优化可能是合适的。假定没有启发法可用于预测最佳状态，则因此对参数空间的完整搜索对于使成本最低可能是合适的。
这不同于传统方法，其中不同OFDM信道彼此独立，具有单独的发送链而没有相互同步。此外，传统方法直接对基带信号操作。相反，本方法对经过上变频的、合并了两个或更多OFDM信道的组合信号评估PAPR，并且这些信道中的每个信道的码元周期必须是同步的。这不会引起接收机处的问题，其中每个信道被独立地接收和定时。
针对PAPR的一些传统方法基于限幅，但是这不可避免地产生失真和带外生成。一些其他方法避免了失真，但需要必须在接收端解码的特殊变换。它们要么需要发送附带信息，要么涉及对于标准OFDM通信协议的背离。本优选方法不具有这两个缺点。
在蜂窝通信中使用的OFDM信道在带宽上可能达到10或20MHz。然而，这些信道可以位于宽得多的频带中，例如2.5-2.7GHz。因此，可具有由100MHz或更多分开的两个或更多OFDM信道的组合，每个OFDM信道10MHz宽。可以以低到20MS/s的速率对10MHz数字基带信号进行采样，但必须以至少200MS/s的速率对覆盖100MHz的组合数字信号进行采样。
在优选实施例中，以这样的提高的采样速率在数字域中执行信号组合(包括图3中的上变频)。PAPR阈值测试以及CSR控制也以该较高的速率实现。该速率应当足够快以使可以在单个码元时间(几微秒)内执行多个迭代。
为了验证循环(circular)时间移位允许降低组合OFDM信道的PAPR而不降低系统性能的预期，执行了OFDM传送和接收的完整Mont-Carlo(蒙特卡罗)仿真。在图6中概述了该仿真的框图，其表示“SAA评估测试平台(bench)”，并且示出组合频率F1和F2上的OFDM信号的发射机，对该发射机应用用于降低PAPR的SAA算法。在接收端，其被下变频，并且使用标准OFDM接收机恢复F2上的信号。按照这种方式，对信道添加合适的加性白高斯噪声(AWGN)。该仿真还包括几乎线性的功率放大器(PA)的实际的传递函数，其示出了在饱和附近对于线性的偏离(参阅图7)。增益因子对该仿真不重要，因此其未被包括。
PAPR比特差错率(BER)仿真的参数包括以下内容。每个分组包含根据使用的调制类型而在若干OFDM码元周期上调制的800字节的信息。检查QPSK(正交相移键控)和16-QAM(16正交幅度调制)两者。运行每个SNR点直到250个分组差错发生。循环前缀被设置为总码元时间的1/8。频率F1和F2上的载波被充分地间隔开，使得它们的频谱不重叠。过采样速率是因子8。最后，使用了升余弦滤波器，其具有非常陡峭的滚降(rolloff)，采样频率Fs＝160MHz以及频率截止Fc＝24MHz。使用了对于组合OFDM信道的大约9dB的PAPR阈值。
图8示出了对于QPSK调制，在应用和不应用SAA算法的情况下，作为信噪比(SNR)的函数的(即，改变AWGN功率)的BER性能。图9示出了对于16-QAM的对应分析。在两个情况下，在BER上，相对于零移位曲线存在非常小的降低。
定量地分析，使用SAA的纯性能改善对于QPSK是2.35dB，对于16-QAM是2.9dB，如从BER图示推断的那样。例如，如果没有SAA，则BER在8.5dB的输入回退(backoff)处(对于PA)展现出0.03的差错下限(error floor)，而在6.5dB的SAA的情况下，BER展现出相同的差错下限，性能改善将是8.5-6.5＝2dB。
这些仿真不仅确认了SAA算法允许将组合OFDM信道中的PAPR降低～3dB，还确认了在无信号劣化并且不需要在传送信号中发送关于变换的任何特殊附带信息的情况下实现这一降低。
该技术的一个优选实现方式包括使用快速现场可编程门阵列(FPGA)，其具有用于移位寄存器存储器、数字上变频和阈值测试的块。可替换地，可使用超快速数字技术，例如快速单磁通量子(RSFQ)超导电路。当所组合的OFDM信道的数量增大时，需要增大算法速度，或者可替换地并行执行所述处理的一部分。
该方法也可应用于遵循认知无线电的路线(line)的可重构系统，其中根据用户需求和可用带宽，可动态地重新分配要传送的信道。所传送的信道的数量和它们的频率分配两者可在全软件控制下改变。只要全部信道遵循相同的通用码元协议和定时，就可应用一组相似的移位相加算法来维持用于高效传送的可接受的PAPR。

Claims (21)

1.一种用于控制组合波形的方法，该组合波形表示至少两个正交频率复用信号的组合，该方法包括：
接收定义该至少两个信号的信息；
将定义每个信号的信息变换为正交频率复用信号，使得至少一个正交频率复用信号具有相同信息的至少两个替换表示，并且以至少两种不同方式来组合该至少两个替换表示以定义各自不同的组合；
根据至少一种标准，分析该各自不同的组合；
根据所述至少一种标准，基于所述分析，选择各自的组合；以及
输出基于所选择的各自的组合的信息，以定义对应的组合波形，而不输出附带信息，所述附带信息表示在组合波形中信息的恢复所需的组合中的信息的变换。
2.根据权利要求1的方法，其中变换后的信息的第一组合和第二组合关于以下而不同：相对于第二信号的第一信号的频率分量的调制的相对定时。
3.根据权利要求1的方法，其中变换后的信息的第一组合和第二组合关于以下而不同：信号的频率分量的相对相位。
4.根据权利要求1的方法，其中所述信号包括正交频分复用信号。
5.根据权利要求1的方法，其中至少一个信号是正交频分复用流，其符合IEEE 802协议。
6.根据权利要求1的方法，其中至少一个信号包括WiMax信号。
7.根据权利要求1的方法，其中至少一个信号包括LTE信号。
8.根据权利要求1的方法，其中至少一个信号包括DAB和/或DVB信号。
9.根据权利要求1的方法，其中所述分析步骤针对相应组合的动态范围进行分析。
10.根据权利要求1的方法，其中所述分析步骤针对用于所述信号之一的参考接收机设计的预测差错率进行分析。
11.根据权利要求1的方法，其中所述分析步骤针对组合波形的动态范围以及用于所述信号之一的接收机的预测差错率进行分析。
12.根据权利要求1的方法，其中所述分析步骤分析组合波形的限幅失真。
13.根据权利要求1的方法，其中所述输出包括：以射频输出所选择的组合信号。
14.根据权利要求1的方法，其中所述输出包括输出组合信号的中频表示。
15.根据权利要求1的方法，其中所述输出包括输出用于将数字基带信号转换为所选择的组合信号的一组参数。
16.根据权利要求1的方法，还包括对组合波形的中频或射频表示中的至少一个进行预失真。
17.根据权利要求16的方法，其中所述预失真至少部分地补偿组合波形的非线性、传输信道缺陷、以及接收机特性中的一个或多个。
18.根据权利要求16的方法，其中所述预失真补偿功率放大器的非线性失真。
19.根据权利要求1的方法，其中每个信号包括具有循环前缀的正交频域复用信号。
20.根据权利要求1的方法，其中每个信号被作为符合通信协议的正交频分复用信号而接收，修改所述信号中的至少一个以生成所述至少两个替换表示。
21.一种用于组合相应的多个各自信道中的多个信号的方法，每个信号包括作为各自信道中调制的相位和/或幅度的多个正交频率复用信号分量，该方法包括：
接收定义所述多个信号中的每一个的信息；
将所述多个信号表示为所述多个正交频率复用信号分量；
以至少两种不同方式变换至少一个信号的表示，每个变换后的表示表现相同的信息；
分析该多个信号的多个不同组合，多个表示中的每一个包括所述至少一个信号的表示的不同变换后的表示，其中不需要用于所选择的组合中的信息的恢复的附带信息；
基于所述分析选择组合；以及
输出所选择的组合的标识、所选择的组合以及定义所选择的组合的信息中的至少一个。
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US10616025B2 (en) 2020-04-07
US10193729B2 (en) 2019-01-29
CN102687476B (zh) 2017-06-30
WO2012030319A2 (en) 2012-03-08
CA2763134C (en) 2021-01-19
US8582687B2 (en) 2013-11-12
CN102687476A (zh) 2012-09-19
US20170237594A1 (en) 2017-08-17
CN107302512A (zh) 2017-10-27
CA2763134A1 (en) 2010-12-26
US20190173708A1 (en) 2019-06-06
EP2457354A4 (en) 2017-07-05
JP2012531876A (ja) 2012-12-10
US20140056385A1 (en) 2014-02-27
US9641372B2 (en) 2017-05-02
WO2012030319A3 (en) 2012-04-26
US20100329401A1 (en) 2010-12-30
EP2457354B1 (en) 2020-09-09
KR101936533B1 (ko) 2019-04-03
US20160036612A1 (en) 2016-02-04
US9160593B2 (en) 2015-10-13
EP2457354A2 (en) 2012-05-30

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