CN101971484A - 具有宽频率变化范围的电谐振器装置 - Google Patents
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Abstract
本发明公开了一种电谐振器装置,其能够在可变频率ω下操作,所述电谐振器装置至少包括:声波谐振器;第一电路,其并联耦合到所述谐振器,且具有虚部等于的可调节复阻抗,其中C1≤0;第二电路,其并联耦合到所述谐振器和所述第一电路,且具有虚部等于的复阻抗,其中C2<0,ω为所述装置的操作频率。
Description
本申请案涉及且主张2008年3月11日申请的第61/035,437号美国临时专利申请案的优先权,所述美国临时专利申请案出于所有目的以引用方式并入本文。本申请案还主张2008年1月18日申请的第0850327号法国专利申请案的优先权,所述法国专利申请案出于所有目的以引用方式并入本文。
技术领域
本文档涉及具有宽的频率调谐范围或频率变化范围的电谐振器装置,其例如用于产生频率和噪声稳定参考源,因此产生压控振荡器(VCO)。此VCO可例如用于移动通信终端的发射和接收链。本文档还适用于具有宽频带的选择性滤波器的产生,其也应用于移动通信系统中。
背景技术
一般来说,VCO特征在于四个参数:
-相位噪声:此表征振荡器的频谱纯度。振荡器的固有振动以频谱噪声密度的形式量化,所述谱噪声密度在移动远离其基本频率时减小。以dBc/Hz表达的此谱密度的水平是在相对于基本频率的某一频率差提供。良好的相位噪声确保良好的接收灵敏度,以及良好的调制质量。
-频率变化范围:这是振荡器调谐到给定频带中的能力。一般来说,振荡器预期能涵盖对应于已创建系统的标准的所有频带。在频带彼此接近的多标准系统的情形下,大的频率变化范围是优点。
-输出功率额定值:这是振荡器产生的参考信号的功率电平。其电平越高,其相位噪声越有效,且其与系统的其它块的接合越简单。
-消耗:这是VCO操作所需的持续功率。其与系统中可用的DC电压有关。在技术致密化的情况下,电源电压正在减小,其相当大地限制了振荡器提供功率的能力。
这四个参数的优化大体上是基于若干折衷。因此,VCO的相位噪声的改善损害频率变化范围和消耗。此外,高输出功率电平增加VCO的消耗。一般来说,VCO依据为其设计的发射/接收系统的特定特征而具有基于这些约束中的一个或多个的优化性能。为了比较VCO的性能,存在使各种约束联系的数学指示符:品质因数。品质因数越高,可认为VCO越有效。
为制成VCO,使例如通过串联或并联连接到负电阻器的RLC电路(电阻器+电感+电容器)而模型化的谐振器关联到额外的复阻抗元件,其修改VCO的谐振条件以适合于命令。负电阻器意味着一电组件,其至少在某一范围内的行为使得在施加到其端子的电压增加时通过其的电流下降。复阻抗元件例如为例如用变容二极管获得的可变电容,或可变电感性元件。如此产生的VCO的相位噪声首先取决于谐振器和可变复阻抗元件的质量系数的关联,其次取决于用以产生谐振器的负电阻器的晶体管专有的噪声。
为了改善RF系统(尤其为数字系统)中的VCO的振荡器的稳定性,一种技术是使集成VCO具备例如FBAR类型(薄膜体声波谐振器)的BAW谐振器(体声波),或关联到电压可控的可变复阻抗元件的SAW谐振器(表面声波)。因此可能满足与当前移动通信系统兼容的尤其在高操作频率下的稳定性、相位噪声和功率消耗的高约束。
例如BAW或SAW等具有高质量系数的谐振器具有在彼此接近的两个频率下具有突出值的阻抗:串联谐振频率,在所述频率下谐振器的阻抗最低,以及反谐振频率,在所述频率下谐振器的阻抗最高。具备BAW或SAW谐振器的VCO的可变复阻抗元件准许VCO的谐振或反谐振频率变化。
在移动通信系统中,VCO的典型频率变化范围达到约5%。此外,维持良好质量系数是基本的,因为其首先涉及于功能的相位噪声中。2006年10月的IEEE固态电路期刊第41期第10号等人的公开案“使用串联IC上方FBAR和并联LC谐振的新颖的VCO架构”描述了具备BAW谐振器的VCO。即使此电路具有优良的相位噪声和可接受的频率变化范围,其由于向BAW谐振器添加串联元件而引起的串联上的电阻性损失的增加而具有高消耗。
而且,此VCO的品质因数受到BAW谐振器的固有性质的限制,其相当大地限制VCO的频率变化范围。举例来说,对于在接收中使用60MHz到2.14GHZ的频带的UMTS标准,与具有高质量系数的谐振器一起操作的集成VCO均不能满足针对使用这些宽频带的数字移动通信系统的频率变化范围约束。用于这些应用的集成VCO因此当前在具有小于10的质量系数的集成谐振器的辅助下操作。
BAW或SAW类型的高质量系数谐振器还用于产生用于移动通信装置的多标准发射和/或接收架构中的滤波器。这些滤波器例如由一个或若干耦合的谐振器制成,其中此耦合可为串联和/或并联以获得梯式滤波器或格型滤波器。
然而,这些滤波器难以覆盖所需频率范围,尤其对于移动通信系统。
发明内容
因此需要提出一种电谐振器装置,其不具有现有技术的缺点,即其归功于高质量系数谐振器而具有良好质量系数,同时提供宽频率变化范围。
为实现此目的,一个实施例提出一种电谐振器装置,其能够在可变频率ω下操作或在可变频率ω下操作,所述电谐振器装置至少包括:
-声波谐振器,
-第一电路,其并联耦合到所述谐振器,且具有正的且可调节的电容,
-第二电路,其并联耦合到所述谐振器和所述第一电路,且具有严格为负的电容。
一个实施例还提出一种电谐振器装置,其能够在可变频率ω下操作或在可变频率ω下操作,所述电谐振器装置至少包括:
-声波谐振器,
ω为所述装置的谐振频率或操作频率。
ω对应于所述电装置的操作频率。举例来说,当电谐振器装置为压控振荡器时,操作频率的值取决于施加到振荡器的控制电压的值。
因此,归功于具有严格为负的电容和电容性行为的第二电路,无论装置的操作频率如何,装置的反谐振频率均移动到高于声波谐振器的自然反谐振频率的频率,而不改变其串联谐振频率。换句话说,可能增加声波谐振器的机电耦合,且因此增加装置的频率变化范围。因此装置在宽频率范围上起作用。
根据此实施例,可能产生具有串联谐振频率和反谐振频率的振荡布置,例如压控振荡器或滤波器,其包括声波谐振器、具有实部可为负且虚部等效于负电容的复阻抗的电子功能、以及正可变电容。此振荡布置准许例如产生压控振荡器,其相位噪声主要由声波谐振器的高质量决定,且其频率变化范围显著高于使用一谐振器例如RLC谐振器获得的变化范围。此振荡布置还准许产生滤波器,其过滤比已知滤波器宽的频带,而不会使滤波器的插入损失和抑制降级。
此外,此实施例准许获得具有低相位噪声的宽频率调谐范围的VCO,且得益于以并联耦合的声波谐振器的质量系数增加的可变电容的质量系数,其中调谐频率可对应于VCO的反谐振频率。
最终,通过在反谐振频率下操作,如此产生的VCO维持低消耗,同时维持低相位噪声。
电装置且尤其是第二电路可以由小尺寸组件制成,因此使得能够实现完全集成的例如VCO等电装置,其例如以微电子技术实现,即具有微米尺寸。
第二电路具有电容性行为,无论装置的操作频率如何。事实上,与具有正虚部且在操作频率增加期间在Smith列线图上绘制时根据顺时针方向转动的正电感的阻抗相反,负电容的阻抗具有正虚部,但在操作频率增加期间在Smith列线图上绘制时根据逆时针方向转动。
虽然对于给定操作频率,正电感值存在,使得阻抗的位置(在Smith列线图上)对应于负电容的阻抗的位置,获得此电感的限制比获得对应负电容重要得多。举例来说,对于例如等于100MHz左右的低操作频率,-1pF的电容具有对应于2.5μH的电感的阻抗,其无法以集成方式实现。
而且,负电容的阻抗的衍生物不同于电感的衍生物。
-在装置的低频率操作期间不使声波谐振器短路,且因此保持组件性质与直流循环相反;
-呈现良好品质因数。
第二电路可进一步具有严格为负的电阻器。
第二电路的复阻抗可包括值严格为负的实部。
第二电路可包括耦合到电感性组件的多个场效应晶体管。
第一电路可包括变容二极管类型的至少一个二极管或至少一个开关电容。
所述谐振器可为体声波或表面声波类型。
所述装置可进一步包括第三电路,所述第三电路并联耦合到所述谐振器、所述第一和第二电路,且具有负电阻器或实部具有严格负值的复阻抗。
所述第三电路可包括由至少两个场效应晶体管形成的至少一个差分对。
本文档还涉及一种包括例如上文所述的至少一个装置的压控振荡器(VCO)。
本文档还涉及一种具有例如先前所述的至少一个装置的电子滤波器。
附图说明
在参考附图阅读了仅以说明而非限制的方式提供的实施例的描述之后将更好地理解本发明,其中:
-图1表示根据一个特定实施例的压控振荡器;
-图2表示压控振荡器的第二电路的实施例;
-图3表示图2所示的第二电路的等效电路;
-图4表示由压控振荡器使用的声谐振器的首先模型化的等效电路;
下文描述的不同图的相同、相似或等效部分带有相同的数字参考,以便促进从一个图到另一图的传递。
图中所示的不同部分不一定按均一比例绘制,以便使图更容易阅读。
应了解,不同的可能性(变体和实施例)彼此不排斥且可组合。
具体实施方式
首先参看图1,其表示根据一个特定实施例的压控振荡器(VCO)100的一个实例。
VCO 100具备具有高质量系数(例如在约500与1500之间)的谐振器101。在关于图1描述的实施例中,谐振器101为体声波类型(BAW)。
VCO 100进一步包括第一电路,其具有正可变电容,即具有虚部等于的可调节复阻抗,其中C1≥0且ω为VCO 100的谐振频率,即,VCO 100的操作频率。此第一电路在此处由相对于彼此串联耦合的变抗器或变容二极管类型的一对二极管108、110形成。此第一电路与谐振器101并联耦合。定位于两个二极管108、110之间的命令输入112允许将命令电压施加到两个二极管108、110,其中电容的值(即,由两个二极管108、110呈现的复阻抗的虚部的值)是根据此命令电压的值界定。
VCO 100还包括第二电路114,其具有严格为负的电容,即具有虚部等于的复阻抗,其中C2<0。此第二电路114还与二极管108、110和谐振器101并联耦合。VCO100的电源电压VDD进一步施加到第二电路114。
最终,VCO 100包括并联耦合到谐振器101、第一电路108、110以及第二电路114的第三电路119,其向VCO 100的其它元件呈现负电阻器,即实部具有严格负值的复阻抗。在图1中的实例中,此第三电路119包括由差分安装的MOS类型的两个场效应晶体管102、104形成的差分对。第三电路119还包括电容器103,以及两个电流极化源105。电容器103确保差分对在低频下具有小于1的增益,因此避免其由于对其连续频率的正反应作用而表现为类似于开关,且因此避免差分对阻塞。
图2中展示第二电路114的一个实施例。此第二电路114包括彼此相同的两个晶体管MOS 113a,以及也彼此相同的两个其它晶体管MOS 113b。这四个晶体管由两个电流源115极化。第二电路114进一步包括具有值L的电感117。最终,输入118准许第二电路114并联耦合到VCO 100的其它元件。
图3中展示第二电路114的等效电路。此等效电路包括第一电阻性元件120,其电阻器等于晶体管113a的漏极-源极电阻器Rds1。此第一电阻性元件120并联耦合到第一电容性元件122,其电容等效于晶体管113b的栅极-源极电容Cgs2。第一电容性元件122并联耦合到彼此串联耦合的三个其它元件:
-第二电阻性元件124具有等于-1/(gm2Rds2)的负电阻器,其中gm为晶体管113a和113b的跨导,且Rds2是晶体管113b的漏极-源极电阻器,
-具有等于-Cgs1/gm2的值的电感性元件126,其中Cgs1在此处是晶体管113a的栅极-源极电容,
-具有等于-L.gm2的负电容的第二电容性元件128。
因此可见,第二电路114的复阻抗尤其由等于-1/(gm2Rds2)的负实部和等于的虚部形成,其中c2=-L.gm2,其中第一电阻性元件120和第一电容性元件122的阻抗相对于第二电阻性元件124和第二电容性元件128的阻抗可忽略。此外,假定电感性元件126和第二电容性元件128的阻抗的值取决于gm的值,则此gm的值因此经选择以使得可避免电感性元件126与第二电容性元件128之间的寄生谐振,同时在第二电路114上具有适于VCO 100的复阻抗。
声波谐振器101单独的频率响应可通过图4所示的等效电路以第一程度模型化。此电路包括串联耦合到电阻器Rm 134和电容Cm 136的电感Lm 132,其中这三个元件并联耦合到彼此串联耦合的两个元件:电阻器Ro 138和电容Co 140。这五个元件与表示谐振器101的电损失的两个电阻器Rs 142串联耦合。
电感Lm和电容Cm表示谐振器101的声效果本身。谐振器101的串联谐振频率ωr由以下等式表达:
电容Co表示谐振器101的介电作用,且根据以下表达式而介入谐振器101的反谐振频率ωa的计算:
谐振器101的总体阻抗Z在此情况下等效于:
其中:
且ω:谐振器101的频率
在此处忽略对应于电阻器Rs的损失而表达Φ。
电阻器Ro表示介电损失且Rm表示声损失。因此可能通过以下表达在串联谐振频率ωr下界定谐振器101的质量系数Qr:
Qr=Qm (5)
其中Qm:谐振器的等效电路的声频支所特定的质量系数,其取决于声损失Rm。
在反谐振频率ωa下谐振器101的质量系数Qa由以下表达界定:
Qo为模型的介电分支专有的质量系数,其取决于介电损失Ro。
图5中说明的曲线图200表示根据谐振器101的频率ω在没有VCO 100的其它元件的情况下声谐振器101的阻抗Z的演进。此曲线图200包括较低峰206,其对应于由上文提到的等式(1)表达的声谐振器101的串联谐振频率ωr。此外,曲线图200还包括较高峰208a,其对应于以上由等式(2)表达的声谐振器101的反谐振频率ωa。
图5中说明的曲线图202表示当谐振器101耦合到第二电路114时根据谐振器101的频率ω的声谐振器101的阻抗Z的演进,所述第二电路114具有虚部等于的复阻抗,其中C2<0。两个曲线图200和202包括同一较低峰206,指示串联谐振频率ωr在具有或不具有电路114的情况下保持不变。另一方面,可见曲线图202具有相对于峰208a朝向较高频率偏移的较高峰208b,从而反映反谐振频率通过将电路114与谐振器101耦合而已朝向较高频率移动的事实。此新的反谐振频率ωa′可在此情况下由以下等式表达:
其中C2:第二电路114的负电容的值或复阻抗的虚部(其中在图2所示实例的情况下C2=-L.gm2)。
在第一程度中,VCO 100的第二电路114的负电容并联到介电电容Co、谐振器101的电容Cm和电感Lm、以及由两个二极管108、110形成的正可变电容。
而假定在VCO 100中由二极管108、110形成的电容的变化将反谐振频率朝向较低频率带动,则第二电路114的负电容因此准许可能的反谐振频率的范围增加,其中此范围在第一配置与第二配置之间,在第一配置中二极管108、110的等效电容为零(对应于例如曲线图202),在第二配置中二极管108、110的等效电容使得反谐振频率达到大体上等于串联谐振频率的值。
VCO 100的反谐振频率的变化范围的此增加伴随着对由元件101、108、110和114形成的布置的反谐振质量系数的修改,所述系数随后采用以下值(在反谐振频率下):
其中C1:二极管108、110的等效电容;
Cn:第二电路114的负电容的绝对值,即Cn=|C2|;
Q//:声谐振器101的介电分支Qo和可变电容(二极管)Qv的经加权质量系数的和,使得:
Rv表示二极管108、110的电损失。
在等式(8)中等于的质量系数Q//的增加因数因负电容Cn的存在而减小。严格来说,质量系数Q//的公式表达应考虑第二电路114的电阻性损失。然而,这些电阻性损失为负的,且贡献于产生振荡条件。其因此不应被考虑,使得该分析保持与使用不具有负电容的谐振器的VCO的分析相当。
在先前关于图2描述的第二电路114的实施例中,自然地其具有负电阻器,即实部具有负值的复阻抗,其大体上在使用此功能时在可能的情况下最小化。然而,在对于本文描述的VCO的应用中,此负电阻器相反以高值来选择,以便满足振荡条件。如果第二电路114的负电阻器足够,即其准许补偿谐振器101的损失,那么可能制成不具有第三电路119的VCO 100。
举例来说,对于VCO 100,通过选择谐振器101以使得单独的谐振器的反谐振质量系数等于600,且其初始频率为2.306GHz,则添加具有虚部等效于-0.7pF的负电容的虚部的复阻抗的第二电路将反谐振频率带到2.43GHz。质量系数随后降低到约220。通过随后改变二极管108、110的复阻抗的虚部的值,所述值等效于从0到2.8pF的正电容,其固有质量系数为100,可见此覆盖大于160MHz的频率变化范围,其具有在其接近串联谐振频率时增加的质量系数。此160MHz的变化范围对应于当前数字移动通信系统所需的变化范围。
具有高质量系数的谐振器(即VCO 100的BAW谐振器101)特征在于串联谐振频率和反谐振频率,使得这两个频率之间的频率差取决于谐振器的物理特性。对于移动通信系统,高质量系数谐振器优选为体声波谐振器(BAW),其使用的压电材料可为氮化铝或适于制成此高质量系数谐振器的任何其它压电材料。在VCO 100的一个变体中,谐振器101可为表面声波谐振器(SAW)。
制成VCO可导致根据若干可用微电子技术(例如倒装芯片、结合或甚至后处理)的高质量系数谐振器的集成。
先前描述的第二电路114是以使用CMOS技术制成的晶体管制成。然而,这些晶体管也可使用SOI、BiCMOS或甚至AsGa技术制成。
此外,由二极管108、110产生的可变电容也可使用其它组件(例如,开关电容)制成。
朝向使用具有虚部等效于负电容的虚部的复阻抗的电路的声波谐振器的反谐振频率的较高值的位移是增加谐振器的机电耦合系数的电学等效物。布置的阻抗因此也增加。这些性质可用以制成对RF频率具有非常宽频带的选择性滤波器。事实上,压电谐振器滤波器的带宽直接取决于此耦合系数。谐振器与具有负电容的电路的耦合允许产生具有插入损失和抑制的滤波器,其几乎与典型滤波器相同,但其带宽可达到150MHz以上(与已知滤波器的约60MHz相比)。
VCO 100可例如通过以下方式来获得:首先在衬底上制成例如电路114、119以及二极管108和110等不同电子元件,随后在这些电子元件附近或这些电子元件上例如通过线结合或倒装芯片而制成谐振器101和连接器。
Claims (10)
2.根据权利要求1所述的装置,其特征在于,所述第二电路的所述复阻抗包括值严格为负的实部。
3.根据权利要求1所述的装置,其特征在于,所述第二电路包括耦合到电感性组件的多个场效应晶体管。
4.根据权利要求1所述的装置,其特征在于,所述第一电路包括变容二极管类型的至少一个二极管。
5.根据权利要求1所述的装置,其特征在于,所述第一电路包括至少一个开关电容。
6.根据权利要求1所述的装置,其特征在于,所述谐振器为体声波或表面声波类型。
7.根据权利要求1所述的装置,其特征在于,其进一步包括第三电路,所述第三电路并联耦合到所述谐振器、所述第一和所述第二电路,且具有实部具有严格负值的复阻抗。
8.根据权利要求7所述的装置,其特征在于,所述第三电路包括由至少两个场效应晶体管形成的至少一个差分对。
9.一种至少包括根据权利要求1所述的装置的压控振荡器。
10.一种至少包括根据权利要求1所述的装置的电子滤波器。
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- 2008-01-18 FR FR0850327A patent/FR2926689A1/fr not_active Withdrawn
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2009
- 2009-01-16 CN CN2009801023474A patent/CN101971484A/zh active Pending
- 2009-01-16 JP JP2010542639A patent/JP2011510551A/ja active Pending
- 2009-01-16 EP EP09701710A patent/EP2243215A1/en not_active Withdrawn
- 2009-01-16 US US12/812,559 patent/US20110018649A1/en not_active Abandoned
- 2009-01-16 WO PCT/EP2009/050495 patent/WO2009090244A1/en active Application Filing
Patent Citations (3)
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US20040130404A1 (en) * | 2001-06-07 | 2004-07-08 | Csem Centre Suisse D'electronique Et De | Differential oscillator circuit including an electro-mechanical resonator |
US7030718B1 (en) * | 2002-08-09 | 2006-04-18 | National Semiconductor Corporation | Apparatus and method for extending tuning range of electro-acoustic film resonators |
US20050174198A1 (en) * | 2003-12-29 | 2005-08-11 | Stmicroelectronics S.A. | Electronic circuit comprising a resonator to be integrated into a semiconductor product |
Cited By (7)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
CN107248852A (zh) * | 2015-06-03 | 2017-10-13 | 威盛日本株式会社 | 声波装置 |
CN108736906A (zh) * | 2017-04-13 | 2018-11-02 | 意法半导体有限公司 | 具有用于限制通过阻抗失配的损失的装置的传输线 |
CN108736906B (zh) * | 2017-04-13 | 2020-06-16 | 意法半导体有限公司 | 具有用于限制通过阻抗失配的损失的装置的传输线 |
CN112154304A (zh) * | 2018-05-23 | 2020-12-29 | Iee国际电子工程股份公司 | 电容式测量中补偿温度影响的方法 |
CN112154304B (zh) * | 2018-05-23 | 2024-01-12 | Iee国际电子工程股份公司 | 电容式测量中补偿温度影响的方法 |
CN110635817A (zh) * | 2018-06-05 | 2019-12-31 | 恒玄科技(上海)有限公司 | 一种增强发射信号的lc匹配电路 |
CN110635817B (zh) * | 2018-06-05 | 2021-08-06 | 恒玄科技(上海)股份有限公司 | 一种增强发射信号的lc匹配电路 |
Also Published As
Publication number | Publication date |
---|---|
WO2009090244A1 (en) | 2009-07-23 |
EP2243215A1 (en) | 2010-10-27 |
US20110018649A1 (en) | 2011-01-27 |
JP2011510551A (ja) | 2011-03-31 |
FR2926689A1 (fr) | 2009-07-24 |
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