WO2023174613A1 - Unité électronique pour appareil électrique - Google Patents

Unité électronique pour appareil électrique Download PDF

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Publication number
WO2023174613A1
WO2023174613A1 PCT/EP2023/052435 EP2023052435W WO2023174613A1 WO 2023174613 A1 WO2023174613 A1 WO 2023174613A1 EP 2023052435 W EP2023052435 W EP 2023052435W WO 2023174613 A1 WO2023174613 A1 WO 2023174613A1
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WIPO (PCT)
Prior art keywords
power transistor
diode
electronic unit
electrode
common source
Prior art date
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PCT/EP2023/052435
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German (de)
English (en)
Inventor
Philipp Zipf
Soenke SCHUCH
Original Assignee
Robert Bosch Gmbh
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Robert Bosch Gmbh filed Critical Robert Bosch Gmbh
Publication of WO2023174613A1 publication Critical patent/WO2023174613A1/fr

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/16—Modifications for eliminating interference voltages or currents
    • H03K17/161—Modifications for eliminating interference voltages or currents in field-effect transistor switches
    • H03K17/165—Modifications for eliminating interference voltages or currents in field-effect transistor switches by feedback from the output circuit to the control circuit
    • H03K17/166—Soft switching

Definitions

  • the invention relates to an electronic unit for controlling an electrical load of an electrical device according to the preamble of independent claim 1.
  • the invention relates to an electrical device with an electronic unit according to the invention.
  • MOSFETs Metal Oxide Semiconductor Field-Effect Transistor
  • the speed of the switching operations of the power transistor must be limited.
  • the electronic unit should still be able to switch sufficiently quickly to minimize any switching losses in the power transistors.
  • CMOS complementary metal-oxide-semiconductor
  • CMOS complementary metal-oxide-semiconductor
  • DRV8303 adjustable driver current and/or time-controlled current profiles
  • each phase of the inverter switching device has a pair of power transistors that convert a DC voltage into a pulse width modulated signal for driving the AC motor by clocking the power transistors.
  • both power transistors must never be switched on at the same time. This danger arises in particular from steep switching edges and from any disturbances, such as electrical noise or magnetic coupling between the electrical components of the inverter switching device and the signal paths of the control inputs of the power transistors.
  • correspondingly high clock rates are also required for high speeds of the AC motor.
  • a blocking circuit is provided in the inverter switching device, which can actively block the control input of the power transistor.
  • the operation of this active blocking circuit can be improved by specifically increasing a so-called common source inductance of the power transistor, since in this way the switching losses are reduced while voltage peaks remain the same.
  • the common source inductance results from the structural conditions of the power transistor and results from inductances that are both of a control current as well as a load or power current of the power transistor flows through it. An increase in the common source inductance is possible through measures in the layout or by using another component with appropriate casing (housing or package).
  • the common source inductance essentially consists of two parts. A first portion, which results from the semiconductor housing, and a second portion, which results from the structure on a circuit board on which the semiconductor housing is soldered.
  • An increase in the common source inductance leads, among other things, to advantages in that the insensitivity to voltage transients (dU/dt) increases, especially in half bridges ("Effect and Utilization of Common Source Inductance in Synchronous Rectification", International Rectifier, APEC 2005) and the Rate of change of the load or power current (dl/dT) in a component, especially in a MOSFET in hard-switching applications, can be reduced to safe/non-critical values (“Reverse Recovery Operation and Destruction of MOSFET Body Diode”, Toshiba, 09/01/2018) because the body diode is less loaded.
  • the reduced current change (dl/dT) in the component enables better EMC behavior to be achieved.
  • the invention relates to an electronic unit for controlling an electrical load of an electrical device, in particular a motor-driven electrical device, wherein the electrical load, in particular each phase of the electric motor, is assigned at least one power transistor, each with at least one control electrode, an outflow electrode and an inflow electrode, wherein a control circuit of the Power transistor a driver circuit controlling the control electrode and a power circuit of the power transistor comprising the outflow electrode and the inflow electrode, and wherein an inductance of the control circuit and an inductance of the power circuit form at least one common source inductance.
  • the switching behavior of the at least one power transistor can be influenced by means of the at least one common source inductance between the drain electrode and the driver circuit of the control circuit via a first diode and at least one second diode connected in anti-parallel to it.
  • This means that power voltage peaks during the switch-on or switch-off process can be adjusted more precisely and the EMC can be specifically improved, while in hard-switching applications the load on any (body) diodes can be reduced through a lower dl/dT.
  • the circuit can also be optimized in such a way that the switch-off voltage peaks of the at least one power transistor are limited to a permitted or desired level, while the so-called inrush current can be limited when capacitive loads are switched on.
  • the at least one power transistor of the electronic unit is preferably designed as a MOS ET.
  • power transistors in the form of other field effect transistors (FET), bipolar junction transistors (BJT), IGBTs (insulated gate bipolar transistors), wide bandgap semiconductors or the like are also possible.
  • FET field effect transistor
  • BJT bipolar junction transistors
  • IGBTs insulated gate bipolar transistors
  • the control electrode is designed as a gate
  • the inflow electrode as a drain and the outflow electrode as a source.
  • the control electrode, inflow electrode and outflow electrode are designed as base, collector or emitter and in the case of an IGBT as gate, collector or emitter.
  • Electrical devices in the context of the invention should be understood to mean all electrically operated devices with an electrical load, in particular with an electric motor drive, which can be powered by mains power or energy storage, such as batteries, removable battery packs or permanently integrated batteries, and in which the electronic unit according to the invention is used can.
  • electrically commutated electric motors (so-called EC or BLDC motors) come into consideration as electric motor drives, the individual phases of which are controlled via at least two power transistors of the electronic unit via pulse width modulation to control or regulate their speed and/or their torque.
  • the invention can be used on battery-operated and/or mains-operated machine tools for machining workpieces using an electrically driven insert tool.
  • the electrical processing device can be designed both as a hand-held machine tool and as a stationary machine tool.
  • Typical machine tools in this context are hand or drilling machines, screwdrivers, impact drills, planers, angle grinders, orbital grinders, polishing machines or the like.
  • Electrical devices also come with electric motors. powered garden and construction equipment such as lawn mowers, grass trimmers, pruning saws, motor and trenchers, blowers, robot breakers and excavators or the like.
  • the invention can be applied to electric motors of household appliances such as vacuum cleaners, mixers, etc.
  • the term electrical device can also be understood to mean road and rail vehicles powered by electric motors, as well as aircraft and ships.
  • the invention can also be used without restriction for predominantly capacitive electrical loads 12 or corresponding mixed forms.
  • the electronic unit according to the invention can also be used in switching regulators or inverters, such as those in chargers or power supplies, step-up, step-down, flyback or forward converters, buck-boost converters and Sepie, Zeta, H5 or HERIC® topologies can be used.
  • a series circuit consisting of the at least one second diode and the at least one common source inductor is connected in anti-parallel to the first diode.
  • the at least one second diode enables an adaptation to the corresponding common source inductance in such a way that only the switch-off process of the power transistor is specifically influenced without affecting the switch-on process.
  • the common source inductance of the power circuit inserted into the control circuit during the switch-off process thus behaves like a resistance that is dependent on a change in a load current over time.
  • the first diode and the at least one second diode are connected between the drain electrode and the driver circuit of the control circuit in such a way that no additional voltage drops across the at least one common source inductance when the power transistor is switched on and that when the power transistor is switched off an induction voltage drops across the at least one common source inductor, which counteracts a drive voltage of the driver circuit.
  • the induction voltage generated by a change in the control current over time leads therefore in a particularly advantageous manner to delay the switch-off process.
  • the first diode and the at least one second diode are connected between the drain electrode and the driver circuit of the control circuit in such a way that no additional voltage drops across the at least one common source inductance when the power transistor is switched off and that when the power transistor is switched on Temporal change in a load current flowing through the electrical load causes the control electrode to open more slowly.
  • This concept is particularly applicable to power transistors designed as P-channel MOSFETs or PNP bipolar transistors.
  • Schottky diodes are preferably used as the first and at least one second diode.
  • the power transistor, the first diode and the at least one second diode are arranged locally adjacent on the same substrate of a printed circuit board of the electronic unit, their PN junctions have essentially the same negative temperature coefficient with respect to the diode forward voltages and the threshold voltage of the control electrode, resulting in a thermally stable system.
  • a third diode in particular a series circuit consisting of a third diode and a resistor, is connected in parallel to a series resistor of the control electrode of the power transistor in such a way that the series resistor is bridged when the power transistor is switched on or switched off.
  • the switching time of the power transistor in the current change phase of the power current during the switching process is largely determined by the common source inductance, provided the internal resistance of the driver circuit is sufficiently small.
  • the switching times of the power transistor until the threshold voltage and the final voltage are reached can then be carried out as quickly as possible in order to achieve this To minimize dead time while simultaneously setting the switching time via the layout for optimized E MV behavior.
  • control electrode and the inflow electrode of the power transistor can also be bridged by a capacitor, in particular by a series connection of a capacitor and a resistor, in order to extend the residence time in the Miller plateau.
  • the invention also relates to an electrical device, in particular a motor-driven electrical device, with an electronic unit according to the invention for controlling an electrical load, in particular a single- or multi-phase electric motor.
  • the electric motor can be designed as a single-phase DC motor and the electronic unit for controlling the DC motor can be designed as a half bridge with a low-side and a high-side power transistor, with the switching behavior being influenced either for only one of the two power transistors or for both power transistors via which at least one common source inductance occurs.
  • the electric motor of the electrical device can be designed as a three-phase EC motor, with the electronic unit for controlling the EC motor having a B6 bridge with one low-side and one high-side power transistor per bridge branch.
  • the switching behavior can be influenced via the at least one common source inductance either for only one of the two power transistors or for both power transistors of a bridge branch.
  • Fig. 2 Time diagrams of the switching times of a power transistor designed as a MOSFET (Fig. 2a) as well as the associated signal curves of the voltages and currents when switching off (Fig. 2b) and switching on (Fig. 2c) of the MOSFET according to the prior art,
  • FIG. 3 a circuit diagram of a first exemplary embodiment of the electronic unit according to the invention for inductively controlled switching off and ohmic switching on of a power transistor designed as a MOSFET,
  • Fig. 4 Time diagrams of the switching times of the MOSFET (Fig. 4a) of the electronic unit according to the invention according to Figure 3 as well as the signal curves of the associated voltages and currents when switching off (Fig. 4b) and when switching on (Fig. 4c) of the MOSFET,
  • Fig. 6 a circuit diagram of a second exemplary embodiment of the electronic unit according to the invention for inductively controlled switching off and for accelerated ohmic switching on Power transistor designed as a MOSFET,
  • FIG. 7 a circuit diagram of a third exemplary embodiment of the electronic unit according to the invention for accelerated, inductively controlled switching on and ohmic switching off of a power transistor designed as a MOSFET,
  • Fig. 11 a circuit diagram of a sixth exemplary embodiment of the electronic unit according to the invention for ohmic switching on and for inductively controlled switching off of a power transistor designed as a MOSFET with slower switching times and
  • Fig. 12 a circuit diagram of a seventh exemplary embodiment of the electronic unit according to the invention for inductively controlled switching off and for accelerated ohmic switching on of a power transistor designed as a MOSFET with maximum configuration.
  • Description of the exemplary embodiments 1 shows a circuit diagram of an electronic unit 10 for an electrical device not shown in detail according to the prior art.
  • any electrically operated device with an electrical load 12 that can be powered by mains power or energy storage such as batteries, removable battery packs or permanently integrated batteries, can serve as an electrical device in the context of the invention.
  • electrical devices that can be used to process workpieces using an electric motor-driven application tool such as hand or stand drills, screwdrivers, impact drills, hammer drills, planers, angle grinders, orbital grinders, polishing machines, circular saws, table saws, chop saws and jigsaws or the like, can be used as electrical devices come into use.
  • an application of the electronic unit 10 is also conceivable in household appliances such as vacuum cleaners, mixers, kitchen machines, hobs or the like, garden equipment such as lawn mowers, shredders, pruning saws, etc., construction machines such as concrete mixers or electric motor-driven vehicles and aircraft, etc .
  • the electrical load 12 can be designed, for example, as a single- or multi-phase electric motor 14.
  • a design as an electrically commutated (EC) or brushless (BLDC) direct current motor 16 is conceivable, whose three phases U, V, W each have at least two power transistors 18 of a power stage 22 of the electronic unit designed as a B6 bridge 20 10 are assigned.
  • the windings of the EC motor 16 associated with the phases U, V, W are marked with 24.
  • the windings 24 of a phase U, V, W can be distributed over several stator teeth, not shown, of a stator of the EC motor 16, with the stator teeth of a phase U, V, W each forming a stator pole.
  • the invention can also be used without restriction for predominantly capacitive electrical loads 12 or corresponding mixed forms.
  • the power stage 22 and the driver circuit 26 of the electronic unit 10 are connected via a first reference potential Vi, in particular a supply potential V+, and a second reference potential V2, in particular a ground potential GND, supplied with energy.
  • a first reference potential Vi in particular a supply potential V+
  • a second reference potential V2 in particular a ground potential GND
  • a shunt resistor 30 is used to measure the load current I that flows through the windings 24 of the EC motor 16.
  • the driver circuit 26 can be regulated by a control circuit (not shown) of the electrical device.
  • the windings 24 of the EC motor 16, which are connected in a delta connection, are switched for each phase U, V, W by means of a high-side power transistor 30 and a low-side power transistor 32.
  • the power transistors 18 each have control electrodes 34 for generating a pulse width modulated energy signal via their outflow electrodes 36 and inflow electrodes 38 by means of the driver circuit 26.
  • the terms outflow and inflow electrodes should refer to the technical current flow direction and not to the physical current flow direction of the electrons.
  • the PWM control of the windings 24 of the stator poles of the EC motor 16 takes place in a known manner via corresponding power contact points 39 between the high-side and low-side power transistors 30, 32 of each bridge branch of the B6 bridge 20 in such a way that the high-side and the Low-side power transistors 30, 32 of a bridge branch are switched on and off alternately with one another, the transition from one to the next phase U, V, W (commutation) having a phase offset of 120 ° el., so that the current supply to the windings 24 a corresponding rotational movement of a rotor, not shown, of the EC motor 16. Since the PWM control of an EC motor 16 by means of a B6 bridge 20 is well known to those skilled in the art, this will not be discussed further.
  • the electronic unit 10 can have a power stage 22 designed as a half bridge or a single power transformer.
  • the person skilled in the art is aware of the different circuit topologies and types of electric motors 14, so it does not appear necessary to go into this in further detail.
  • each power transistor 18 of the power stage 22 of the electronic unit 10 can be designed as a MOSFET 40.
  • the control electrode 34 should be referred to as gate G, the outflow electrode 36 as source S and the inflow electrode 38 as drain D.
  • power transistors in the form of bipolar junction transistors (BJT), IGBTs (insulated gate bipolar transistors) or the like are also possible.
  • BJT bipolar junction transistors
  • IGBT insulated gate bipolar transistors
  • control electrode 34, outflow electrode 36 and inflow electrode 38 are designed as base B, emitter E or collector C and in the case of an IGBT as gate G, emitter E or collector C.
  • the various variants of power transistors 18 are known to those skilled in the art, so this will not be discussed in further detail.
  • Figure 2 shows various time diagrams of the switching times of a MOSFET 40 (Figure 2a) as well as the associated signal curves of the measured voltages and currents when switching off ( Figure 2b) and switching on ( Figure 2c) of the MOSFET 40 according to the prior art.
  • the switch-on phase ON and the switch-off phase OFF of the MOSFET 40 are each composed of four time periods ti to t4 and ts to ts.
  • ti defines the period of time in which, after the control voltage U GS has been applied via the gate-source transition of the MOSFET 40, there is initially no change in the drain current l D or the blocking voltage U DS on or in the MOSFET 40.
  • a further increase in the control voltage UGS beyond the threshold voltage UTK causes a current change in the drain current ID until the control voltage UGS has reached the plateau voltage U PI of the so-called Miller plateau at the beginning of the third time period ta.
  • the blocking voltage U DS drops AT the beginning of the second time period t2.
  • the fourth time period t4 finally defines the duration after the MOSFET 40 is switched on until the final value of the control voltage U GS is reached.
  • the internal resistance of the MOSFET 40 also decreases slightly.
  • the MOSFET 40 behaves exactly the other way around.
  • FIG. 2b shows the curves of the control voltages U G s, Hi g h and UGS.LOW of a MOSFET 40 designed as a high-side power transistor 30 and a low-side power transistor 32 of a branch of the power stage 22, the blocking voltage U DS, high across the drain -Source transition of the high-side power transistor 30 and the load current I flowing through the electrical load 12 and the corresponding switched-on power transistors 18 when the high-side power transistor 30 is switched off are plotted over time t.
  • Figure 2c shows the corresponding time curves when the high-side power transistor 30 is switched ON. Two adjacent graduation lines correspond to a time division of 50 ns.
  • the distribution is 10 A per division, for the control voltages U GS ,Hi g h, UGS.LOW it is 1 V and for the blocking voltage U D s,Hi g h it is 5 V per division.
  • the values are only to be understood as examples and can vary depending on the power transistor 18 used and the electrical load 12 of the electrical device. The switching behavior can also depend on the temperature or other external influences (e.g. EMC).
  • the present invention offers the advantage over the prior art of an optimized relationship between the switching speed and the power loss of a power transistor 18, whereby the power voltage peaks during the switch-on or switch-off process can be minimized and the EMC can be improved if the common source inductance is used for braking is used without the ratio between the on and off time of the power transistor 18 being significantly influenced, since only the dl / dt phase is extended.
  • FIG. 3 shows a first exemplary embodiment of the invention based on a section of a branch of the power stage 22 of the electronic unit 10, designed as a B6 bridge 20, for the high-side power transistor 30, designed as a MOSFET 40.
  • the electrical load 12 has not been shown for clarity; In this regard, reference is made to Figure 1.
  • the control electrode 34 of the MOSFET 40 designed as a gate G, is controlled in a clocked manner via a series resistor 42 with the resistance value Ri in order to pulse-width modulate the MOSFET 40. and turn off.
  • the MOSFET 40 When switching on, the MOSFET 40 is charged by means of a gate current l G via a control circuit 44, which, in addition to the driver circuit 26 and the series resistor 42, has the gate G and the drain electrode 36 of the MOSFET 40, which is designed as a source S, and one of another resistor 46 with the resistance value R2 and a first diode Di marked 48 in series, to which a second diode D2 marked 50 is connected in anti-parallel. Due to their low forward voltage are the first and the second diode 48, 50 is preferably designed as a Schottky diode.
  • the control circuit 44 controls a power circuit 52, which, in addition to the electrical load 12 (not shown), includes the source electrode S and the inflow electrode 38 of the MOSFET 40, which is designed as a drain D. If the MOSFET 40 is switched on, the load current I can flow through its drain-source path and the electrical load 12.
  • the control circuit 44 and the power circuit 52 each have an inductance, which together form a so-called common source inductance Les.
  • the common source inductance Les forms a series circuit with the second diode 50, which is connected in anti-parallel to the series circuit consisting of the first diode 48 and the further resistor 46.
  • a further, so-called layout source inductance LLS results from the construction of the MOSFET 40 on a circuit board 54 (see Figure 8) of the electronic unit 10. It is a geometric quantity and is determined via the geometry of the layout. Usual values of the layout source inductance LLS are between 0.5 and 30 nH. Apart from the layout source inductance LLS, every component - except those with a so-called Kelvin connection - has an intrinsic common source inductance Les in the range of 0.5 to 20 nH due to its geometric structure.
  • the charging process of the MOSFET 40 can take place very quickly depending on the resistance value R2 of the further resistor 46, since the increasing gate current l G does not cause a voltage drop across the common source inductance L C s during the ON process, so that for the switching voltage U G s of MOSFET 40 applies:
  • Vnigh is the control potential for the high-side power transistor 30;
  • V 3 describes a reference potential of the driver circuit 26, which may also be connected to the second reference potential V 2 or the ground potential GND can be identical.
  • VL OW and V3 define the potentials for controlling the low-side power transistor 32, not shown, of the corresponding bridge branch of the B6 bridge 20.
  • the discharging process of the MOSFET 40 takes place via the driver circuit 26, the series resistor 42 and the second diode 50 of the control circuit 44. Since the second diode 50 is connected after the common source inductance Les, the increasing gate current l G generates an induction voltage Ucs, which the driver voltage U Dr of the driver circuit 26 counteracts. In this way, the OFF switch-off process can be extended inductively without significantly influencing the ON switch-on process. This applies to the control voltage of the MOSFET
  • UGS Uor ⁇ U RI - UD2 - Ucs.
  • the switch-off delay of the MOSFET 40 is therefore based on a change in the drain current l D or the load current I, which in turn causes feedback via the common source inductance L C s as a result of the second diode 50.
  • FIG. 4 shows, corresponding to FIG .
  • Figure 4a makes it clear that the common source circuit according to the invention determines the time period t? within which the control voltage UGS changes from the plateau voltage UPI of the Miller switch when the MOSFET 40 is switched OFF. Plateaus drop to the threshold voltage Uih and the drain current ID has reached its final value of 0 A, is significantly longer than in Figure 2, while all other time periods have remained almost unchanged. Accordingly, a slower turn-off behavior of the voltages and currents can be seen in FIG. 4b compared to FIG. 2b, while there are no significant differences in the voltage and current curves in FIGS. 4c and 2c during the ON process.
  • ti and ts denote the time periods in ns in which the control voltage U GS , Hi g h reloads the input capacitances of the MOSFET 40 when the high-side power transistor 30 is switched ON or when the high-side power transistor 30 is switched off without it changing its capacity
  • t2 and t? are the time periods in ns in which the drain current ID or power current I flowing through the MOSFET 40 changes to the respective final value and in which the invention, depending on the polarity of the two diodes Di and D2, during one of the two switching processes (ON or OFF) becomes effective.
  • the MOSFET 40 only switches the blocking voltage UDS due to the Miller effect.
  • the two remaining time periods t4 and ts were not included in the table for the sake of clarity, as they have no particular significance for the functioning of the invention.
  • the bottom two lines of the table show, on the one hand, the measured values with the diode circuit according to the invention according to Figure 3 (“with Di, D2”) and without diode circuit according to the prior art according to Figure 1 (“w/o diodes”).
  • the switch-on process ON is only minimally delayed compared to the circuit without diodes according to FIG. 1 for all time periods ti to ts (t4 correspondingly). It is also noticeable that, regardless of the type of switching process (ON, OFF), the deviation between the two circuit variants while the drain current ID changes within the time period t2 is very small. Under the influence of the second diode 54, however, the MOSFET 40 is switched OFF during the time period t? significantly extended because a change in the Drain current ID causes a negative feedback due to the common source voltage Ucs falling across the common source inductance Les.
  • FIG. 6 shows a second exemplary embodiment of the electronics unit 10 according to the invention for the high-side power transistor 30 designed as a MOSFET 40.
  • a series circuit consisting of a third diode D 3 marked with 56 and a further resistor 58 with the resistance value R 3 is now connected in parallel to the gate series resistor 42 in such a way that the third diode 56 passes through the series resistor 42 bridges the further resistor 58 when switching off OFF.
  • the switch-off time is determined almost exclusively by the common source inductance Les (time period t?) and the Miller effect (time period t 3 ), while the remaining time periods ts and ts depend on the resistance value R 3 of the further resistance 58 can be passed through more or less accelerated.
  • Figure 7 shows a third exemplary embodiment of the electronics unit 10 according to the invention, the polarities of the two diodes Di and D2 marked 48 and 50 being swapped compared to the first two exemplary embodiments.
  • the MOSFET 40 is thus switched off via the first diode 48 and the series resistor 42 in such a way that no additional common source voltage Ucs drops across the common source inductance Les during the switch-off process OFF.
  • the switching on of the MOSFET 40 is controlled inductively via the second diode 50 and the common source inductance Les in such a way that a temporal change in the drain current dlo/dt or the load current dl/dt during the switching ON process results in a slow Mere opening of the gate G results in a reduction in the rate of current change without significantly affecting the OFF switch-off process.
  • This circuit variant is particularly useful if the capacitive loads are to be switched in order to limit the inrush current in the capacity.
  • FIG. 8 shows a schematic representation of a section of the circuit board 54 of the electronics unit 10 according to the invention according to FIG.
  • the detail shows the high-side power transistor 30, designed as a MOSFET 40, of one of the three bridge branches of the B6 bridge 20.
  • the gate G of the MOSFET 40 is controlled via two plug contacts 56, 58 by the driver circuit 26, not shown (see also FIG. 7).
  • the corresponding phase U, V, W of the EC motor 16, not shown, is electrically connected to the source connection S of the MOSFET 40 via the power contact point 39.
  • the MOSFET 40, the first diode 48 and the second diode 50 are arranged locally adjacent on the same substrate of the circuit board 54 such that their PN junctions have essentially the same negative temperature coefficient with respect to the diode forward voltages and the threshold voltage UTK of the control electrode 34. This leads to a particularly advantageous thermally stable system.
  • the fourth exemplary embodiment according to FIG. 9 causes, analogously to the exemplary embodiments according to FIGS. 3 and 6, an inductively controlled switch-off process OFF of the MOSFET 40 via the second diode D 2 , the series resistor 48 and the common source inductance Les.
  • the third diode D 3 marked with 56 is now poled in such a way that it bridges the series resistor 48 during the switch-on process ON, so that it is accelerated to a greater or lesser extent depending on the resistance value R 3 of the third resistor 58.
  • a capacitor 60 connected in parallel to the gate-source junction of the MOSFET 40 the switching times during the switch-on process ON and the switch-off process OFF can be further slowed down depending on a capacitance value Ci of the capacitor 60.
  • a similar mode of operation as in the exemplary embodiment according to FIG. 10 can be achieved in the sixth exemplary embodiment according to FIG. 11 with an additional capacitor 64 parallel to the drain-gate path of the MOSFET 40.
  • the time periods ta and te that the MOSFET 40 remains in the Miller plateau can be extended in order to switch ON and to slow down the switch-off process OFF in the dll/dt phase (see Figures 2a and 4a).
  • FIG. 12 shows a seventh exemplary embodiment of the electronic unit 10 according to the invention with maximum configuration, in which, analogous to the exemplary embodiment according to FIG.
  • the inductively controlled switching off of the MOSFET 40 is now controlled via a multiple starting point in the circuit layout in such a way that different common source inductances are provided via several second diodes D 2 .I, D 2.2 , D 2 .3 marked 50a, 50b and 50c Lcs.i, Les, 2, Les, 3 can be selected.
  • electrical switches can alternatively be used for specifically switching the additional common source inductors Les, 2, Les, 3 on and off.
  • the switching behavior can be supplemented via the further resistor 46 connected in series with the first diode 48 (ON when switched on) and via additional resistors 66a, 66b, 66c connected in series with the second diodes 50a, 50b, 50c with the respective resistance values Rs. i, Rs,2, R ⁇ ,3 (when switching off OFF). It is also alternatively or additionally possible to have a common further resistor 68 with the resistance value Re in the source path of the control circuit 44 of the MOSFET 40 to be provided in such a way that it influences both the ON and OFF switching processes. Accordingly, to slow down the ON and OFF switching processes of the MOSFET 40, the capacitors 62, 64 and the resistor 64 shown in the exemplary embodiments according to FIGS. 10 and 11 can also be provided.
  • circuit variants of the electronics unit 10 can be used to influence the switching behavior of the power transistors 16 of the electronics unit 10 by means of the common source inductance Les.
  • Various combinations of correspondingly polarized first and second diodes 48, 50 and possibly third diodes 56 are therefore conceivable.
  • the effects that can be achieved depend on the change in the load current dl/dt over time. The larger this is, the greater the effect it has on the switching behavior of the power transistors 16.
  • the invention can also be applied in an analogous manner to a low-side power transistor 32 or to a single power transistor 18 for controlling the electrical load 12.

Abstract

L'invention concerne une unité électronique (10) destinée à commander une charge électrique d'un appareil électrique, en particulier d'un appareil électrique motorisé. La charge électrique est appariée à au moins un transistor de puissance (18, 30, 40) ayant au moins une électrode de commande respective (34, G), une électrode de sortie (36, S) et une électrode d'entrée (38, D), un circuit de commande (44) du transistor de puissance (18, 30, 40) comprenant un circuit d'attaque (26) qui active l'électrode de commande (34, G), et un circuit de puissance (52) du transistor de puissance (18) comprenant l'électrode de sortie (36, S) et l'électrode d'entrée (38, D). Une inductance du circuit de commande (44) et une inductance du circuit de puissance (52) forment au moins une inductance de source commune (Lcs). Le comportement de commutation du ou des transistors de puissance (18, 30, 40) peut être influencé par la ou les inductances de source commune (Lcs) entre l'électrode de sortie (36, S) et le circuit d'attaque (26) du circuit de commande (44) par l'intermédiaire d'une première diode (48, D1) et d'au moins une seconde diode (50, D2) qui est montée en antiparallèle par rapport à la première diode (48, D1) dans une branche constituée de la seconde diode et de l'inductance de source commune. L'invention concerne en outre un appareil électrique, en particulier un appareil électrique motorisé, comprenant une unité électronique (10) selon l'invention destinée à commander une charge électrique, en particulier un moteur électrique monophasé ou polyphasé.
PCT/EP2023/052435 2022-03-18 2023-02-01 Unité électronique pour appareil électrique WO2023174613A1 (fr)

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DE102022202702.7A DE102022202702A1 (de) 2022-03-18 2022-03-18 Elektronikeinheit für ein Elektrogerät

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