WO2021240657A1 - Dispositif de conversion de puissance, dispositif d'entraînement de moteur, soufflante, compresseur et climatiseur - Google Patents

Dispositif de conversion de puissance, dispositif d'entraînement de moteur, soufflante, compresseur et climatiseur Download PDF

Info

Publication number
WO2021240657A1
WO2021240657A1 PCT/JP2020/020782 JP2020020782W WO2021240657A1 WO 2021240657 A1 WO2021240657 A1 WO 2021240657A1 JP 2020020782 W JP2020020782 W JP 2020020782W WO 2021240657 A1 WO2021240657 A1 WO 2021240657A1
Authority
WO
WIPO (PCT)
Prior art keywords
voltage
capacitor
switching
power conversion
duty ratio
Prior art date
Application number
PCT/JP2020/020782
Other languages
English (en)
Japanese (ja)
Inventor
浩一 有澤
基 豊田
貴昭 ▲高▼原
修 森
啓介 植村
貴彦 小林
Original Assignee
三菱電機株式会社
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 三菱電機株式会社 filed Critical 三菱電機株式会社
Priority to JP2022527330A priority Critical patent/JP7297157B2/ja
Priority to PCT/JP2020/020782 priority patent/WO2021240657A1/fr
Publication of WO2021240657A1 publication Critical patent/WO2021240657A1/fr
Priority to JP2023018913A priority patent/JP7399331B2/ja

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode

Definitions

  • the present invention includes a power converter that converts an AC voltage output from an AC power supply into a DC voltage, a motor drive device equipped with a power converter, a blower and a compressor provided with a motor drive device, and a blower or a compressor. Regarding the air conditioner equipped.
  • Patent Document 1 discloses a multi-level power conversion device.
  • the power conversion device described in Patent Document 1 includes a sub-converter composed of four switching elements and one capacitor between the reactor and the main converter.
  • a high power factor is provided while the capacitor voltage of the sub-converter is interposed between the output DC voltage and the input AC voltage. Power control is performed.
  • the applied voltage of the reactor is reduced as compared with the high power factor converter having a general bridgeless configuration, and the capacity and loss of the reactor are reduced as compared with the high power factor converter of the same specifications.
  • the present invention has been made in view of the above, and an object of the present invention is to reduce the switching loss of a switching element and the high frequency loss of a reactor to obtain a highly efficient power conversion device.
  • the power conversion device includes at least one reactor and a plurality of switching elements. Further, the power conversion device includes a first capacitor, a second capacitor provided between the reactor and the first capacitor, and a controller for controlling the conduction of the switching element.
  • the reactor, switching element, first capacitor and second capacitor are provided between the AC power supply and the DC load.
  • the power conversion device performs power conversion between the AC voltage of the AC power supply and the voltage of the first capacitor, which is the voltage of the first capacitor.
  • the controller controls the switching of the switching element by switching the switching element once or more and 20 times or less in a half cycle of the AC voltage.
  • the present invention by reducing the switching loss of the switching element and the high frequency loss of the reactor, it is possible to drive the power conversion device with high efficiency.
  • the block diagram used for explaining the operation of the addition / subtraction determination device shown in FIG. A block diagram showing an internal configuration of the controller according to the second embodiment.
  • FIG. 19 A block diagram showing a detailed configuration of the carrier wave generator shown in FIG. A block diagram showing a detailed configuration of the high power factor controller shown in FIG. A block diagram showing a detailed configuration of the gate signal generator shown in FIG.
  • FIG. 1 is a diagram showing a basic circuit configuration of the power conversion device 100 according to the first embodiment.
  • the power conversion device 100 according to the first embodiment includes a main circuit 110 and a controller 8.
  • the main circuit 110 is a power conversion circuit that converts AC power generated by AC voltage output from AC power source 1 into DC power generated by DC voltage and applies it to the load 7.
  • the load 7 is a DC load.
  • a DC load is a load that operates by being supplied with DC power.
  • a load 7 including an inverter that converts DC power into AC power is also included in the DC load referred to here.
  • the main circuit 110 includes a reactor 2 for limiting current, a subconverter 3, a main converter 5, and a smoothing capacitor 6, which is a first capacitor.
  • the smoothing capacitor 6 is connected in parallel to each of the main converter 5 and the load 7 between the main converter 5 and the load 7.
  • the subconverter 3 includes a first leg in which the switching element 3a and the switching element 3b are connected in series, a second leg in which the switching element 3c and the switching element 3d are connected in series, and a direct current which is a second capacitor. It has a capacitor 4. The first leg, the second leg and the DC capacitor 4 are connected in parallel with each other.
  • Examples of the DC capacitor 4 and the smoothing capacitor 6 are an aluminum electrolytic capacitor and a film capacitor.
  • the main converter 5 includes a third leg in which the diode 5a and the switching element 5b are connected in series, and a fourth leg in which the diode 5c and the switching element 5d are connected in series. Both the third leg and the fourth leg have a configuration in which the anode of each diode is connected to the switching element. The third and fourth legs are connected in parallel with each other.
  • the diodes 5a and 5c may be replaced with switching elements.
  • One end of the reactor 2 is connected to one of the AC power supplies 1, and the other end of the reactor 2 is connected to the midpoint of the first leg.
  • the midpoint of the first leg is the connection point between the switching element 3a and the switching element 3b. Also in other legs, the connection point between the switching elements and the connection point between the diode and the switching element are called "midpoints".
  • the midpoint of the third leg is connected to the midpoint of the second leg of the subconverter 3, and the midpoint of the fourth leg is connected to the other of the AC power supply 1.
  • switching elements 3a, 3b, 3c, 3d (hereinafter, appropriately referred to as “3a to 3d”) and the switching elements 5b, 5d is a metal oxide semiconductor field effect transistor (Metal) in which diodes are connected in antiparallel.
  • Metal Oxide Semiconductor (Field Effect Transistor: MOSFET).
  • the anti-parallel connection means that the drain of the MOSFET and the cathode of the diode are connected, and the source of the MOSFET and the anode of the diode are connected.
  • a parasitic diode contained in the MOSFET itself may be used. Parasitic diodes are also called body diodes.
  • an insulated gate bipolar transistor (IGBT) or a high electron mobility transistor (HEMT) may be used.
  • IGBT insulated gate bipolar transistor
  • HEMT high electron mobility transistor
  • a cascode type GaN (Gallium Nitride) -HEMT is suitable.
  • the smoothing capacitor 6 smoothes and holds the DC voltage converted by the main converter 5.
  • the polarity of the smoothing capacitor voltage Vdc which is the voltage held in the smoothing capacitor 6, is indicated by an arrow.
  • the tip of the arrow is the high potential side, and the opposite side is the low potential side. This definition is the same for the subconverter 3. Further, in the AC power supply 1, the case where the tip of the arrow has a high potential is defined as the positive electrode property, and the case where the opposite side has a high potential is defined as the negative electrode property.
  • the power conversion device 100 further includes voltage detectors 30, 31, 33, and a current detector 32.
  • the voltage detector 30 detects the smoothing capacitor voltage Vdc.
  • the smoothing capacitor voltage may be referred to as “first capacitor voltage”, and the voltage detector 30 may be referred to as “first voltage detector”.
  • the detected value of the smoothing capacitor voltage Vdc detected by the voltage detector 30 is input to the controller 8.
  • the voltage detector 31 detects the DC capacitor voltage Vsub, which is the voltage of the DC capacitor 4.
  • the DC capacitor voltage may be referred to as a "second capacitor voltage”
  • the voltage detector 31 may be referred to as a "second voltage detector”.
  • the detected value of the DC capacitor voltage Vsub detected by the voltage detector 31 is input to the controller 8.
  • the current detector 32 detects the alternating current iac flowing in the reactor 2.
  • the detected value of the alternating current iac detected by the current detector 32 is input to the controller 8.
  • the voltage detector 33 detects the AC voltage vac output by the AC power supply 1.
  • the voltage detector 33 may be referred to as a "third voltage detector”.
  • the detected value of the AC voltage vac detected by the voltage detector 33 is input to the controller 8.
  • the controller 8 has gate signals G3a, G3b, G3c, for controlling conduction of the switching elements 3a to 3d based on the detected values of the smoothing capacitor voltage Vdc, the DC capacitor voltage Vsub, the AC voltage vac, and the AC current iac.
  • G3d (hereinafter, appropriately referred to as “G3a to G3d”) and gate signals G5b and G5d for controlling the conduction of the switching elements 5b and 5d are generated.
  • Each of the sub-converter 3 and the main converter 5 has a gate drive circuit (not shown).
  • Each gate drive circuit of the subconverter 3 generates a drive pulse using the gate signals G3a to G3d output from the controller 8, and applies the generated drive pulse to the gate of the corresponding switching element to control the switching element.
  • Drive Each gate drive circuit of the main converter 5 generates a drive pulse using the gate signals G5b and G5d output from the controller 8, and applies the generated drive pulse to the gate of the corresponding switching element to control the switching element.
  • Drive
  • controller 8 The internal configuration of the controller 8 and the detailed operation of the controller 8 will be described later.
  • the processor 8a is an arithmetic unit such as an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor).
  • arithmetic unit such as an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor).
  • the memory 8b is a non-volatile or volatile semiconductor such as RAM (Random Access Memory), ROM (Read Only Memory), flash memory, EPROM (Erasable Project ROM), and EEPROM (registered trademark) (Electrically EPROM).
  • RAM Random Access Memory
  • ROM Read Only Memory
  • flash memory EPROM (Erasable Project ROM)
  • EEPROM registered trademark
  • the memory 8b stores a function of the controller 8 described later and a program for executing the function of the controller 8 described later.
  • the processor 8a exchanges necessary information via an interface including an analog-to-digital converter and a digital-to-digital converter (not shown), and the processor 8a executes a program stored in the memory 8b to perform necessary processing.
  • the calculation result by the processor 8a can be stored in the memory 8b.
  • the function of the controller 8 may be realized by using a processing circuit.
  • the processing circuit referred to here corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof. Even in a configuration using a processing circuit, some processing in the controller 8 may be performed by the processor 8a.
  • the power conversion device 100 configured as described above performs power conversion between the AC voltage output by the AC power supply 1 and the first capacitor voltage which is the DC voltage held in the smoothing capacitor 6.
  • the function of this power conversion is carried out by the reactor 2, the sub-converter 3, and the main converter 5. Further, the power conversion device 100 controls the conduction of the switching elements 3a to 3d, 5b, 5d so that the second capacitor voltage held in the DC capacitor 4 matches the command voltage.
  • the controller 8 is responsible for this function.
  • FIG. 1 discloses a configuration in which one reactor 2 is connected to one side of the AC power supply 1, but the configuration is not limited to this configuration.
  • One reactor 2 may be connected to the other side of the AC power supply 1.
  • the two divided reactors 2 may be connected to both one side and the other side of the AC power supply 1.
  • the divided reactor 2 may be wound around the same core to form one magnetically coupled reactor, and may be connected to either one side or the other side of the AC power supply 1.
  • FIG. 2 is a diagram used to explain the concept of an operating region when the power conversion device 100 according to the first embodiment operates for boosting.
  • FIG. 3 is a diagram used to explain the concept of an operating region when the power conversion device 100 according to the first embodiment operates in a step-down operation.
  • FIG. 4 is a diagram showing the relationship between the operating region defined in FIGS. 2 and 3 and the operating state of the main circuit 110.
  • FIG. 5 is a diagram showing an example of a current path when the power conversion device 100 according to the first embodiment performs power conversion.
  • FIG. 5 is a diagram showing an example of a current path when the power conversion device 100 according to the first embodiment performs power conversion.
  • FIG. 6 is a diagram showing an example of a switching pattern when the power conversion device 100 according to the first embodiment operates for boosting.
  • FIG. 7 is a diagram showing an example of a switching pattern when the power conversion device 100 according to the first embodiment operates in a step-down operation.
  • FIG. 2 shows the voltage relationship when the smoothing capacitor voltage Vdc is larger than the absolute value
  • AC half cycle the smoothing capacitor voltage Vdc is larger than the absolute value
  • the mode in which the power conversion device 100 is stepped up is referred to as a “boosting operation mode”.
  • FIG. 3 shows the voltage relationship in the case where the smoothing capacitor voltage Vdc is smaller than the absolute value
  • the mode in which the power conversion device 100 is stepped down is referred to as a "stepping down operation mode".
  • . Further, ⁇ 1 and ⁇ 2 are phases in which Vdc
  • the operating region in which the phase ⁇ satisfies 0 ⁇ ⁇ ⁇ 1 and ⁇ 2 ⁇ ⁇ ⁇ is the operating region in which the relationship of
  • This operating area is defined as area 1.
  • the region of ⁇ 1 ⁇ ⁇ ⁇ ⁇ 2 in FIG. 2 and the operating region in FIG. 3 that satisfies ⁇ 1 ⁇ ⁇ ⁇ 1 and ⁇ 2 ⁇ ⁇ ⁇ 2 are operating regions in which Vsub ⁇
  • the operating region satisfying ⁇ 1 ⁇ ⁇ ⁇ ⁇ 2 is the operating region where Vsub ⁇ Vdc ⁇
  • FIG. 4 shows the reactor 2, the operating state of the DC capacitor 4, and the magnitude of the reactor applied voltage, which is the voltage applied to the reactor 2, corresponding to each operating region defined in FIGS. 2 and 3. ..
  • Excitation is an operation of accumulating electromagnetic energy in the reactor 2.
  • Reset is an operation of releasing the electromagnetic energy stored in the reactor 2.
  • FIG. 5 shows the current path when the reactor applied voltage becomes “vac-Vdc + Vsub” in the region 2 of FIG.
  • the arrow in FIG. 5 indicates the current path through which the alternating current iac flows.
  • the controller 8 controls the switching elements 3b, 3c, 5d to be on and the switching elements 3a, 3d, 5b to be off. Be controlled.
  • the current path of the alternating current iac can be shown as in FIG. The explanation here is omitted.
  • the lower part of FIG. 6 shows an example of a switching pattern in the boost operation mode.
  • An example of the switching pattern in the step-down operation mode is shown in the lower part of FIG. 7.
  • the smoothing capacitor voltage Vdc, the smoothing capacitor voltage command Vdc * which is the command value of the smoothing capacitor voltage Vdc, the DC capacitor voltage Vsub, and the DC which is the command value of the DC capacitor voltage Vsub are shown.
  • the switching pattern of the section A in the region 2 is the switching pattern at the time of the operation of FIG. Further, according to the switching pattern of the section B in the region 1 of FIG. 6, the operation in which the DC capacitor 4 shown in the region 1 of FIG. 4 is “through” and the reactor applied voltage is “vac” is performed.
  • FIG. 8 is a block diagram showing an internal configuration of the controller 8 according to the first embodiment.
  • the controller 8 includes an operating area determination device 9, a feed forward (FF) duty (Duty) (hereinafter referred to as “FF_Duty”) arithmetic unit 10, a DC capacitor voltage controller 11, and an addition / subtraction determination device 12. , The adder 12a, and the gate signal generator 13.
  • FF_Duty feed forward
  • FF_Duty feed forward
  • the adder 12a and the gate signal generator 13.
  • the operating area determination device 9 generates the area determination signal Sig_SP based on the detected values of the AC voltage vac, the smoothing capacitor voltage Vdc, and the DC capacitor voltage Vsub.
  • the area determination signal Sig_SP is a signal indicating in which region of FIG. 2 or FIG. 3 the operating state of the power conversion device 100 at the time of determination is.
  • the area determination signal Sig_SP generated by the operation area determination device 9 is input to the FF_Duty calculator 10, the addition / subtraction determination unit 12, and the gate signal generator 13. In each arithmetic unit, an arithmetic operation is performed according to the operating area.
  • the FF_Duty calculator 10 calculates the FF_Duty ratio D_Vdc based on the detected value of the AC voltage vac and the region determination signal Sig_SP.
  • the charging and discharging in the DC capacitor 4 are set to be the same number of times in each AC half cycle in each region shown in FIG. 2 or FIG. For example, in a certain AC half cycle, if the DC capacitor 4 is charged once, the DC capacitor 4 is also discharged once, and if the DC capacitor 4 is charged twice, the DC capacitor 4 is also discharged twice. Will be. In addition, "0" is also included in the charge / discharge count which is the total value of the charge count and the discharge count.
  • Charging and discharging in each operating area are set with reference to FIG. When the number of charge / discharge cycles is "0", "through” in FIG. 4 is selected. It is preferable, but not limited to, the set of “charging” and “discharging” of the same number of times is selected for each operating region. As long as it is within the period of the AC half cycle, the set of "charge” and “discharge” may be selected across the operating region.
  • the number of charge / discharge cycles can be determined based on the capacity of the DC capacitor 4, the withstand voltage of the main circuit component, and the power factor of the main circuit 110.
  • the switching loss can be reduced, but the responsiveness of the control deteriorates, resulting in a decrease in the power factor and an increase in the amount of voltage ripple. Therefore, in order to operate the power conversion device 100 stably, it is necessary to increase the capacity of the capacitor.
  • the capacitor capacity can be small and the responsiveness of control is improved, but the switching loss increases.
  • FIG. 9 is a block diagram showing a detailed configuration of the DC capacitor voltage controller 11 shown in FIG.
  • the DC capacitor voltage controller 11 includes a pretreatment device 11a and a sample hold device 11b.
  • the DC capacitor voltage controller 11 controls the DC capacitor voltage Vsub to follow the DC capacitor voltage command Vsub *, which is a command value of the DC capacitor voltage Vsub, based on the detected value of the DC capacitor voltage Vsub. Generate a command duty ratio D_Vsub.
  • the preprocessing device 11a proportionally controls the deviation between the DC capacitor voltage command Vsub * and the detected value of the DC capacitor voltage Vsub *, and divides the control value by the DC capacitor voltage command Vsub * to obtain a standardized duty ratio. Is calculated.
  • the sample hold device 11b updates the value of the output of the preprocessing device 11a in the sample hold cycle, and outputs the updated value to the addition / subtraction determination device 12 as the DC capacitor voltage command duty ratio D_Vsub.
  • the above-mentioned FF_Duty ratio may be referred to as a "first duty ratio”
  • the above-mentioned DC capacitor voltage command duty ratio may be referred to as a "second duty ratio”.
  • FIG. 10 is a time chart used for explaining the operation of the addition / subtraction determination device 12 shown in FIG.
  • FIG. 11 is a block diagram used for explaining the operation of the addition / subtraction determination device 12 shown in FIG.
  • the waveform in FIG. 10 is an example of the operation of region 2 in the positive half wave of the AC voltage vac.
  • FIG. 10 shows the waveforms of the alternating current iac and the waveforms of the gate signals G3a to G3d, G5b, and G5d for controlling each of the switching elements 3a to 3d, 5b, and 5d in order from the upper stage side. .. Further, the operating state of the DC capacitor 4 is shown in the lower part of FIG. 10. At times t0 to t1 and t2 to t3, the reactor 2 is excited, and at times t1 to t2 and t3 to t4, the excitation of the reactor 2 is reset. Therefore, in the example of FIG. 10, two pairs of switching control with excitation and reset as one pair, that is, four times of switching control are performed.
  • the DC capacitor voltage command duty ratio D_Vsub is input to the addition / subtraction determination device 12.
  • the addition / subtraction determination device 12 multiplies the DC capacitor voltage command duty ratio D_Vsub by a value of "1" or "-1" based on the area determination signal Sig_SP. That is, the addition / subtraction determination device 12 outputs a non-inverted control signal “+ D_Vsub” whose sign is not inverted or a control signal “ ⁇ D_Vsub” whose sign is inverted according to the area determination signal Sig_SP.
  • the upper part of FIG. 11 shows a situation in which a non-inverting control signal “+ D_Vsub” is output.
  • the lower part of FIG. 11 shows a situation in which the inverted control signal “ ⁇ D_Vsub” is output.
  • voltage control is performed by changing the time t1 and the time t3 in FIG.
  • the DC capacitor voltage Vsub becomes lower than the DC capacitor voltage command Vsub * due to disturbance, as shown in the upper part of FIG. 11, at time t1, the DC capacitor voltage with respect to the FF_Duty ratio D_Vdc in the adder 12a.
  • the command duty ratio D_Vsub is positively added.
  • the DC capacitor voltage command duty ratio D_Vsub is negatively added to the FF_Duty ratio D_Vdc in the adder 12a.
  • the DC capacitor voltage command duty ratio D_Vsub is negatively added to the FF_Duty ratio D_Vdc in the adder 12a. Will be done. Further, at time t3, the DC capacitor voltage command duty ratio D_Vsub is positively added to the FF_Duty ratio D_Vdc in the adder 12a. As a result, the operation of the main circuit 110 is such that the amount of charge to the DC capacitor 4 decreases and the amount of discharge increases.
  • the DC capacitor voltage Vsub is controlled according to the DC capacitor voltage command Vsub *.
  • the operation of the region 2 in the positive half wave of the AC voltage vac has been described in FIG. 10, the same operation is performed in the regions 1 and 3 in the positive half wave and the regions 1 to 3 in the negative half wave. It is possible to perform constant voltage control with.
  • the AC voltage vac, the area determination signal Sig_SP, and the total duty ratio D_total output from the adder 12a are input to the gate signal generator 13.
  • the gate signal generator 13 generates gate signals G3a to G3d, G5b, and G5d based on the AC voltage vac, the area determination signal Sig_SP, and the total duty ratio amount D_total output from the adder 12a.
  • the gate signals G3a to G3d, G5b, and G5d are applied to the switching elements 3a to 3d, 5b, and 5d, respectively, and the conduction of the switching elements 3a to 3d, 5b, and 5d is controlled.
  • the required power can be supplied to the load 7 through the control of the first capacitor voltage. Further, it is possible to control the voltage constant so that the second capacitor voltage follows the command value while controlling the first capacitor voltage.
  • the number of switchings in the AC half cycle is at most 20 times. That is, in the first embodiment, switching control is performed for the switching elements 3a to 3d, 5b, and 5d of the main circuit 110 by switching the number of times of switching to a dozen or less times in an AC half cycle.
  • the AC half cycle is 10 [ms].
  • the time required for one switching control is 0.5 [ms]. If 0.5 [ms] is set as one switching cycle, the switching frequency is 5 [kHz], and the switching frequency can be reduced as compared with the conventional case. As a result, the switching loss of the switching element and the high frequency loss of the reactor can be reduced, so that the power conversion device can be driven with higher efficiency than in the conventional case.
  • the DC capacitor 4 is charged and discharged at least once in the AC half cycle, but the present invention is not limited to this.
  • the main circuit 110 In the case of the boosting operation using only the region 1 and the region 2, it is possible to operate the main circuit 110 only by controlling the smoothing capacitor voltage Vdc, with the number of charge / discharge of the DC capacitor 4 in the AC half cycle being 0 times.
  • the FF_Duty calculator 10 only the “through” operation is selected in FIG. 4, and the FF_Duty ratio D_Vdc is calculated. Then, the gate signals G3a to G3d, G5b, and G5d are generated using the calculated FF_Duty ratio D_Vdc. In this case, feedback control of the DC capacitor voltage Vsub becomes unnecessary. Therefore, the number of switching times and the conduction time according to the operating conditions may be stored in the memory 8b of the controller 8 in advance, and the stored information may be read out to operate the main circuit 110.
  • the controller switches a plurality of switching elements once or more and a dozen times or less (less than 20 times) in a half cycle of the AC voltage. Switching control is performed with. As a result, the switching loss of the switching element and the high frequency loss of the reactor can be reduced, and the power conversion device can be driven with high efficiency. Further, in this switching control, the controller controls the charge amount and the discharge amount of the second capacitor in the half cycle of the AC voltage to match the second capacitor voltage with the command value of the second capacitor voltage. Take control. By this control, the power factor of the main circuit operation can be increased while reducing the switching loss of the switching element and the high frequency loss of the reactor. As a result, the power conversion device can be driven with high efficiency.
  • the controller can perform switching control while changing the charging time for charging the second capacitor and the discharging time for discharging the second capacitor. Since this control can be performed without using carrier waves, it is possible to easily carry out high-efficiency driving of the power conversion device.
  • the controller charges and discharges the second capacitor once or more within a half cycle of the AC voltage, and the number of times of charging and the number of times of discharging are within the half cycle of the AC voltage. It is preferable to perform switching control so that they are equal to each other. As a result, constant voltage control that causes the second capacitor voltage to follow the command value can be reliably performed.
  • the controller preferably adds or subtracts the second duty ratio to the first duty ratio so that the total amount of duty ratios in the half cycle of the AC voltage is kept constant. As a result, it is possible to carry out both the control of the first capacitor voltage and the constant voltage control for making the second capacitor voltage follow the command value.
  • FIG. 12 is a block diagram showing an internal configuration of the controller 8A according to the second embodiment.
  • the FF_Duty calculator 10 is replaced with the high power factor controller 17, and the gate signal generator 13 is the gate signal. It has been replaced by the generator 18.
  • a carrier wave generator 16 has been added.
  • the other configurations are the same as or equivalent to the configurations shown in FIG. 8, and the same or equivalent components are designated by the same reference numerals, and duplicate explanations are omitted.
  • the basic circuit configuration is the same as or equivalent to that in FIG.
  • FIG. 13 is a diagram showing an example of a switching pattern when the power conversion device 100 according to the second embodiment operates for boosting.
  • FIG. 14 is a diagram showing an example of a switching pattern when the power conversion device 100 according to the second embodiment operates in a step-down operation.
  • the concept of the operating area described in the first embodiment is the same in the second embodiment.
  • the power conversion device 100 according to the second embodiment performs switching control for the switching elements 3a to 3d, 5b, 5d while changing the carrier frequency for each operating region. When the carrier frequency is changed, so is the switching frequency.
  • the period in which the same switching frequency is maintained is defined as the "first switching period".
  • the DC capacitor 4 is operated so as to be charged and discharged once or more in each operating region.
  • the width of each operating region is determined by the magnitude relationship between the smoothing capacitor voltage Vdc and the DC capacitor voltage Vsub and the AC voltage vac, the width of each operating region varies. Therefore, in a region with a narrow period, power factor control is performed by switching control at a higher frequency, and in a region with a wide period, power factor control is performed by switching control at a lower frequency.
  • the number of switching times is at most 20 times, which is significantly reduced as compared with the conventional method. Therefore, it is possible to significantly reduce the loss as compared with the conventional case, and it is possible to drive the power conversion device with high efficiency.
  • the operating area increases as compared with the case of the step-up operation shown in FIG. Therefore, if the number of switchings for each operating region is the same, the number of switchings in the AC half cycle is larger in the step-down operation. Further, the step-down operation is different from the step-up operation, and it is necessary to operate while charging / discharging the DC capacitor 4. Therefore, in the region 3, switching control for charging / discharging is always performed. This point is the same in the first embodiment. As shown in region 3 of FIG. 4, there is no "through" operation in region 3, and either discharge or charge operation is selected.
  • FIG. 15 is a block diagram showing a detailed configuration of the carrier wave generator 16 shown in FIG.
  • the carrier wave generator 16 includes a switching frequency calculator 20 and a frequency converter 21.
  • the switching frequency calculator 20 calculates the time of each operating region in the AC half cycle based on the AC voltage vac, the DC capacitor voltage Vsub, and the smoothing capacitor voltage Vdc. Further, the switching frequency calculator 20 calculates a switching cycle which is the reciprocal of the switching frequency based on the number of times of charging / discharging of the DC capacitor 4 set by the user.
  • the frequency converter 21 generates a carrier wave based on the switching cycle calculated by the switching frequency calculator 20 and the region determination signal Sig_SP.
  • the carrier wave may be a sawtooth wave or a triangular wave.
  • the switching frequency calculator 20 includes a first time calculator 20a, a second time calculator 20c, a third time calculator 20e, a fourth time calculator 20i, and the like.
  • the dividers 20b, 20d, 20g, 20j and the adder 20h are provided.
  • the first time calculator 20a calculates the time Rt1 in the region 1 by dividing the DC capacitor voltage Vsub by the AC voltage vac, converting it into time by an inverse trigonometric function, and further dividing by the angular frequency 2 ⁇ fac.
  • the time Rt1 is the time corresponding to the phase difference from 0 to ⁇ 1 in FIG. fac is the frequency of the AC voltage vac.
  • the frequency of the AC voltage vac is referred to as "AC voltage frequency”.
  • the switching frequency calculator 20 calculates the switching cycle Tsw_1 of the region 1 by dividing the time Rt1 by the number of charge / discharge cycles in the divider 20b.
  • the second time calculator 20c calculates the time Rt2b of the region 2 by subtracting the double value of the time Rt1 of the region 1 from the value obtained by dividing the value of 1 by the double value of the AC voltage frequency fac.
  • the double value of the time Rt1 is the time corresponding to the phase difference obtained by adding the phase difference from 0 to ⁇ 1 in FIG. 2 and the phase difference from ⁇ 2 to ⁇ .
  • the value obtained by dividing the value of 1 by a double value of the AC voltage frequency fac is the time corresponding to the phase difference from 0 to ⁇ in FIG. Therefore, by the processing of the second time calculator 20c, the time Rt2b of the region 2 corresponding to the phase difference from ⁇ 1 to ⁇ 2 in FIG. 2 is calculated.
  • the switching frequency calculator 20 calculates the switching cycle Tsw_2b of the region 2 in the boost operation mode by dividing the time Rt2b by the number of charges and discharges in the divider 20d.
  • the third time calculator 20e divides the smoothing capacitor voltage Vdc by the AC voltage vac, converts it into time by an inverse trigonometric function, further divides it by the angular frequency 2 ⁇ fac, and further subtracts the time Rt1 in the region 1.
  • the time Rt2s of the region 2 is calculated.
  • the value divided by the angular frequency 2 ⁇ fac is the time corresponding to the phase difference from 0 to ⁇ 1 in FIG. Therefore, the time Rt2s of the region 2 can be calculated by subtracting the time Rt1 of the region 1 from the value divided by the angular frequency 2 ⁇ fac.
  • the switching frequency calculator 20 calculates the switching cycle Tsw_2s of the region 2 in the step-down operation mode by dividing the time Rt2s by the number of charges and discharges in the divider 20g.
  • the adder 20h recalculates the time corresponding to the phase difference from 0 to ⁇ 1 in FIG. 3 by adding the time Rt2s in the region 2 and the time Rt1 in the region 1.
  • the fourth time calculator 20i calculates the time Rt3 in the region 3 by subtracting the double value of the output of the adder 20h from the value obtained by dividing the value of 1 by the double value of the AC voltage frequency fac.
  • the double value of the output of the adder 20h is the time corresponding to the phase difference obtained by adding the phase difference from 0 to ⁇ 1 in FIG. 3 and the phase difference from ⁇ 2 to ⁇ .
  • the value obtained by dividing the value of 1 by a double value of the AC voltage frequency fac is the time corresponding to the phase difference from 0 to ⁇ in FIG. Therefore, by the processing of the fourth time calculator 20i, the time Rt3 of the region 3 corresponding to the phase difference from ⁇ 1 to ⁇ 2 in FIG. 3 is calculated.
  • the switching frequency calculator 20 calculates the switching cycle Tsw_3 of the region 3 in the step-down operation mode by dividing the time Rt3 by the number of charges and discharges in the divider 20j.
  • the switching frequency calculator 20 outputs the calculated switching cycles Tsw_1, Tsw_2b, Tsw_2s, and Tsw_3 to the frequency converter 21, but is not limited to this.
  • the reciprocals of the switching cycles Tsw_1, Tsw_2b, Tsw_2s, and Tsw_3 may be calculated, and each calculated value may be output to the frequency converter 21 as the switching frequency corresponding to the operating region.
  • the switching cycles Tsw_1, Tsw_2b, Tsw_2s, and Tsw_3 are used as switching frequency information
  • the times Rt1, Rt2b, Rt2s, and Rt3 can be regarded as reference switching frequencies for calculating each switching frequency.
  • the switching frequency calculator 20 shown in FIG. 15 the front stage portion calculates the reference switching frequency corresponding to the operating region, and the rear stage portion determines the reference switching frequency and the number of times of charging / discharging of the second capacitor. It is configured to calculate the switching frequency based on it.
  • the frequency converter 21 generates a carrier wave corresponding to the operating region based on the switching cycles Tsw_1, Tsw_2b, Tsw_2s, Tsw_3 calculated by the switching frequency calculator 20 and the region determination signal Sig_SP, and is a gate signal generator. Output to 18.
  • the number of charge / discharge cycles is set by an integer including 0 so as to be 1 or more in an AC half cycle.
  • the number of charge / discharge cycles can be determined based on the capacity of the DC capacitor 4, the withstand voltage of the main circuit component, and the power factor of the main circuit 110.
  • the switching loss can be reduced, but the responsiveness of the control deteriorates, resulting in a decrease in the power factor and an increase in the amount of voltage ripple. Therefore, in order to operate the power conversion device 100 stably, it is necessary to increase the capacity of the capacitor.
  • the capacitor capacity can be small and the responsiveness of control is improved, but the switching loss increases.
  • FIG. 16 is a block diagram showing a detailed configuration of the high power factor controller 17 shown in FIG.
  • the high power factor controller 17 includes a current command calculator 17a, a current controller 17b, an FF_Duty calculator 17c, and an adder 17d.
  • the high power factor controller 17 commands the smoothing capacitor voltage Vdc while controlling the power factor of the main circuit 110 to approach 1 based on the detected values of the AC voltage vac, the smoothing capacitor voltage Vdc, and the AC current iac.
  • a control signal D_PFC that controls to follow the value is generated.
  • the current command calculator 17a calculates the current command amplitude Iac * by controlling the deviation between the smoothing capacitor voltage command Vdc * and the detected value of the smoothing capacitor voltage Vdc by proportional integral (PI).
  • the current command calculator 17a multiplies the current command amplitude Iac * by the AC voltage vac generated by the phase-locked loop (PLL) control and the sinusoidal signal Sin ( ⁇ t) having the same phase, and the AC current command iac. * Is calculated.
  • the current controller 17b performs PI control of the deviation between the AC current command iac * and the AC current iac, and calculates the standardized control duty ratio D_PFC1 by dividing the control value by the DC capacitor voltage command Vsub *.
  • the adder 17d adds the control duty ratio D_PFC1 and the FF_Duty ratio D_PFC_FF calculated by the FF_Duty calculator 17c, and outputs the added value as the FF_Duty ratio D_PFC for high power rate control to the adder 12a in FIG. do.
  • the control duty ratio D_PFC1 is switched according to the operating region. Therefore, by inserting the FF control, the current fluctuation at the time of switching the control can be suppressed.
  • the FF_Duty ratio D_PFC_FF for FF control calculates the theoretical duty ratio so that the amount of increase / decrease in the AC current iac due to the excitation and reset of the reactor 2 becomes equal.
  • the method for calculating the theoretical duty ratio is described in detail in Patent Document 1 described above, so please refer to the description. The contents of the description are incorporated in the present specification and form a part of the present specification. The method for calculating the theoretical duty ratio is not limited to the contents described in the publication, and any method may be used as long as the theoretical duty ratio can be obtained.
  • a set of excitation and reset is selected so that the charging and discharging of the DC capacitor 4 have the same number of times in the AC half cycle.
  • the FF_Duty calculator 17c calculates the theoretical duty ratio based on information about the selected excitation and reset pairs.
  • FIG. 17 is a block diagram showing a detailed configuration of the gate signal generator 18 shown in FIG.
  • the gate signal generator 18 includes a comparison unit 18a and a pulse calculator 18b. Further, the comparison unit 18a includes a first comparator 18a1, a multiplier 18a2, and a second comparator 18a3.
  • the total duty ratio amount D_total is input to the + terminal of the second comparator 18a3 via the multiplier 18a2, and the carrier wave is input to the-terminal of the second comparator 18a3.
  • the total duty ratio D_total and the amplitude value of the carrier wave are compared, and if the total duty ratio D_total is larger than the amplitude value of the carrier wave, an on signal for conducting the switching element is generated. ..
  • the pulse calculator 18b generates gate signals G3a to G3d, G5b, and G5d using the on signal output from the comparison unit 18a and the region determination signal Sig_SP.
  • the gate signals G3a to G3d, G5b, and G5d are applied to the switching elements 3a to 3d, 5b, and 5d, respectively, and the conduction of the switching elements 3a to 3d, 5b, and 5d is controlled.
  • a first comparator 18a1 and a multiplier 18a2 are provided in order to realize control when the number of charge / discharge cycles is 0.
  • the output of the first comparator 18a1 becomes 0 and the output of the multiplier 18a2 also becomes 0. Therefore, the total duty ratio input to the second comparator 18a3 is D_total. Is also 0.
  • the configuration of FIG. 17 is an example and is not limited to these configurations. For example, an on-signal for conducting the switching element may be generated when the total duty ratio D_total is smaller than the amplitude value of the carrier wave.
  • the required power can be supplied to the load 7 through the control of the first capacitor voltage. Further, it is possible to control the voltage constant so that the second capacitor voltage follows the command value while controlling the first capacitor voltage.
  • the number of switchings in the AC half cycle is at most 20 times. Therefore, in the second embodiment, the first capacitor voltage is controlled by the switching control of the switching elements 3a to 3d, 5b, 5d of the main circuit 110 by the number of switchings of 10 or less times in the AC voltage half cycle. It is possible to carry out both constant voltage control that causes the second capacitor voltage to follow the command value.
  • the controller responds to the magnitude relationship between the detected values of the first and second capacitor voltages and the detected values of the AC voltage.
  • the operating region is determined, and switching control is performed while changing the switching frequency for each operating region based on the detected values of the first and second capacitor voltages and the detected value of the AC voltage.
  • the controller charges and discharges the second capacitor once or more in the first switching period, which is the period in which the same switching frequency is maintained, and the number of times of charging. It is preferable to perform switching control so that the number of discharges becomes equal within the first switching period. As a result, constant voltage control that causes the second capacitor voltage to follow the command value can be reliably performed.
  • the controller may add or subtract the second duty ratio to the first duty ratio so that the total amount of duty ratios in the first switching period is kept constant. As a result, it is possible to carry out both the control of the first capacitor voltage and the constant voltage control for making the second capacitor voltage follow the command value.
  • the controller refers from the first phase at which the AC voltage and the second capacitor voltage intersect and the second phase at which the AC voltage and the first capacitor voltage intersect for each operating region. Calculate the switching frequency.
  • the controller may calculate the switching frequency based on the predetermined number of times of charging and discharging of the second capacitor and the reference switching frequency obtained by the calculation.
  • the controller has a theory that the amount of increase / decrease in the alternating current due to the excitation and reset of the reactor becomes equal to the feedback duty ratio for controlling the second capacitor voltage within the first switching period.
  • a duty ratio may be added. This makes it possible to smoothly switch between operating areas.
  • FIG. 18 is a diagram showing a basic circuit configuration of the power conversion device 100A according to the third embodiment.
  • the diode bridge 24 is provided on the AC power supply 1 side of the reactor 2 in the configuration of the power conversion device 100 according to the first embodiment shown in FIG.
  • the switching elements 3a and 3d provided in the subconverter 3 of the main circuit 110 are replaced with the diodes 3a'and 3d', respectively.
  • the switching element 5d provided in the main converter 5 of the main circuit 110 is replaced with the diode 5d'.
  • the other configurations are the same as or equivalent to the configurations shown in FIG. 1, and the same or equivalent components are designated by the same reference numerals, and duplicate explanations are omitted.
  • a voltage waveform rectified by a diode bridge 24 is applied to the subconverter 3. Therefore, the operation of the controller 8 in the third embodiment has no negative half-wave operation and is only a positive half-wave operation. Therefore, the function of the controller 8 in the first embodiment or the second embodiment can be used as it is.
  • the positive half-wave operation is the same as that of the first and second embodiments, and the description thereof is omitted here.
  • the number of switching elements can be reduced as compared with the first embodiment and the second embodiment, so that the switching loss can be reduced. Further, since the diode can be obtained at a lower cost than the switching element, the cost of the device can be reduced.
  • the operation of the controller 8 is only a positive half wave operation. Therefore, the function of the controller 8 can be simplified as compared with the first embodiment and the second embodiment. This makes it possible to reduce the cost of the device.
  • FIG. 19 is a diagram showing a configuration example of the motor drive device 150 according to the fourth embodiment.
  • the inverter 7a and the motor 7b are added to the configuration of the power conversion device 100 shown in FIG.
  • a motor 7b is connected to the output side of the inverter 7a.
  • the motor 7b is an example of a load device.
  • the inverter 7a drives the motor 7b by converting the DC power stored in the smoothing capacitor 6 into AC power and supplying the converted AC power to the motor 7b.
  • the motor drive device 150 shown in FIG. 19 can be applied to products such as blowers, compressors and air conditioners.
  • the power conversion device 100 according to the first embodiment is applied to configure the motor drive device 150, but the present invention is not limited to this.
  • the power conversion device 100 according to the first embodiment the power conversion device 100 according to the second embodiment or the power conversion device 100A according to the third embodiment may be used.
  • FIG. 20 is a diagram showing an example in which the motor drive device 150 shown in FIG. 19 is applied to an air conditioner.
  • a motor 7b is connected to the output side of the motor drive device 150, and the motor 7b is connected to the compression element 504.
  • the compressor 505 includes a motor 7b and a compression element 504.
  • the refrigeration cycle unit 506 is configured to include a four-way valve 506a, an indoor heat exchanger 506b, an expansion valve 506c, and an outdoor heat exchanger 506d.
  • the flow path of the refrigerant circulating inside the air conditioner is from the compression element 504 via the four-way valve 506a, the indoor heat exchanger 506b, the expansion valve 506c, the outdoor heat exchanger 506d, and again via the four-way valve 506a. , It is configured to return to the compression element 504.
  • the motor drive device 150 receives electric power from the AC power supply 1 and rotates the motor 7b.
  • the compression element 504 can execute the compression operation of the refrigerant by rotating the motor 7b, and the refrigerant can be circulated inside the refrigeration cycle unit 506.
  • the motor drive device 150 according to the fourth embodiment is configured to include the power conversion device according to the first to third embodiments. Thereby, in the products such as the blower, the compressor and the air conditioner to which the motor drive device according to the fourth embodiment is applied, the effects described in the first to third embodiments can be obtained.

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Rectifiers (AREA)
  • Inverter Devices (AREA)

Abstract

L'invention concerne un dispositif de conversion de puissance (100) pourvu : d'un réacteur (2) ; d'éléments de commutation (3a-3d, 5b, 5d) ; d'un condensateur de lissage (6) ; d'un condensateur CC (4) disposé entre le réacteur (2) et le condensateur de lissage (6) ; et d'un dispositif de commande (8) pour commander la conduction des éléments de commutation (3a-3d, 5b, 5d). Le dispositif de conversion de puissance (100) effectue une conversion de puissance entre une tension alternative et une tension de condensateur de lissage. Le dispositif de commande (8) effectue une commande de commutation des éléments de commutation (3a-3d, 5b, 5d) avec le nombre de temps de commutation de 1 à 20, inclus, dans un demi-cycle de la tension alternative.
PCT/JP2020/020782 2020-05-26 2020-05-26 Dispositif de conversion de puissance, dispositif d'entraînement de moteur, soufflante, compresseur et climatiseur WO2021240657A1 (fr)

Priority Applications (3)

Application Number Priority Date Filing Date Title
JP2022527330A JP7297157B2 (ja) 2020-05-26 2020-05-26 電力変換装置、モータ駆動装置、送風機、圧縮機及び空気調和機
PCT/JP2020/020782 WO2021240657A1 (fr) 2020-05-26 2020-05-26 Dispositif de conversion de puissance, dispositif d'entraînement de moteur, soufflante, compresseur et climatiseur
JP2023018913A JP7399331B2 (ja) 2020-05-26 2023-02-10 電力変換装置、モータ駆動装置、送風機、圧縮機及び空気調和機

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
PCT/JP2020/020782 WO2021240657A1 (fr) 2020-05-26 2020-05-26 Dispositif de conversion de puissance, dispositif d'entraînement de moteur, soufflante, compresseur et climatiseur

Publications (1)

Publication Number Publication Date
WO2021240657A1 true WO2021240657A1 (fr) 2021-12-02

Family

ID=78723047

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/JP2020/020782 WO2021240657A1 (fr) 2020-05-26 2020-05-26 Dispositif de conversion de puissance, dispositif d'entraînement de moteur, soufflante, compresseur et climatiseur

Country Status (2)

Country Link
JP (2) JP7297157B2 (fr)
WO (1) WO2021240657A1 (fr)

Citations (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP6529707B1 (ja) * 2018-12-13 2019-06-12 三菱電機株式会社 電力変換装置

Family Cites Families (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH0789742B2 (ja) * 1987-02-18 1995-09-27 三菱電機株式会社 電力変換装置
JPH11164561A (ja) * 1997-11-26 1999-06-18 Toshiba Corp 直流電源装置および空気調和機
JP2000278955A (ja) * 1999-01-19 2000-10-06 Matsushita Electric Ind Co Ltd 電源装置及びこの電源装置を用いた空気調和機
JP4337316B2 (ja) * 2001-09-28 2009-09-30 ダイキン工業株式会社 電力変換装置
DE112012003766T5 (de) 2011-09-08 2014-09-18 Mitsubishi Electric Corp. Energie-Umwandlungsvorrichtung
JP6712104B2 (ja) 2015-09-10 2020-06-17 日立ジョンソンコントロールズ空調株式会社 直流電源装置および空気調和機

Patent Citations (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP6529707B1 (ja) * 2018-12-13 2019-06-12 三菱電機株式会社 電力変換装置

Also Published As

Publication number Publication date
JPWO2021240657A1 (fr) 2021-12-02
JP7399331B2 (ja) 2023-12-15
JP7297157B2 (ja) 2023-06-23
JP2023054066A (ja) 2023-04-13

Similar Documents

Publication Publication Date Title
JP6026049B2 (ja) 電力変換装置
JP5626435B2 (ja) 直接形交流電力変換装置
JP5958531B2 (ja) インバータ装置
JP6569839B1 (ja) 電力変換装置
US9184674B2 (en) Power conversion apparatus that provides power conversion between AC and DC power
JP2016029884A (ja) 直接形交流電力変換装置
US11811332B2 (en) Direct-current power supply apparatus, motor drive control apparatus, blower, compressor, and air conditioner
JP6103031B1 (ja) 電力変換装置の制御装置
WO2021240657A1 (fr) Dispositif de conversion de puissance, dispositif d'entraînement de moteur, soufflante, compresseur et climatiseur
WO2021240658A1 (fr) Dispositif de conversion de puissance, dispositif d'entraînement de moteur, soufflante, compresseur et appareil de climatisation
JP6968315B1 (ja) 電力変換装置
US11967909B2 (en) Power conversion device performing first and second conversion operations
US10243498B2 (en) Power converter control device
JP5950970B2 (ja) 電力変換装置
US20230283219A1 (en) Power converting apparatus, motor driving apparatus, blower, compressor, and air conditioner
JP7267450B2 (ja) 電力変換装置、モータ駆動装置、送風機、圧縮機及び空気調和機
JP7130024B2 (ja) 電力変換装置
Neba et al. Single-phase voltage-quadrupler rectifier with sinusoidal input current

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 20938359

Country of ref document: EP

Kind code of ref document: A1

ENP Entry into the national phase

Ref document number: 2022527330

Country of ref document: JP

Kind code of ref document: A

NENP Non-entry into the national phase

Ref country code: DE

122 Ep: pct application non-entry in european phase

Ref document number: 20938359

Country of ref document: EP

Kind code of ref document: A1