WO2021057730A1 - 混合定位电磁感应式位移传感器 - Google Patents

混合定位电磁感应式位移传感器 Download PDF

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Publication number
WO2021057730A1
WO2021057730A1 PCT/CN2020/116882 CN2020116882W WO2021057730A1 WO 2021057730 A1 WO2021057730 A1 WO 2021057730A1 CN 2020116882 W CN2020116882 W CN 2020116882W WO 2021057730 A1 WO2021057730 A1 WO 2021057730A1
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Prior art keywords
phase
pitch
winding
displacement
circuit
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PCT/CN2020/116882
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English (en)
French (fr)
Inventor
陆取辉
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桂林广陆数字测控有限公司
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Priority to PL20867052.1T priority Critical patent/PL3865812T3/pl
Priority to RU2021113638A priority patent/RU2759209C1/ru
Priority to US17/421,331 priority patent/US11965757B2/en
Priority to EP20867052.1A priority patent/EP3865812B1/en
Priority to KR1020217014570A priority patent/KR102650230B1/ko
Priority to JP2021531884A priority patent/JP7076645B2/ja
Publication of WO2021057730A1 publication Critical patent/WO2021057730A1/zh

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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01B—MEASURING LENGTH, THICKNESS OR SIMILAR LINEAR DIMENSIONS; MEASURING ANGLES; MEASURING AREAS; MEASURING IRREGULARITIES OF SURFACES OR CONTOURS
    • G01B7/00—Measuring arrangements characterised by the use of electric or magnetic techniques
    • G01B7/02—Measuring arrangements characterised by the use of electric or magnetic techniques for measuring length, width or thickness
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01D—MEASURING NOT SPECIALLY ADAPTED FOR A SPECIFIC VARIABLE; ARRANGEMENTS FOR MEASURING TWO OR MORE VARIABLES NOT COVERED IN A SINGLE OTHER SUBCLASS; TARIFF METERING APPARATUS; MEASURING OR TESTING NOT OTHERWISE PROVIDED FOR
    • G01D5/00—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable
    • G01D5/12—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means
    • G01D5/14—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage
    • G01D5/20—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage by varying inductance, e.g. by a movable armature
    • G01D5/204—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage by varying inductance, e.g. by a movable armature by influencing the mutual induction between two or more coils
    • G01D5/2086—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage by varying inductance, e.g. by a movable armature by influencing the mutual induction between two or more coils by movement of two or more coils with respect to two or more other coils
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01B—MEASURING LENGTH, THICKNESS OR SIMILAR LINEAR DIMENSIONS; MEASURING ANGLES; MEASURING AREAS; MEASURING IRREGULARITIES OF SURFACES OR CONTOURS
    • G01B7/00—Measuring arrangements characterised by the use of electric or magnetic techniques
    • G01B7/02—Measuring arrangements characterised by the use of electric or magnetic techniques for measuring length, width or thickness
    • G01B7/023—Measuring arrangements characterised by the use of electric or magnetic techniques for measuring length, width or thickness for measuring distance between sensor and object
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01D—MEASURING NOT SPECIALLY ADAPTED FOR A SPECIFIC VARIABLE; ARRANGEMENTS FOR MEASURING TWO OR MORE VARIABLES NOT COVERED IN A SINGLE OTHER SUBCLASS; TARIFF METERING APPARATUS; MEASURING OR TESTING NOT OTHERWISE PROVIDED FOR
    • G01D5/00—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable
    • G01D5/12—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means
    • G01D5/14—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage
    • G01D5/20—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage by varying inductance, e.g. by a movable armature
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01B—MEASURING LENGTH, THICKNESS OR SIMILAR LINEAR DIMENSIONS; MEASURING ANGLES; MEASURING AREAS; MEASURING IRREGULARITIES OF SURFACES OR CONTOURS
    • G01B7/00—Measuring arrangements characterised by the use of electric or magnetic techniques
    • G01B7/003—Measuring arrangements characterised by the use of electric or magnetic techniques for measuring position, not involving coordinate determination
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01B—MEASURING LENGTH, THICKNESS OR SIMILAR LINEAR DIMENSIONS; MEASURING ANGLES; MEASURING AREAS; MEASURING IRREGULARITIES OF SURFACES OR CONTOURS
    • G01B7/00—Measuring arrangements characterised by the use of electric or magnetic techniques
    • G01B7/16—Measuring arrangements characterised by the use of electric or magnetic techniques for measuring the deformation in a solid, e.g. by resistance strain gauge
    • G—PHYSICS
    • G06—COMPUTING; CALCULATING OR COUNTING
    • G06F—ELECTRIC DIGITAL DATA PROCESSING
    • G06F17/00—Digital computing or data processing equipment or methods, specially adapted for specific functions
    • G06F17/10—Complex mathematical operations
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
    • H01F38/00—Adaptations of transformers or inductances for specific applications or functions
    • H01F38/14—Inductive couplings

Definitions

  • the invention relates to an electromagnetic induction type displacement measurement technology, in particular to an electromagnetic induction type displacement sensor realized by a hybrid positioning method.
  • Patent US5804963 discloses an incremental electromagnetic induction displacement sensor, which completes displacement measurement by cyclically driving three groups of inducing windings and receiving signals from two groups of inductive windings that are equivalently orthogonal to the inducing winding being driven.
  • Patent CN100491896C improves the structure of the induction winding and the induction winding in US5804963, so that the induction conductor (winding) and the induction conductor (winding) are no longer shared but are arranged separately to reduce the direct coupling between the two.
  • Patent US5886519 discloses an absolute position electromagnetic induction displacement sensor, which derives the measurement result by sequentially measuring the spatial phase of the measured position in multiple different wavelengths.
  • Patent CN1198111C improves the scale structure and the layout of each transmitting winding in US5886519.
  • the new scale uses multiple interconnected coupling loops to generate positive and negative symmetrical spatial magnetic fields in the receiving winding area, and the measurement method is unchanged.
  • the absolute position electromagnetic induction type displacement sensor has no minimum measurement frequency requirement, so it can work intermittently at long intervals, and the average working current is very small; the disadvantage is that its limited range restricts its application range, such as the slot width of a conventional caliper
  • the wavelength layout is only about 12.5mm and can accommodate at most two improved structures. With the current manufacturing process, it cannot be applied to conventional calipers over 200mm.
  • the purpose of the present invention is to solve the technical shortcomings of the existing electromagnetic induction displacement sensor, such as large working current, limited range, high production cost, and high manufacturing difficulty, and discloses an existing electromagnetic induction displacement sensor.
  • a hybrid positioning electromagnetic induction displacement sensor that works intermittently, has no range limit and is easy to implement.
  • Hybrid positioning electromagnetic induction type displacement sensor characterized in that the sensor includes a transceiver board (1) and an excitation board (2) that can move relative to each other along the measurement path;
  • the transceiver board (1) is arranged with a measuring circuit and at least one transmitting winding extending along the measuring path and at least two three-phase receiving windings with different pitches, one more than the number of transmitting windings; each transmitting winding surrounds two at the same time. Three-phase receiving windings with different pitches. Each three-phase receiving winding includes three phase windings with the same structure and a phase difference of 120° in sequence.
  • the measurement circuit includes There are a central control unit, an interface unit and a measurement unit;
  • the central control unit includes a microcontroller (13);
  • the interface unit includes a key input circuit connected to the microcontroller (13), a liquid crystal drive circuit, a measurement interface circuit and a power conversion circuit ;
  • the excitation board (2) is arranged with at least two rows of excitation coils that are equal in number to the three-phase receiving windings on the transceiver board (1) along the measurement path, and each row of excitation coils corresponds to the corresponding one on the transceiver board (1).
  • the pitch of the three-phase receiving windings is equal, the center lines are coincident, and the size along the measuring path is half of the respective pitches.
  • the measurement unit includes an oscillator, a frequency dividing circuit (3), a signal generator composed of a driving and sampling pulse forming circuit (4) and a line voltage scanning control signal generator (5), and an analog switch group (6) , Sample-and-hold capacitors (C1, C2), differential amplifiers (7), low-pass filters (8), zero-crossing detectors (9) composed of analog signal processing circuits, synchronous delay circuits, addition counters (10), Phase quantization circuit composed of random access memory (11, 12) and synchronous capture circuit, and transmit winding drive power tube (T1, T2), two sets of analog signal processing circuit, random access memory and synchronous capture circuit each form two parallel processing aisle;
  • the oscillator directly or through the frequency dividing circuit (3) is the driving and sampling pulse forming circuit (4), the line voltage scanning control signal generator (5), the low-pass filter (8), the synchronous capture circuit and the addition counter (10) Provide input clock; driving and sampling pulse forming circuit (4) is connected to the analog switch group (6) and at the same time is connected directly or via a multiple switch (S14) to the transmitting winding driving power tube (T1, T2), line voltage scanning control
  • the signal generator (5) is respectively connected to the analog switch group (6) and the synchronization delay circuit; the analog switch group (6), differential amplifier (7), low-pass filter (8), zero-crossing detector (9), synchronization
  • the capture circuits are connected in sequence, the sample and hold capacitors (C1, C2) are connected between the input of the differential amplifier (7) and the analog signal ground; the synchronous delay circuit is connected to the synchronous capture circuit and the addition counter (10), the addition counter ( 10)
  • the synchronous capture circuit is connected to the random access memory (11, 12) at the same time
  • Each transmitting winding uses two approximately closed coils to surround two three-phase receiving windings with different pitches in series in the same direction; all excitation coils adopt the shape of a short-circuit ring; each phase receiving winding is composed of at least 2
  • the M sub-windings with the same structure but with a phase shift of 60°/M are connected in series to form a distributed winding.
  • Each sub-winding that constitutes the distributed winding can also be a distributed winding; each time the measured position is measured in parallel.
  • the transceiver board (1) and the excitation board (2) can move relative to each other along the measurement axis; the transceiver board (1) is arranged with a transmitter winding (1.3) and two Three-phase receiving windings (1.1, 1.2) with different pitches; the transmitting winding (1.3) is surrounded by two approximately closed rectangular coils (1.3.1, 1.3.2) in series in the same direction, and the pitch is P 1
  • the three-phase receiving winding (1.1) and the three-phase receiving winding (1.2) with a pitch of P 2 are arranged on the excitation plate (2).
  • Two rows of excitation coils (2.1, 2.2) are arranged along the measuring axis, which are connected to the transceiver board respectively.
  • the two three-phase receiving windings (1.1, 1.2) on the upper side have the same pitch, the center lines coincide, and the shape of the excitation coil adopts a rounded rectangular short-circuit ring;
  • the transceiver board (1) and the excitation board (2) can rotate relative to each other around the axis, and the pitch is calculated by angle;
  • the transceiver board (1) is arranged with a transmitter that spreads along a concentric arc The winding (1.3) and two three-phase receiving windings (1.1, 1.2) with different pitches;
  • the transmitting winding (1.3) uses two approximately closed concentric arc-shaped coils (1.3.1, 1.3.2) in series in the same direction manner to surround each of the pitch P of the three-phase winding 1 received (1.1) and a pitch P 2 receives the three-phase winding (1.2);
  • the transceiver board (1) and the excitation board (2) can move relative to each other along the measurement axis; the transceiver board (1) is arranged with two transmitter windings (1.4, 1.5) that are deployed along the measurement axis. ) And three three-phase receiving windings (1.1, 1.2, 1.3) with different pitches, the first transmitting winding (1.4) uses two approximately closed rectangular coils (1.4.1, 1.4.2) in series in the same direction a three-phase manner surround each pitch P 1 of the reception winding (1.1) and a pitch P 2 receives the three-phase winding (1.2), the second transmitting winding (1.5) is closed by two approximately rectangular coils (1.4.
  • 1,1.5.2) to surround the same pitch, respectively, in series to the three-phase receiver windings.
  • 1 is P (1.1) and receiving a three-phase winding pitch P (1.3) 3; a field plate disposed on (2)
  • the three rows of excitation coils (2.1, 2.2, 2.3) spread out along the measuring axis are respectively the same in pitch with the three three-phase receiving windings (1.1, 1.2, 1.3) on the transceiver board (1), and the center lines coincide.
  • the shape uses a rounded rectangular short-circuit ring.
  • the driving and sampling pulse forming circuit (4) includes an odd number of cascaded inverters and a NAND gate (G NA ); the first clock signal (S CLK ) output by the frequency dividing circuit (3)
  • the trigger pulse of this circuit is connected to the input terminal of the first inverter and one input terminal of the NAND gate (G NA ).
  • the output of the cascaded odd number of inverters and the NAND gate (G NA ) The other input terminal is connected.
  • the NAND gate (G NA ) At each rising edge of the trigger pulse (S CLK ), the NAND gate (G NA ) outputs a narrow pulse of negative polarity, the width of which is equal to the total transmission delay of the cascaded odd number of inverters ;
  • the output (Y) of the NAND gate (G NA ) is inverted to obtain a positive polarity sample and hold control signal (SAH), and the output (Y) of the NAND gate (G NA ) is inverted and buffered to obtain a positive polarity drive Signal (TG).
  • SAH positive polarity sample and hold control signal
  • TG positive polarity drive Signal
  • the line voltage scanning control signal generator (5) is composed of 4 D-type flip-flops (FF 11 , FF 12 , FF 13 , FF 14 ); the second clock signal ( P CLK ) is connected to the clock terminals of the 4 D-type flip-flops at the same time, and the initialization signal (INI) output by the microcontroller (13) presets the 4 D-type flip-flops to 1, 0, 0, 1;
  • the first three D-type flip-flops (FF 11 , FF 12 , FF 13 ) constitute a circular counter with a cyclic shift, and the three output signals (Q 0 , Q 1 , Q 2 ) are high in turn;
  • the fourth D-type flip-flop (FF 14 ) counts the output (Q 0 ) of the first D-type flip-flop (FF 11 ), and each cycle of the ring counter causes its output (Q 3 ) to flip once .
  • the analog switch group (6) forms a three-stage switch in series structure, and completes the selection, exchange and sampling of the input three-phase voltage in sequence: the first-stage switch (S 1 -S 6 ) is configured as ⁇ S 1 , S 4 ⁇ , ⁇ S 2 , S 5 ⁇ and ⁇ S 3 , S 6 ⁇ three pairs, respectively select the phase voltage pair ⁇ u A , u B ⁇ , ⁇ u A , u C ⁇ and ⁇ u B , u C ⁇ ; the second-level switch (S 7- S 8 ) exchanges the sequence of the phase voltage pairs selected by the first-level switch as needed; the third-level switch (S 9- S 10 ) outputs one of the second-level switches Sampling the phase voltage and storing the results in the sample and hold capacitors (C 1 , C 2 ) respectively;
  • the differential amplifier (7) performs subtraction and amplification on the two phase voltage samples stored in the sample and hold capacitors (C 1 , C 2 ) to obtain a corresponding sampling and zero-order hold signal of a line voltage; Under the control of the output signal of the signal generator, the line voltage cyclic scanning sampling in the order of A—B, A—C, B—C, B—A, C—A, C—B is obtained in sequence, and the input of the three receiving phase winding do not change with time (when the sensor does not move), but to do with the measured position of the phase voltage cycle to a discrete-time synthetic sinusoidal signal (u s (n)), said measured position in the three-phase receiver windings the pitch of the phase space is converted to the discrete-time sine signal (u s (n)) of the initial phase.
  • the synchronous delay circuit is composed of 3 D-type flip-flops (FF 21 , FF 22 , FF 23 ); the last output signal (Q 3 ) of the line voltage scanning control signal generator (5) and the first
  • the clock terminal of the D-type flip-flop (FF 21 ) is connected as the input clock of the circuit, and the initialization signal (INI) output by the microcontroller (13) presets the three D-type flip-flops to 1, 1, respectively.
  • the first two D-type flip-flops (FF 21 , FF 22 ) form a 2-bit asynchronous subtraction counter, which counts down the rising edge of the input clock (Q 3 ); When the rising edge of the fourth input clock (Q 3 ) arrives, make the output signal (C E ) jump to a high level;
  • the addition counter (10) starts counting from 0, and the synchronous capture circuit releases the capture lock
  • the synchronous capture circuit is composed of 2 D-type flip-flops (FF 31 , FF 32 ) and an AND gate (AG 31 ); the initialization signal (INI) output by the microcontroller (13) triggers the 2 D-type flip-flops
  • the device (FF 31 , FF 32 ) is asynchronously cleared to 0, and the capture of the counting value of the addition counter (10) is blocked when the delay time has not expired; after the delay time is reached, the output signal (C) of the synchronous delay circuit E ) changes to a high level, and the square wave signal (U Z ) output by the analog signal processing circuit changes the output signal (C) of the first D-type flip-flop (FF 31) after the first rising edge S ) is also set to high level, and the output signal (C P ) of the AND gate (AG 31 ) is the same as the square wave signal (U Z ) thereafter; the second D-type flip-flop (FF 32 ) will The output signal (C P ) of the AND gate (AG 31 ) and the
  • the following absolute positioning algorithms can also be used: map the middle displacement x M to the semi
  • the phase quantization codes N 11 and N 12 in the two different pitches P 1 and P 2 surrounded by the first transmitting winding (1.4) are measured in parallel, and the phase quantization codes N 11 and N 12 are measured in parallel.
  • the distance P 1 is the fine distance P F )
  • the middle displacement x M m ⁇ (N 12 -N 11 )
  • the coarse displacement x C m ⁇ n ⁇ (N 23 -N 21 )
  • the coarse displacement x C is mapped to half closed interval [0, P C)
  • the pitch P M K M is an integer number from the relationship x C ⁇ K M ⁇ P M + x M x C obtained contained coarse offset
  • ⁇ K F ⁇ x M according to the relationship P F +x F find the integer number K F of the fine pitch P F contained in
  • the present invention can be used for linear displacement measurement and angular displacement measurement; it can use a 2-pitch structure or a 3-pitch structure; it can use a hybrid positioning algorithm or an absolute Location Algorithm.
  • the difference frequency operation is performed on the spatial frequencies of two different pitches, and the longer spatial period corresponding to the frequency difference—medium pitch or coarse pitch can be obtained; the measured position is in the middle pitch and coarse pitch.
  • the displacement within the pitch can be directly obtained by subtracting and amplifying the spatial phase of the measured position in the two different pitches of the difference frequency operation (absolute positioning) without relying on the measurement history or process.
  • the mid-pitch scale is limited.
  • the total displacement of the mid-pitch wavelength ratio without range limitation can be obtained by accumulating the mid-pitch displacement increments of two adjacent measurements; when using a 3-pitch structure
  • the scale of the coarse pitch can generally meet the requirements of conventional measurement, but in the application of a large range (beyond the range of the coarse pitch), it can also be obtained by accumulating the absolute displacement increments of two adjacent measurements to obtain no range limit and step The measured displacement at a distance of 1.
  • This hybrid positioning method of both absolute positioning and incremental accumulation can achieve the purpose of intermittent operation of the measurement circuit and no limit on the sensor range.
  • the transmitting winding is connected to both ends of the power supply via a drive switch (T 1 or T 2 ).
  • the driving pulse output by the driving and sampling pulse forming circuit (4) has a short duration of excitation (on the order of 10 ns) and a rising speed in the transmitting winding.
  • the eddy currents in the two rows of excitation coils respectively generate linear time-varying magnetic fields in the overlapping three-phase receiving winding regions with respective pitches as the period and periodically changing along the measuring path, thereby inducing in the two overlapping three-phase receiving windings
  • a three-phase electromotive force that does not change with time (when the sensor is not moving), but uses the pitch of each three-phase receiving winding as the period to change periodically with the measured position (relative position of the transceiver board and the excitation board); use a three-stage series analog switch
  • the differential amplifier (7) performs subtraction to synthesize the three-phase voltages of each three-phase receiving winding surrounded by the transmitting winding, which does not change with time, into a discrete Time sine signal, the space phase of the measured position in the two different pitches surrounded by the transmitting winding is converted into the initial phase of the two discrete time sine signals; the discrete time sine signal is passed through
  • Fig. 1 is a schematic diagram of the sensor structure when using a 2-pitch structure to measure linear displacement according to the present invention
  • 2A is an exploded view of the connection of the centralized receiving winding for each phase when the present invention is used for linear displacement measurement;
  • 2B is a schematic diagram of the connection of the centralized receiving winding of each phase when the present invention is used for linear displacement measurement;
  • 2C is a schematic diagram of the connection of the three-phase centralized receiving winding when the present invention is used for linear displacement measurement;
  • 3A is a schematic diagram of the connection of the distributed receiving winding realized in an overlapping manner when the present invention is used for linear displacement measurement;
  • 3B is a schematic diagram of the connection of the distributed receiving winding realized in a tiled manner when the present invention is used for linear displacement measurement;
  • 3C is a schematic diagram of the connection of the distributed receiving winding realized in a hybrid manner when the present invention is used for linear displacement measurement;
  • Fig. 4 is a schematic diagram of the sensor structure when measuring angular displacement using a 2-pitch structure according to the present invention.
  • FIG. 5 is a schematic diagram of the connection of the distributed receiving winding realized in a hybrid manner when the present invention is used for angular displacement measurement;
  • Fig. 6 is a schematic diagram of the measuring circuit when the 2-pitch structure is used in the present invention.
  • Figure 7 is a signal waveform diagram of the measuring circuit of the present invention.
  • Fig. 8 is a schematic diagram of the driving and sampling pulse forming circuit of the present invention.
  • Fig. 9 is an electrical schematic diagram of the line voltage scanning control signal generator of the present invention.
  • Fig. 10 is a flowchart of timer interrupt processing when using a 2-pitch structure according to the present invention.
  • Fig. 12 is a flowchart of the absolute positioning algorithm capture interrupt processing flow chart when the 2-pitch structure is used in the present invention.
  • FIG. 13 is a schematic diagram of the sensor structure when using a 3-pitch structure to measure linear displacement according to the present invention.
  • Fig. 14 is a schematic diagram of the measuring circuit when a 3-pitch structure is used in the present invention.
  • Fig. 15 is a flowchart of timer interrupt processing when the 3-pitch structure is used in the present invention.
  • Fig. 16 is a flowchart of the capture interrupt processing when the 3-pitch structure is used in the present invention.
  • the hybrid positioning electromagnetic induction type displacement sensor of the present invention includes a transceiver board 1 and an excitation board 2 that can move relatively along the measurement path; refer to Figure 1, which is a schematic structural diagram of the present invention when a 2-pitch structure is used to measure linear displacement.
  • the transceiver board 1 is arranged with a first pitch three-phase receiving winding 1.1, a second pitch three-phase receiving winding 1.2 and a transmitting winding 1.3.
  • the transmitting winding 1.3 uses two approximately closed rectangular coils 1.3.1 and 1.3.2 to surround the first-pitch three-phase receiving winding 1.1 and the second-pitch three-phase receiving winding 1.2 in series in the same direction. Therefore, for the transmitting One drive of winding 1.3 can induce three-phase electromotive force containing position information in the two three-phase receiving windings 1.1 and 1.2 surrounded by it at the same time, so as to measure the spatial phase of the measured position in these two different pitches in parallel Or displacement.
  • the inductance of the transmitting winding 1.3 is doubled in series in the same direction, and the rising rate of the driving current is doubled, and the power consumption remains unchanged. Under the circumstance, the duration of the driving current can be prolonged by more than 50%, which is beneficial to the stability and processing of the sensing signal.
  • the first pitch three-phase receiving winding 1.1 contains three A-phase windings 1.1.1 and B-phase windings 1.1.3 with the same structure, which are shifted by 1/6 pitch in the order of A-C-B, and have opposite polarities.
  • C-phase winding 1.1.2 the pitch of each phase winding is P 1 , and the phases differ by 120° in the order of A-B-C; because each pitch corresponds to 360°, the three-phase receiving winding can also be
  • the sequence space of A-B-C is sequentially shifted by 1/3 pitch.
  • Figure 2A is an exploded view of the connection of the receiving windings of each phase.
  • the windings are connected by coils, and the coils are composed of two mirror-symmetric upper and lower sides.
  • the dotted line in the figure indicates that each coil is located on one side of the bottom layer of the PCB, and the solid line indicates that each coil is located on the other side of the layer above the bottom layer of the PCB, which is later called the upper side.
  • the thickness between the two layers is very thin. , Only about 0.1mm; the entire coil spans 1 pitch, but the middle parallel line segments of the two sides are only half a pitch apart, and the different layers of the PCB are connected by vias (Via, the black dots in the figure).
  • the three-phase winding is divided into 60° phase band (that is, the space occupied by each phase winding under each half pitch corresponds to the 60° angle, that is, the three-phase winding is divided into each Half pitch) divide the space position.
  • the phase bands are arranged in the order of A, Z, B, X, C, and Y, where X, Y, and Z are the negative phases of A, B, and C respectively.
  • Belt, that is, the phase difference is half a pitch or 180°; all upper-layer parallel line segments belonging to the same phase are connected in series to form a group, and all upper-layer parallel line segments belonging to the same phase are located in the positive phase zone.
  • the coils are connected in series to form another group; the two sets of coils are connected head-to-head and tail-to-tail to form a closed loop, and two terminals are cut from any position to obtain the phase winding.
  • a slight adjustment of the head and tail ends can obtain a more consistent structure and a higher symmetry winding layout, as shown in FIG. 2B, and the three-phase receiving winding formed thereby is shown in FIG. 2C.
  • the waveform of the three-phase receiving winding arranged in this way does not include even harmonics in the induced electromotive force along the measurement axis, and the line voltage does not include 3, 6, 9 and other multiples of harmonics; the harmonic amplitude increases with its order It decreases monotonously, so the main harmonic components in the three-phase receiving winding line voltage are the 5th and 7th harmonics.
  • At least 2 M concentrated windings shown in Figure 2B can be sequentially shifted in space by 60°/M, and then connected in series to form a distributed winding; the fundamental wave and sub-harmonics contained in the induced electromotive force of these M sub-windings
  • the sub-winding electromotive force amplitude) and the distribution factor of each harmonic are different. It is not difficult to deduce its suppression ratio R v to the v-th harmonic:
  • the spatial phase shift is 60°/M.
  • the pitch P 1 When the pitch P 1 is large, it can be realized in an overlapping manner by shifting P 1 /(6M), as shown in Figure 3A; when the pitch P 1 is small, the overlapping method cannot Wiring can only be realized in a tiled manner by shifting L ⁇ P 1 +P 1 /(6M), where: L and M are integers, M ⁇ 2, and L ⁇ P 1 is greater than the length of the sub-winding, as shown in Figure 3B. It is also possible to use M distributed windings with the same structure as sub-windings to sequentially shift phases by 60°/M in space, and then connect them in series to form a mixed distributed winding, thereby obtaining a multiplied harmonic suppression ratio, as shown in Fig. 3C.
  • the second-pitch three-phase receiving winding 1.2 also includes three A-phase windings 1.2.1, B-phase windings 1.2.3, and polarities with the same structure, which are sequentially shifted by 1/6 pitch in the order of A-C-B.
  • the opposite C-phase winding 1.2.2, the pitch of each phase winding is P 2 , and the phases differ by 120° in the order of A—B—C.
  • the winding connection method is exactly the same as the first pitch three-phase receiving winding 1.1 .
  • the transceiver board 1 should also contain the electronic circuits needed to complete the measurement, which are used to drive the transmitter winding 1.3, process the sensing signals in the receiver windings 1.1 and 1.2, and display the measurement results. Therefore, the transceiver board 1 is generally made on a 4-layer PCB board. on.
  • the excitation board 2 is arranged with two rows of excitation coils 2.1 and 2.2 (for clarity, the excitation coil located under the receiving winding has been omitted in the figure), the excitation coil sequence 2.1 and the first pitch on the transceiver board 1 are arranged
  • the three-phase receiving winding 1.1 not only has the same pitch (P 1 ), but also has the same center line; similarly, the excitation coil sequence 2.2 has the same pitch (P 2 ) and the center line coincides with the second pitch three-phase receiving winding 1.2.
  • the shapes of the two rows of excitation coils are rounded rectangular short-circuit rings to weaken the influence of the adjacent magnetic field.
  • the width of the sides of the excitation coils along the measuring axis is equal to half of the respective pitches.
  • the excitation board 2 has a simple structure and can be fabricated on a single-sided PCB board.
  • the transmitting winding is driven
  • the rectangular coil can be approximately infinitely long, and the magnetic field generated by the current in each receiving winding area is approximately a two-dimensional magnetic field that does not change along the measurement axis, so it is directly induced in each receiving winding.
  • the electromotive force is zero.
  • the sensor structure is shown in FIG. 4. It is composed of two parts that can rotate relative to the axis A: the transceiver board 1 and the excitation board 2.
  • the transceiver board 1 is arranged with a first-pitch three-phase receiving winding 1.1, a second-pitch three-phase receiving winding 1.2, and a transmitting winding 1.3 that are spread along concentric arcs.
  • the transmitting winding 1.3 uses two approximately closed concentric arc-shaped coils 1.3.1 and 1.3.2 to respectively surround the first pitch three-phase receiving winding 1.1 and the second pitch three-phase receiving winding 1.2 in series in the same direction.
  • the three-phase receiving windings 1.1 and 1.2 both use distributed windings, and their pitches are P 1 and P 2 respectively , and both are composed of three phase windings with a phase difference of 120° in sequence.
  • the connection diagram of the first-pitch distributed three-phase receiving winding realized in a hybrid manner is shown in FIG. 5.
  • Arranged on the excitation board 2 are two rows of excitation coils 2.1 and 2.2 (for clarity, the excitation coil under the receiving winding has been omitted in the figure), which are connected to the two three-phase receiving windings on the transceiver board 1.
  • the pitches of 1.1 and 1.2 are equal, and the center lines coincide; the shapes of the two rows of excitation coils are all short-circuit rings surrounded by two concentric circular arcs and two radial straight lines, and the angle spanned along the measuring arc is equal to the respective pitch half. Except for the distribution based on concentric arcs (circle) and the pitch calculation based on the central angle, the other conditions are exactly the same as those for linear displacement measurement, and will not be repeated here.
  • the eddy current generates a linear time-varying magnetic field in the overlapping three-phase receiving winding area with its pitch as a period and periodically changing along the measuring path, so that the induction in the overlapping three-phase receiving winding does not change with time (when the sensor is not moving) , But take the three-phase receiving winding pitch as a three-phase electromotive force that periodically changes with the measured position (the relative position of the transceiver board 1 and the exciter board 2). After the distributed winding is adopted, the spatial harmonic components in the three-phase receiving winding line voltage have been greatly attenuated. Therefore, when pushing the relationship between the conductor and the phase voltage, it can be assumed that the phase voltage of the three-phase receiving winding also contains only the fundamental wave component. For the coordinate origin of, the phase voltage of the three-phase receiving winding can be set as follows:
  • E m is the amplitude of the electromotive force of each phase
  • the pitch P of the three-phase windings for receiving x is the measured displacement
  • A, B, C are sequentially lag phase voltage 2 ⁇ / 3 radians (120 °).
  • I the line voltage amplitude
  • the above formula has converted the spatial phase 2 ⁇ x/P of the measured position within the pitch P into the initial phase of the continuous-time sinusoidal signal u(t) (including the fixed offset ⁇ radians).
  • the sampled continuous-time sinusoidal signal u(t) can be restored, and the time difference between its zero-crossing point and the phase zero point can be obtained by measuring the time difference between the zero-crossing point and the phase zero point.
  • the spatial phase or displacement within the pitch By filtering out the harmonics of the discrete-time sinusoidal signal u s (n), the sampled continuous-time sinusoidal signal u(t) can be restored, and the time difference between its zero-crossing point and the phase zero point can be obtained by measuring the time difference between the zero-crossing point and the phase zero point.
  • the spatial phase or displacement within the pitch is the spatial phase or displacement within the pitch.
  • the specific circuit to realize the above assumption is shown in Figure 6.
  • the whole circuit is powered by a 3V lithium battery and can be divided into three units: central control, interface and measurement according to their functions.
  • the central control unit only includes a low-power microcontroller 13.
  • the interface unit includes key input, liquid crystal (LCD) drive circuit, measurement interface circuit and power conversion circuit. Key input and liquid crystal drive circuit are used to interact with the user; the measurement interface is used to respond to and process measurement events; the power conversion circuit is responsible for supplying power to the microcontroller 13 and generating half of the power supply voltage V CC /2 as the analog signal ground AGND .
  • LCD liquid crystal
  • the remaining circuit is the measurement unit, which includes an oscillator, a frequency dividing circuit 3, a signal generator composed of a driving and sampling pulse forming circuit 4 and a line voltage scanning control signal generator 5, and an analog switch group 6 (S 1 — S 10 ), sample and hold capacitors C 1 and C 2 , differential amplifier 7, low-pass filter 8, zero-crossing detector 9 composed of analog signal processing circuit, by a synchronous delay circuit (by D-type flip-flop FF 21 , FF 22 , FF 23 ), addition counter 10, random access memories 11 and 12, synchronous capture circuit (composed of D-type flip-flops FF 31 , FF 32 , AND gate AG 31 ) composed of phase quantization circuit and transmit winding drive power tube T 1 , the analog signal processing circuit, the random access memory and the synchronous capture circuit each have two parallel processing channels; the oscillator generates a system clock with a frequency of f M , which is a frequency divider circuit 3 and a switched capacitor low-pass filter 8 , 12K carry addition counter 10, D
  • the analog switch group 6, the differential amplifier 7, the low-pass filter 8, the zero-crossing detector 9, and the synchronous capture circuit are connected in sequence, and the sample and hold Capacitors C 1 and C 2 are respectively connected between the two input terminals of the differential amplifier 7 and the analog signal ground;
  • the synchronous delay circuit is respectively connected to the synchronous capture circuit and the 12K carry addition counter 10, the 12K carry addition counter 10 and the synchronous capture circuit and At the same time, it is connected to the random access memory 11 and 12; in order to reduce power consumption, the measurement unit can be enabled (Enable) and disabled (Disable), and the method of disabling can be simply disconnecting the power supply or stopping the system clock and turning off ( Shutdown) analog circuits, etc.
  • the three-phase receiving windings 1.1 and 1.2 on the transceiver board 1 are all connected in star (Y), and the two neutral points are connected to the analog signal ground AGND, and the transmitting winding 1.3 is connected to the power supply via the drive switch (NMOS power tube T 1) Both ends.
  • the microcontroller 13 For each measurement, the microcontroller 13 first enables the measurement unit, clears the initialization signal INI and then sets it to make the measurement unit start to work; after the measurement is completed, the microcontroller 13 disables the measurement unit to make it work Stop the operation and reduce the power consumption of the circuit by this intermittent operation.
  • the inductance of the transmitting winding 1.3 is very small (less than 1 ⁇ H). After the power supply voltage V CC is applied to it, the current in the winding rises sharply (the rising rate is on the order of 10mA/ns), so the conduction time of the power tube T 1 Should be precisely controlled, taking into account the switching characteristics of the power tube and the transient process of the sampling circuit, the pulse width of the driving signal TG should be about 30ns.
  • the driving and sampling pulse forming circuit 4 generates the driving switch of the transmitting winding 1.3-the gate driving signal TG of the NMOS power tube T 1 and the analog switches S 9 and S 10 (and the second pitch three-phase receiving winding 1.2 processing channel 2 corresponding switches) sample and save the control signal SAH.
  • the input capacitance of the power tube T 1 is relatively large (on the order of 100 pF), so the driving signal TG needs to be output through the buffer to enhance its driving capability.
  • the resistance-capacitance delay is difficult to achieve such a short and precise narrow pulse without fine-tuning. In addition, it is obviously affected by factors such as power supply voltage and temperature.
  • the best solution is to use inverter transmission delay to form batch dispersion.
  • the NAND gate G NA input terminal 2 changes to high level, the input terminal 1 remains high level (due to the transmission delay), the output Y of G NA jumps Low level; suppose the transmission delay of each stage of inverter is t pd (on the order of ns), and the number of the odd number of inverters used is represented by (2N+1), then (2N+1)t pd
  • the input terminal 1 of the NAND gate G NA transitions to low level (due to odd number of inversions), and the output Y transitions to high level and remains until the rising edge of S CLK arrives again (due to S CLK G NA still outputs high level when it becomes low level).
  • each rising edge of the input signal S CLK triggers the NAND gate G NA to output a narrow pulse of negative polarity with a pulse width of (2N+1)t pd .
  • the output Y of the NAND gate G NA is inverted to obtain a positive sample and hold control signal SAH, and the output Y of G NA is inverted and buffered to obtain the gate drive signal TG of the power tube T 1.
  • Each drive of the transmitting winding 1.3 induces three-phase electromotive force in the three-phase receiving windings 1.1 and 1.2 at the same time, so the output signals of the two three-phase receiving windings can be processed in parallel.
  • the circuits used are exactly the same, so the following discussion is only for the first The one-phase three-phase receiving winding 1.1 is expanded.
  • the analog switch group 6 (S 1 -S 10 ), the sample and hold capacitors C 1 and C 2 , and the differential amplifier 7 jointly complete the sampling and amplification of the line voltage of the first-pitch three-phase receiving winding 1.1 to synthesize the expression ( d) The discrete-time sinusoidal signal u s (n) described.
  • the control signal Q 0 is responsible for turning on the analog switches S 1 and S 4 to generate u 1A- u 1B (or its inverse u 1B- u 1A ), and the control signal Q 1 is responsible for turning on the analog switches S 2 and S 5 to generate u 1A- u 1C (or its inverse u 1C -u 1A ), the control signal Q 2 is responsible for turning on the analog switches S 3 and S 6 to generate u 1B -u 1C (or its inverse u 1C -u 1B ), and the control signal Q 3 Responsible for the channel switching of single-pole double-throw analog switches S 7 and S 8 to determine whether to reverse phase, the control signal SAH is responsible for turning on the analog switches S 9 and S 10 to sample the two input phase voltages and save the results to the sample and hold capacitors.
  • the differential amplifier 7 performs subtraction and amplification to obtain a sampling and zero-order hold signal of the line voltage. It can be seen that the 10 analog switches of the analog switch group 6 are formed into a three-stage switch series structure to sequentially complete the selection, exchange, and sampling of the phase voltages, which greatly simplifies the circuit without decoding the control signal.
  • each cycle of Q 0 , Q 1 , and Q 2 is inverted by Q 3 to obtain the press A-B, A-C, B-C , B-A, C-A, C-B sequence line voltage cyclic scanning sampling, synthesis of discrete-time sinusoidal signal u s (n) described in expression (d).
  • the realization circuit of the line voltage scanning control signal generator 5 that generates the above Q 0 , Q 1 , Q 2 and Q 3 signals is shown in FIG. 9, and each signal waveform is shown in FIG. 7.
  • D-type flip-flops FF 11 , FF 12 , and FF 13 constitute a circular counter of cyclic shift.
  • the initialization signal INI presets it to 1, 0, and 0, so there is only one high level at any time; D-type flip-flop FF 14 counts the rising edge of Q 0 , so its output Q 3 flips every cycle of Q 0 , Q 1 , and Q 2 ; the clock signal P CLK comes from the frequency divider circuit 3, which is the trigger clock S CLK for 2 Frequency division output; the driving and sampling pulse forming circuit 4 outputs a driving and a sampling pulse at each rising edge of the trigger clock S CLK , so each line voltage is continuously sampled twice to increase the discrete time sine of the relative movement of the sensor The number of samples of the signal u s (n).
  • Patent CN101949682B discloses an absolute position capacitive displacement sensor, which includes low-pass filtering of discrete-time sinusoidal signals and zero-crossing detection using an addition counter to measure the displacement of the measured position within each wavelength and the absolute positioning of the measured displacement Method, introduce its related content here.
  • the discrete-time sinusoidal signal u s (n) synthesized by the differential amplifier 7 is filtered by the switched capacitor (for ease of integration) and the low-pass filter 8 to restore the continuous-time sinusoidal signal u r (t) described in the expression (e).
  • the zero-crossing detector 9 shapes the continuous-time sinusoidal signal u r (t) square wave signal U Z, which corresponds to the rising edge of u r (t) from the negative to positive zero crossing, a corresponding falling u r (t) from positive to negative zero crossing, both of phase 180 °, the sensor is not
  • a synchronous delay circuit composed of D-type flip -flops FF 21 , FF 22 and FF 23 and a D-type flip-flop are set
  • the synchronous capture circuit composed of FF 31 , FF 32 and AND gate AG 31 , see Figure 7 for its signal waveform.
  • Synchronous delay circuit, synchronous capture circuit, 12K carry (because each pitch is subdivided into 12K equal parts) addition counter 10, random access memories 11 and 12 jointly complete the phase quantization task.
  • the initialization signal INI presets the D-type flip-flops FF 21 , FF 22 , and FF 23 to 1, 1, 0, and the low level C E output by FF 23 asynchronously clears the 12K carry addition counter 10 and prevents it from counting;
  • D type The flip-flops FF 21 and FF 22 constitute a 2-bit asynchronous subtraction counter, which counts down the rising edge of the output signal Q 3 from the line voltage scanning control signal generator 5; when the fourth rising edge of Q 3 arrives, the D type
  • the output signal W 1 of the flip-flop FF 22 generates a positive transition to set the output signal C E of the FF 23 to 1:
  • the 12K carry addition counter 10 starts counting from 0, and the synchronous capture circuit releases the capture lock; therefore, the synchronous delay circuit generates Delay of 4 Q 3 cycles and set the phase zero point on the rising edge of Q 3.
  • the initialization signal INI clears the D-type flip-flops FF 31 and FF 32 to 0 asynchronously.
  • the low level C E output by the synchronous delay circuit makes the D-type flip-flop FF 31 always output low level C.
  • S , AND gate AG 31 always output low level C P
  • D-type flip-flop FF 32 always output low level [P 1 rising edge capture] and high level [P 1 falling edge capture] signals, thereby blocking Capture the count value of the 12K carry addition counter 10; after the delay time is reached, the output signal C E of the synchronous delay circuit is set to high level, and the square wave signal U Z will be D-shaped on the first rising edge after this.
  • the output signal C S of the flip-flop FF 31 is also set to high level, and the output signal C P of the AND gate AG 31 is the same as the square wave signal U Z from then on ; the 12K carry addition counter 10 does not generate a count down of the system clock Edge, the D-type flip-flop FF 32 synchronizes the output signal C P of the AND gate AG 31 and outputs the same phase [P 1 rising edge capture] and reverse phase [P 1 falling edge capture] signals, respectively in the square wave
  • the rising and falling edges of the signal U Z capture the count value of the 12K carry addition counter 10 in a synchronous manner, and save them in the random access memories 11 and 12 respectively, and at the same time notify the microcontroller 13 of the occurrence of the capture event, the microcontroller 13 Then read in the captured value and judge whether the measurement is complete.
  • the microcontroller 13 disables the measurement unit to stop working, and then calculates the measured displacement through software.
  • the linear displacement resolution is 10 ⁇ m
  • P 1 3.84 mm
  • P 2 3.6 mm
  • ⁇ is the proportional coefficient
  • mod is the modulo operation
  • the difference m ⁇ ( ⁇ N 2 - ⁇ N 1) for adding or subtracting the pitch P M (unit conversion is required, the same below) correction which is the main used both to capture the rising edge, falling edge and a capture.
  • K M is the number of medium pitches
  • K F is the number of fine pitches, both of which are integers.
  • the number of integers K M of the middle pitch included in the total displacement x T the number of integers K F of the fine pitch included in the middle displacement x M , and the measured displacement x with a step of 1:
  • This hybrid positioning algorithm needs to continuously accumulate the middle displacement increment, and the sensor displacement between two measurements cannot exceed 1 middle pitch range, so there is a minimum measurement frequency requirement.
  • the sensor using the 2-pitch structure is applied to short-range measuring instruments such as indicators and small calipers, only the middle displacement x M and the fine displacement x F can be used to determine the measured displacement x in the middle pitch range without the need to accumulate the middle displacement increase.
  • the measuring range of ⁇ x M is extended to become an inductive displacement sensor for absolute positioning. Compared with the existing absolute position induction displacement sensor, it has the following advantages:
  • the transmitting winding surrounds two three-phase receiving windings with different pitches at the same time, and two sets of sensing signals with different pitches can be obtained at the same time by one drive, so that the displacement measurement of the measured position in two different pitches can be completed in parallel .
  • This not only saves the driving power consumption of a set of driving circuits and the primary transmitting winding, but also enhances the ability to track the rapid movement of the sensor;
  • the distributed receiving winding has a strong ability to suppress spatial harmonics, so there is no need to use a sinusoidal winding with a large area and complex shape, which increases the wiring density of the receiving winding, increases the number of coils, and enhances the received signal;
  • the excitation coil adopts a simple short-circuit loop shape, which is shorter than the interconnected multi-coupling loop structure, the loop resistance is smaller, the induced eddy current is larger, and the received signal is stronger; compared to the interference of the rectangular copper foil structure adjacent to the magnetic field Smaller and higher precision;
  • the drive pulse formed by the transmission delay excites a linear time-varying current with a short duration in the transmitting winding, and induces an electromotive force that does not change with time (when the sensor is not moving) in each receiving winding, which greatly reduces the timing requirements for the sampling circuit;
  • the sensing signal is synthesized into a discrete-time sinusoidal signal through a three-stage series-connected analog switch and a differential amplifier, the space where the measured position is within two different pitches is completed through simple circuits such as low-pass filtering, zero-crossing detection, and addition counting. Phase or displacement measurement, so the measurement circuit is easy to implement and integrate.
  • the middle displacement x M m ⁇ (N 2 -N 1 ) needs to be mapped to the semi-closed interval [0, P M ) to correspond to the fine displacement x F ⁇ [0 , P F) match, i.e., when x M is negative when it is combined with the pitch P M becomes positive so, the details contained in the displacement x M is then determined by the relation of formula (j) from the integer number K F , And finally calculate the measured displacement x as follows:
  • the present invention does not need to make any changes to the structure of the sensor and the measurement circuit, and only through software algorithms can achieve two measurement methods of absolute positioning and hybrid positioning.
  • absolute positioning has low power consumption but low range Short (limited to mid-pitch range), hybrid positioning has no range limit but has a minimum measurement frequency requirement.
  • the microcontroller 13 adopts an interrupt-driven mode: that is, it enters sleep immediately every time the interrupt returns, until a new interrupt wakes it up again.
  • Two interrupt vectors are required to complete the measurement: timer interrupt and capture interrupt.
  • Timer interrupt is used to initiate measurement
  • capture interrupt is used to respond to capture events: [P 1 rising edge capture], [P 1 falling edge capture], [P 2 rising edge capture] and [P 2 falling edge capture] four external Interrupt request.
  • the timer generates an interrupt every 20ms, initiates a measurement with a maximum frequency of 50 times/second, and halves the measurement frequency when the sensor is stationary or in standby; in order to respond to the capture event in time, it is also enabled (Enable) [P 1 rising edge capture], [ P 1 falling edge capture], [P 2 rising edge capture] and [P 2 falling edge capture] 4 external interrupt requests, the processing flow is shown in Figure 10.
  • the capture interrupt service routine processes the measurement process and data.
  • the captured interrupt request will be disabled after reading the capture value of the interrupt source; if the measurement has been completed (4 captured interrupt requests are all closed): disable the measurement unit to stop running ,
  • the flow chart is shown in Figure 11.
  • the timer generates an interrupt every 125ms, and initiates a measurement with a maximum frequency of 8 times per second.
  • the measurement frequency is also halved when the sensor is stationary or in standby. Therefore, the processing flow is exactly the same as that of hybrid positioning, as shown in Figure 10.
  • the processing flow of capturing interrupt is similar to that of hybrid positioning, except that the method of determining the measured displacement is different and the measurement result is displayed for each measurement, as shown in Figure 12.
  • Read in the capture value of the interrupt source and close the interrupt request of the capture; if the measurement has been completed: disable the measurement unit, calculate the displacement x M m ⁇ (N 2 -N 1 ), determine whether the sensor is moving and set the state Mark, map the middle displacement x M to the semi-closed interval [0, P M ), obtain the integer number of fine distances K F contained in the middle displacement x M from the relation (j), and calculate the measured value according to formula (l) Displacement x and display measurement results according to user requirements.
  • the present invention can also use a 3-pitch structure.
  • the sensor structure when measuring linear displacement is shown in Figure 13. It is composed of two parts that can move relative to each other along the measurement axis: the transceiver board 1 and the excitation board 2.
  • the transmitting winding 1.4 uses two approximately closed rectangular coils 1.4.1 and 1.4.2 to surround the first pitch three-phase receiving winding 1.1 and the second pitch three-phase receiving winding 1.2 in series in the same direction; the transmitting winding 1.5 is used Two approximately closed rectangular coils 1.4.1 and 1.5.2 respectively surround the first pitch three-phase receiving winding 1.1 and the third pitch three-phase receiving winding 1.3 in series in the same direction.
  • the three-phase receiving windings 1.1, 1.2, and 1.3 all use distributed windings with pitches P 1 , P 2 and P 3 , respectively, and they are all composed of three phase windings with a difference of 120° in sequence.
  • excitation coils 2.1, 2.2 and 2.3 There are three rows of excitation coils 2.1, 2.2 and 2.3 arranged along the measuring axis on the excitation plate 2 (for clarity, the excitation coils under the receiving winding have been omitted in the figure), which are respectively connected with the three three-phases on the transceiver board 1.
  • the receiving windings 1.1, 1.2 and 1.3 have the same pitch and the center lines coincide; the shapes of the 3 rows of excitation coils are all rounded rectangular short-circuit rings, and the width of the sides along the measuring axis is equal to half of the respective pitches.
  • the transmitting winding 1.4 and the first-pitch three-phase receiving winding 1.1 and the second-pitch three-phase receiving winding 1.2 surrounded by it constitute a part of the transceiver board using the 2-pitch structure that has been described in detail, and is used to measure the middle pitch
  • the internal displacement; the transmitting winding 1.5 and the first pitch three-phase receiving winding 1.1 and the third pitch three-phase receiving winding 1.3 surrounded by it constitute another part of the transceiver board that uses a 2-pitch structure to measure the ratio of the middle section Displacement within a coarse pitch with a larger step pitch and a larger range; two structures using two pitches share the rectangular coil 1.4.1 and the three-phase receiving winding 1.1 surrounded by it, so a three-pitch structure is equivalent to two two-pitch structures.
  • the transceiver board 1 and the excitation board 2 are both extended from the linear displacement sensor using the 2-pitch structure, and the design method is exactly the same, so it will not be repeated here.
  • phase quantization codes N 11 and N 12 in the two different pitches P 1 and P 2 in the first 2-pitch structure can be measured in parallel step by step and the measured position is measured in parallel in the second
  • the phase quantization codes N 21 and N 23 in two different pitches P 1 and P 3 in a two-pitch structure are similar to the derivation process of expression (h).
  • the specific circuit for realizing the above step-by-step measurement is shown in Figure 14. It only adds one NMOS power tube T 2 and four multiple analog switches S 11 -S 14 ( Including control signal [pitch selection]), so the working principle is exactly the same.
  • the microcontroller 13 After the microcontroller 13 is enabled measurement unit, outputs a low level signal of the [selection] pitch: connecting the output terminal of the drive power to the driving signal TG tube T so that a transmission gate winding 1.4 (by the coil 1.4.1 And 1.4.2) Accept the drive, connect the output terminal of the second-pitch three-phase receiving winding 1.2 to the second processing channel, then clear the initialization signal INI and set it to initiate the first measurement, and the parallel measurement is Measure the phase quantization codes N 11 and N 12 in the first pitch P 1 and the second pitch P 2 ; then the microcontroller 13 outputs a high-level [pitch selection] signal: output the drive signal TG The terminal is changed to the gate of the driving power tube T 2 to make the transmitting winding
  • the microcontroller 13 also adopts an interrupt drive mode, and two interrupt vectors are required to complete the measurement: timer interrupt and capture interrupt.
  • the timer generates an interrupt every 125ms, and initiates a measurement with a maximum frequency of 8 times per second.
  • the measurement frequency is halved when the sensor is stationary or in standby.
  • the processing flow is one more step "clear" [ Pitch selection] signal", as shown in Figure 15.
  • the processing flow of catching interrupt is shown as in Fig. 16.
  • Read in the capture value of the interrupt source close the interrupt request of the capture; if the first measurement is completed: set the [pitch selection] signal, clear the initialization signal INI, clear each capture interrupt request flag, and reopen each
  • a 3-pitch structure can also be used to measure the angular displacement, but because the angular displacement only needs to be positioned absolutely within 360°, the 2-pitch structure is generally used to meet the requirements, so it will not be explained here.

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Abstract

一种电磁感应式位移传感器，由可沿测量路径相对运动的收发板（1）和励磁板（2）组成。收发板（1）上布置有至少一个发射绕组（1.3）和比发射绕组数量多一个的至少两个节距不同的三相接收绕组（1.1，1.2）。每个发射绕组（1.3）均以同向串联的方式同时包围两个节距不同的三相接收绕组（1.1，1.2），所有接收绕组（1.1，1.2）均使用分布式绕组结构。励磁板（2）上布置有与收发板（1）上三相接收绕组（1.1，1.2）数量相等的至少两列短路环形状的励磁线圈（2.1，2.2），分别与对应的三相接收绕组（1.1，1.2）对齐和节距相等。利用传输延时形成的驱动脉冲在发射绕组（1.3）激发持续时间很短的线性时变电流，同时在被其包围的两个三相接收绕组（1.1，1.2）中感应不随时间变化的电动势，对此三相电动势合成、滤波、整形后利用所得方波的边沿捕捉加法计数器（10）的计数值并行测得被测位置在两个不同节距内的空间相位或位移，最后通过混合定位或绝对定位算法求出被测位移。

Description

混合定位电磁感应式位移传感器 技术领域
本发明涉及电磁感应式位移测量技术，具体是采用混合定位方法实现的电磁感应式位移传感器。
背景技术
为克服电容式位移传感器对脏污、潮湿等因素敏感的缺陷，电磁感应式位移传感器应运而生。
专利US5804963公开了一种增量式电磁感应式位移传感器，它通过循环驱动3组施感绕组、接收与正被驱动的施感绕组等效正交的两组感应绕组中的信号完成位移测量。专利CN100491896C对US5804963中的施感绕组和感应绕组的结构做了改进，使施感导体(绕组)和感应导体(绕组)不再共用而分开设置以减小两者之间的直接耦合，测量方法没有改变。增量式测量实现简单，但必须连续不间断地工作以实时累加位移增量，因而工作电流较大。
专利US5886519公开了一种绝对位置电磁感应式位移传感器，它通过依次测量被测位置在多个不同波长中的空间相位推导测量结果。为获得被测位置在某个波长中的空间相位：首先以LC振荡方式激励该波长的发射绕组、待振荡电流过零时激活该波长接收绕组信号的采样和保持(SAH)控制信号、将采样信号进行放大和模数转换(ADC)、最后经反正切运算求出所需相位。专利CN1198111C对US5886519中的标尺结构和各发射绕组的布局做了改进，新的标尺使用多个互连的耦合回路以便在接收绕组区域产生正、负对称的空间磁场，测量方法没有改变。该绝对位置电磁感应式位移传感器没有最低测量频率要求，因而可长间隔地间歇工作，平均工作电流很小；不足之处是它有限的量程制约了它的应用范围，例如：常规卡尺的槽宽仅约12.5mm，至多容纳2个改进结构的波长布局，以目前的制造工艺，无法将其应用到200mm以上的常规卡尺之中。此外，对纽扣电池供电的手持量具而言，要实现其测量方法中的高速比较器、模数转换器、反正切运算等功能部件并非易事，因而制造成本不低。
发明内容
综上所述，本发明的目的是为了解决现有电磁感应式位移传感器存在的或者工作电流较大，或者量程受限、，生产成本高制造难度较大等技术不足，而公开一种既可间歇工作、 又无量程限制且易于实现的混合定位电磁感应式位移传感器。
为解决本发明所提出的技术问题，采用的技术方案为：
混合定位电磁感应式位移传感器，其特征在于所述传感器包括有可沿测量路径相对运动的收发板(1)和励磁板(2)；
所述的收发板(1)上布置有测量电路以及沿测量路径展开的至少一个发射绕组和比发射绕组数量多一个的至少两个节距不同的三相接收绕组；每个发射绕组同时包围两个节距不同的三相接收绕组，每个三相接收绕组包含3个结构相同、依次相差120°的相绕组，发射绕组和三相接收绕组均与所述测量电路相连；所述测量电路包括有中控单元、接口单元和测量单元；中控单元包含微控制器(13)；接口单元包括有与微控制器(13)连接的按键输入电路、液晶驱动电路、测量接口电路和电源变换电路；
所述的励磁板(2)上布置有沿测量路径展开的与收发板(1)上三相接收绕组数量相等的至少两列励磁线圈，各列励磁线圈分别与收发板(1)上对应的三相接收绕组节距相等、中心线重合，沿测量路径的尺寸为各自节距的一半。
作为对本发明进一步限定的技术方案包括有：
所述的测量单元包括振荡器、分频电路(3)、由驱动和采样脉冲形成电路(4)和线电压扫描控制信号发生器(5)组成的信号发生器、由模拟开关组(6)、采样保持电容(C1、C2)、差分放大器(7)、低通滤波器(8)、过零检测器(9)组成的模拟信号处理电路、由同步延时电路、加法计数器(10)、随机访问存储器(11、12)和同步捕捉电路组成的相位量化电路以及发射绕组驱动功率管(T1、T2)，模拟信号处理电路、随机访问存储器和同步捕捉电路各设两组形成两个并行处理通道；
振荡器直接或经分频电路(3)为驱动和采样脉冲形成电路(4)、线电压扫描控制信号发生器(5)、低通滤波器(8)、同步捕捉电路及加法计数器(10)提供输入时钟；驱动和采样脉冲形成电路(4)与模拟开关组(6)连接的同时还直接或经多路开关(S14)与发射绕组驱动功率管(T1、T2)相连，线电压扫描控制信号发生器(5)分别连接模拟开关组(6)和同步延时电路；模拟开关组(6)、差分放大器(7)、低通滤波器(8)、过零检测器(9)、同步捕捉电路依次顺序相连，采样保持电容(C1、C2)连接在差分放大器(7)的输入端与模拟信号地之间；同步延时电路分别连接同步捕捉电路和加法计数器(10)，加法计数器(10)和同步捕捉电路又同时与随机访问存储器(11、12)相连；各发射绕组均经各自的驱动功率管(T1或T2)和电源连接，各三相接收绕组均按星型(Y)联结且中性点和模拟信号地相连。
每个发射绕组均用两个近似闭合的线圈以同向串联的方式分别包围两个节距不同的三相接收绕组；所有励磁线圈均采用短路环的形状；每相接收绕组均由至少为2的M个结构相同但空间依次移相60°/M的子绕组串联而构成一个分布绕组，构成分布绕组的各子绕组本身还可以是分布绕组；每次都并行测量被测位置在被同一个发射绕组所包围的两个不同节距内的空间相位或位移；测得被测位置在所需不同节距内的空间相位或位移后微控制器(13)禁能测量单元使其停止运行、利用混合定位或绝对定位算法计算被测位移。
在使用2节距结构测量线性位移时，收发板(1)和励磁板(2)可沿测量轴线相对移动；收发板(1)上布置有沿测量轴线展开的一个发射绕组(1.3)和两个节距不同的三相接收绕组(1.1、1.2)；发射绕组(1.3)用两个近似闭合的矩形线圈(1.3.1、1.3.2)以同向串联的方式分别包围节距为P 1的三相接收绕组(1.1)和节距为P 2的三相接收绕组(1.2)；励磁板(2)上布置有沿测量轴线展开的两列励磁线圈(2.1、2.2)，分别与收发板(1)上的两个三相接收绕组(1.1、1.2)节距相同，中心线重合，励磁线圈的形状采用圆角矩形短路环；
将节距P 2和P 1的空间频率差作为中节距空间频率F M＝1/P 2-1/P 1，得到波长比为m的中节距P M＝1/F M＝P 1·P 2/(P 1-P 2)＝m·P 1＝(m+1)·P 2。
在使用2节距结构测量角位移时，收发板(1)和励磁板(2)可围绕转轴相对转动，节距按角度计算；收发板(1)上布置有沿同心圆弧展开的一个发射绕组(1.3)和两个节距不同的三相接收绕组(1.1、1.2)；发射绕组(1.3)用两个近似闭合的同心圆弧形线圈(1.3.1、1.3.2)以同向串联的方式分别包围节距为P 1的三相接收绕组(1.1)和节距为P 2的三相接收绕组(1.2)；励磁板(2)上布置有沿同心圆周展开的两列励磁线圈(2.1、2.2)，分别与收发板(1)上的两个三相接收绕组(1.1、1.2)节距相同，中心线重合，励磁线圈的形状采用由两条同心圆弧和两条径向直线围成的短路环；
将节距P 2和P 1的空间频率差作为中节距空间频率F M＝1/P 2-1/P 1，得到波长比为m的中节距P M＝1/F M＝P 1·P 2/(P 1-P 2)＝m·P 1＝(m+1)·P 2。
在使用3节距结构测量线性位移时，收发板(1)和励磁板(2)可沿测量轴线相对移动；收发板(1)上布置有沿测量轴线展开的两个发射绕组(1.4、1.5)和三个节距不同的三相接收绕组(1.1、1.2、1.3)，第一个发射绕组(1.4)用两个近似闭合的矩形线圈(1.4.1、1.4.2)以同向串联的方式分别包围节距为P 1的三相接收绕组(1.1)和节距为P 2的三相接收绕组(1.2)，第二个发射绕组(1.5)用两个近似闭合的矩形线圈(1.4.1、1.5.2)以同向串联的方式分别包围节距为P 1的三相接收绕组(1.1)和节距为P 3的三相 接收绕组(1.3)；励磁板(2)上布置有沿测量轴线展开的三列励磁线圈(2.1、2.2、2.3)，分别与收发板(1)上的三个三相接收绕组(1.1、1.2、1.3)节距相同，中心线重合，励磁线圈的形状采用圆角矩形短路环。
将节距P 2和P 1的空间频率差作为中节距空间频率F M＝1/P 2-1/P 1，得到波长比为m的中节距P M＝1/F M＝P 1·P 2/(P 1-P 2)＝m·P 1＝(m+1)·P 2；将节距P 3和P 1的空间频率差作为粗节距空间频率F C＝1/P 3-1/P 1，得到波长比为n的粗节距P C＝1/F C＝P 1·P 3/(P 1-P 3)＝m·n·P 1＝(m·n+1)·P 3。
所述驱动和采样脉冲形成电路(4)包括有级联的奇数个反相器和一个与非门(G NA)；所述分频电路(3)输出的第一个时钟信号(S CLK)作为该电路的触发脉冲分别与第一个反相器的输入端子和与非门(G NA)的一个输入端子相连，级联的奇数个反相器的输出和与非门(G NA)的另一个输入端子相连，在触发脉冲(S CLK)的每个上升沿，与非门(G NA)输出一个负极性的窄脉冲，其宽度等于级联的奇数个反相器的总传输延时；对与非门(G NA)的输出(Y)反相得到正极性的采样和保持控制信号(SAH)，对与非门(G NA)的输出(Y)反相缓冲得到正极性的驱动信号(TG)。
所述线电压扫描控制信号发生器(5)由4个D型触发器(FF 11、FF 12、FF 13、FF 14)组成；所述分频电路(3)输出的第二个时钟信号(P CLK)同时和这4个D型触发器的时钟端子相连，所述微控制器(13)输出的初始化信号(INI)将这4个D型触发器分别预设为1、0、0、1；前3个D型触发器(FF 11、FF 12、FF 13)构成一个循环移位的环形计数器，其输出的3个信号(Q 0、Q 1、Q 2)轮流为高电平；第4个D型触发器(FF 14)对第1个D型触发器(FF 11)的输出(Q 0)进行计数，所述环形计数器的每次循环使其输出(Q 3)发生一次翻转。
所述模拟开关组(6)组成一个三级开关串联的结构，对输入的三相电压依次完成相电压的选择、交换和采样：第一级开关(S 1—S 6)被配成{S 1、S 4}、{S 2、S 5}和{S 3、S 6}的三对，分别选择相电压对{u A、u B}、{u A、u C}和{u B、u C}；第二级开关(S 7—S 8)根据需要交换第一级开关所选择的相电压对的顺序；第三级开关(S 9—S 10)对第二级开关输出的一对相电压进行采样并将结果分别保存到所述的采样保持电容(C 1、C 2)之中；
所述差分放大器(7)对保存在采样保持电容(C 1、C 2)中的两个相电压采样实施减法运算和放大得到与之对应的一个线电压的采样和零阶保持信号；在所述信号发生器的输出信号控制下，依次获得按A—B、A—C、B—C、B—A、C—A、C—B顺序的线电压循环扫描采样，将输入的所述三相接收绕组不随时间变化(传感器不动时)、但随被测位置做 周期变化的三相电压合成为一个离散时间正弦信号(u s(n))，被测位置在所述三相接收绕组节距内的空间相位被转换成该离散时间正弦信号(u s(n))的初相位。
所述同步延时电路由3个D型触发器(FF 21、FF 22、FF 23)组成；所述线电压扫描控制信号发生器(5)的最后一个输出信号(Q 3)与第一个D型触发器(FF 21)的时钟端相连作为该电路的输入时钟，所述微控制器(13)输出的初始化信号(INI)将这3个D型触发器分别预设为1、1、0，使输出信号(C E)为低电平；前2个D型触发器(FF 21、FF 22)构成一个2位异步减法计数器，对输入时钟(Q 3)的上升沿进行减计数；当第4个输入时钟(Q 3)的上升沿到达时，使输出信号(C E)跃变为高电平；
所述同步延时电路的输出信号(C E)跃变为高电平之后，所述加法计数器(10)从0开始计数、所述同步捕捉电路解除捕捉封锁；
所述同步捕捉电路由2个D型触发器(FF 31、FF 32)和一个与门(AG 31)组成；所述微控制器(13)输出的初始化信号(INI)将2个D型触发器(FF 31、FF 32)异步清0，在延时时间未到时封锁对所述加法计数器(10)计数值的捕捉；延时时间到达后，所述同步延时电路的输出信号(C E)跃变为高电平、所述模拟信号处理电路输出的方波信号(U Z)在此之后的第一个上升沿将第一个D型触发器(FF 31)的输出信号(C S)也置为高电平、与门(AG 31)的输出信号(C P)自此之后便和所述方波信号(U Z)相同；第二个D型触发器(FF 32)将与门(AG 31)的输出信号(C P)和系统时钟下降沿同步后分别在所述方波信号(U Z)的上升沿和下降沿以同步方式捕捉加法计数器(10)的计数值，将其分别保存在两个随机访问存储器(11、12)中，由此得到被测位置在与该处理通道相连的所述三相接收绕组节距内的空间相位或位移的量化编码。
在使用2节距结构时，并行测得被测位置在被发射绕组(1.3)包围的两个不同节距P 1和P 2内的相位量化编码N 1和N 2后，则细位移x F＝N 1(相应地称节距P 1为细节距P F)、中位移x M＝m·(N 2-N 1)，然后累加相邻两次测量的中位移增量Δx M得到总位移x T＝∑(Δx M)、根据关系x T≈K M·P M+x M求出总位移x T所包含的中节距整数个数K M、根据关系x M≈K F·P F+x F求出中位移x M所包含的细节距整数个数K F、然后根据公式x＝(m·K M+K F)·P F+x F得到没有量程限制的被测位移x；也可使用以下绝对定位算法：将中位移x M映射到半闭区间[0，P M)、根据关系x M≈K F·P F+x F求出中位移x M所包含的细节距整数个数K F、根据公式x＝K F·P F+x F得到在中节距范围绝对定位的被测位移x。
在使用3节距结构时，并行测得被测位置在第一个发射绕组(1.4)所包围的两个不同节距P 1和P 2内的相位量化编码N 11和N 12以及并行测得被测位置在第二个发射绕组(1.5) 所包围的两个不同节距P 1和P 3内的相位量化编码N 21和N 23之后，则细位移x F＝N 21(相应地称节距P 1为细节距P F)、中位移x M＝m·(N 12-N 11)、粗位移x C＝m·n·(N 23-N 21)，将粗位移x C映射到半闭区间[0，P C)、根据关系x C≈K M·P M+x M求出粗位移x C所包含中节距P M的整数个数K M、根据关系x M≈K F·P F+x F求出中位移x M所包含细节距P F的整数个数K F、根据公式x a＝(m·K M+K F)·P F+x F得到在粗节距范围的绝对位移x a、如不扩展量程此即被测位移x＝x a、否则通过累加相邻两次测量的绝对位移增量Δx a得到没有量程限制的被测位移x＝∑(Δx a)。
本发明的有益效果为：本发明既可用于线性位移测量，也可用于角位移测量；既可使用2节距结构，也可使用3节距结构；既可使用混合定位算法，也可使用绝对定位算法。
如前所述，对两个不同节距的空间频率进行差频运算，可得与该频差对应的更长空间周期——中节距或粗节距；被测位置在中节距和粗节距内的位移均可通过对被测位置在做差频运算的两个不同节距内的空间相位相减和放大后直接求出(绝对定位)而不依赖于测量历史或过程。但中节距尺度有限，在使用2节距结构时可通过累加相邻两次测量的中节距位移增量获得没有量程限制但步距为中节距波长比的总位移；在使用3节距结构时，粗节距的尺度一般能满足常规测量的要求，但在大量程(超出粗节距范围)应用时也可通过累加相邻两次测量的绝对位移增量得到没有量程限制且步距为1的被测位移。这种既绝对定位又增量累加的混合定位方法便可达成测量电路间歇工作、传感器量程没有限制的目的。
将发射绕组经驱动开关(T 1或T 2)连接于电源两端，所述驱动和采样脉冲形成电路(4)输出的驱动脉冲在发射绕组激发持续时间很短(10ns量级)、上升速度很快(10mA/ns量级)的线性时变驱动电流；该电流产生的线性时变磁场在与其耦合的励磁板(2)上的两列励磁线圈中感应随时间线性增长的电涡流，这两列励磁线圈中的电涡流分别在与其重叠的三相接收绕组区域产生以各自节距为周期沿测量路径做周期变化的线性时变磁场，从而在与其重叠的两个三相接收绕组中感应不随时间变化(传感器不动时)、但以各三相接收绕组节距为周期随被测位置(收发板和励磁板的相对位置)做周期变化的三相电动势；使用三级串联的模拟开关组(6)依次完成对相电压的选择、交换和采样后，由差分放大器(7)实施减法运算将被该发射绕组包围的每个三相接收绕组不随时间变化的三相电压合成为一个离散时间正弦信号，被测位置在被该发射绕组包围的两个不同节距内的空间相位被分别转换成这两个离散时间正弦信号的初相位；离散时间正弦信号经低通滤波器(8)滤波后还原出被抽样的连续时间正弦信号，过零检测器(9)将该连续时间正弦信号整形 成方波；待电路暂态过程足够衰减之后，所述同步延时电路解除对加法计数器(10)和同步捕捉电路的封锁，由整形所得的两个方波的上升沿和下降沿在经系统时钟同步后分别捕捉同一个从相位零点开始计数的加法计数器(10)的计数值从而得到被测位置在被该发射绕组包围的两个不同节距内的空间相位或位移的量化编码(可能包含固定偏移)。对不随时间变化的信号采样几乎没有时序要求，对采样信号的后续处理也只是使用一些简单电路，因此本发明对电子线路要求不高，易于集成，便于实现低成本的规模化生产。
附图说明
图1为本发明使用2节距结构测量线性位移时的传感器结构示意图；
图2A为本发明用于线性位移测量时每相集中式接收绕组的连接分解图；
图2B为本发明用于线性位移测量时每相集中式接收绕组的连接示意图；
图2C为本发明用于线性位移测量时三相集中式接收绕组的连接示意图；
图3A为本发明用于线性位移测量时以交叠方式实现的分布式接收绕组的连接示意图；
图3B为本发明用于线性位移测量时以平铺方式实现的分布式接收绕组的连接示意图；
图3C为本发明用于线性位移测量时以混合方式实现的分布式接收绕组的连接示意图；
图4为本发明使用2节距结构测量角位移时的传感器结构示意图；
图5为本发明用于角位移测量时以混合方式实现的分布式接收绕组的连接示意图；
图6为本发明使用2节距结构时的测量电路原理图；
图7为本发明测量电路的信号波形图；
图8为本发明驱动和采样脉冲形成电路的原理图；
图9为本发明线电压扫描控制信号发生器的电原理图；
图10为本发明使用2节距结构时的定时器中断处理流程图；
图11为本发明使用2节距结构时的混合定位算法捕捉中断处理流程图；
图12为本发明使用2节距结构时的绝对定位算法捕捉中断处理流程图；
图13为本发明使用3节距结构测量线性位移时的传感器结构示意图；
图14为本发明使用3节距结构时的测量电路原理图；
图15为本发明使用3节距结构时的定时器中断处理流程图。
图16为本发明使用3节距结构时的捕捉中断处理流程图。
具体实施方式
以下结合附图和本发明优选的具体实施例对本发明作进一步地说明。
本发明混合定位电磁感应式位移传感器，包括有可沿测量路径相对运动的收发板1 和励磁板2；参照图1所示，图1为本发明使用2节距结构测量线性位移时的结构示意图，收发板1上布置有第一节距三相接收绕组1.1、第二节距三相接收绕组1.2和发射绕组1.3。
发射绕组1.3用两个近似闭合的矩形线圈1.3.1和1.3.2以同向串联的方式分别包围第一节距三相接收绕组1.1和第二节距三相接收绕组1.2，因此，对发射绕组1.3的一次驱动便可同时在被其包围的两个三相接收绕组1.1和1.2中感应包含位置信息的三相电动势，从而并行测得被测位置在这两个不同节距内的空间相位或位移。这不仅有利于降低功耗，也增强了对传感器快速移动的跟踪能力；此外同向串联使发射绕组1.3的电感量成倍增加，使其驱动电流的上升率成倍降低，在功耗不变的情况下可将其驱动电流的持续时间延长50％以上，有利于传感信号的稳定和处理。布局时应注意保持矩形线圈和三相接收绕组之间的对称，并在测量轴线方向与各接收绕组留有足够间距以确保发射绕组1.3的驱动电流在两个三相接收绕组区域产生的磁场近似为不沿测量轴线变化的二维磁场，因而直接在各接收绕组感应的电动势为0。
第一节距三相接收绕组1.1包含3个结构相同、按A—C—B的顺序空间依次移位1/6节距的A相绕组1.1.1、B相绕组1.1.3和极性相反的C相绕组1.1.2，各相绕组的节距均为P 1，相位按A—B—C的顺序依次相差120°；因每个节距对应360°，因此三相接收绕组也可按A—B—C的顺序空间依次移位1/3节距。图2A为每相接收绕组的连接分解图，绕组由线圈连接而成，线圈由镜像对称的上、下两条边组成。图中虚线表示每个线圈位于PCB底层(Bottom Layer)的一条边，实线表示每个线圈位于PCB底层之上一层的另一条边，后称上层边，这两层之间的厚度很薄，仅约0.1mm；整个线圈跨越1个节距，但两条边的中间平行线段只相隔半个节距，PCB不同层之间由过孔(Via，图中的黑圆点)连通。与三相交流电机的整距波绕组类似，三相绕组按60°相带(即每相绕组在每个半节距下所占的空间对应60°角度，也即三相绕组均分每个半节距)划分空间位置，在每个节距范围，相带按A、Z、B、X、C、Y的顺序排列，其中X、Y、Z分别为A、B、C相的负相带，即相差半个节距或180°；把属于同一相的所有上层边平行线段位于负相带的线圈依次串联起来构成一组，再把属于同一相的所有上层边平行线段位于正相带的线圈依次串联起来构成另一组；将这两组线圈分别头头相连、尾尾相接形成一个闭合回路，从任意位置切开引出两个端子即得该相绕组。对头尾端部略作调整可得结构更为一致、对称度更高的绕组布局，如图2B所示，由此形成的三相接收绕组如图2C所示。
如此布置的三相接收绕组其感应电动势沿测量轴线变化的波形不含偶次谐波，线电压中还不含3、6、9等3的倍数次谐波；谐波幅值随其次数增高单调下降，因此三相接收 绕组线电压中的主要谐波分量是5次和7次谐波。为此可将至少为2的M个图2B所示的集中式绕组在空间依次移相60°/M，然后串联形成一个分布绕组；这M个子绕组的感应电动势所包含的基波及各次谐波的幅值分别对应相等，但基波相位依次相差60°/M而v次谐波相位依次相差v·60°/M，因此其基波分布因数(分布因数＝合成电动势幅值/M倍子绕组电动势幅值)和各谐波分布因数不同。不难推得它对v次谐波的抑制比R v：
Figure PCTCN2020116882-appb-000001
当M＝2时：R 5＝R 7＝3.73，即对5次、7次谐波的衰减均是对基波衰减的3.73倍，可见效果斐然。
空间移相60°/M，在节距P 1较大时，可通过平移P 1/(6M)以交叠方式实现，如图3A所示；当节距P 1较小时，交叠方式无法布线，只能通过平移L·P 1+P 1/(6M)以平铺方式实现，其中：L、M为整数，M≥2，L·P 1大于子绕组长度，如图3B所示。还可将M个结构相同的分布绕组作为子绕组在空间依次移相60°/M，然后串联形成混合分布绕组，籍此获得相乘的谐波抑制比，如图3C所示。图3C中以交叠方式移相30°的两段子集中绕组连成子分布绕组，将两个这样的子分布绕组再以平铺方式移相30°连成混合分布绕组，它对5次、7次谐波的抑制比均为3.73×3.73＝13.91；而直接用4段子集中绕组依次移相15°形成的分布绕组，根据表达式(a)求得的谐波抑制比R 5＝4.66、R 7＝6.08，都不及前者的一半。图3A、3B、3C中上面的单相绕组图，位于PCB不同层的用于串联各子绕组的一对连线本应重叠，但为表明走线，图中故意将其错开绘制。
第二节距三相接收绕组1.2同样包含3个结构相同、按A—C—B的顺序空间依次移位1/6节距的A相绕组1.2.1、B相绕组1.2.3和极性相反的C相绕组1.2.2，各相绕组的节距均为P 2，相位按A—B—C的顺序依次相差120°，绕组连线方法与第一节距三相接收绕组1.1完全相同。
收发板1上还应包含完成测量所需的电子线路，用于驱动发射绕组1.3、处理接收绕组1.1和1.2中的传感信号、显示测量结果等，因此收发板1一般制作在4层PCB板上。
励磁板2上布置有沿测量轴线展开的两列励磁线圈2.1和2.2(为清晰起见，图中已省略位于接收绕组下方的励磁线圈)，励磁线圈序列2.1与收发板1上的第一节距三相接收绕组1.1不仅节距相同(P 1)，而且中心线重合；同样，励磁线圈序列2.2与第二节距三相接收绕组1.2节距相同(P 2)、中心线重合。两列励磁线圈的形状均采用圆角矩形短 路环以削弱邻近磁场的影响，励磁线圈沿测量轴线的含边宽度等于各自节距的一半。励磁板2的结构简单，可制作在单面PCB板上。
由于发射绕组1.3的两个矩形线圈1.3.1和1.3.2的长度(沿测量轴线方向的尺度)远大于宽度，而且在长度方向与各三相接收绕组留有足够间距，故对发射绕组驱动电流在各接收绕组区域产生的磁场而言，矩形线圈可近似为无限长，其电流在各接收绕组区域产生的磁场近似为不沿测量轴线变化的二维磁场，因此直接在各接收绕组感应的电动势为0。
本发明使用2节距结构测量角位移时，其传感器结构如图4所示。它由可围绕转轴A相对转动的两部分组成：收发板1和励磁板2。收发板1上布置有沿同心圆弧展开的第一节距三相接收绕组1.1、第二节距三相接收绕组1.2和发射绕组1.3。发射绕组1.3用两个近似闭合的同心圆弧形线圈1.3.1和1.3.2以同向串联的方式分别包围第一节距三相接收绕组1.1和第二节距三相接收绕组1.2。三相接收绕组1.1和1.2均使用分布式绕组，其节距分别为P 1和P 2，均由3个依次相差120°的相绕组组成。以混合方式实现的第一节距分布式三相接收绕组的连接示意图如图5所示。励磁板2上布置有两列沿同心圆周展开的励磁线圈2.1和2.2(为清晰起见，图中已省略位于接收绕组下方的励磁线圈)，它们分别与收发板1上的两个三相接收绕组1.1和1.2节距相等、中心线重合；两列励磁线圈的形状均采用由两条同心圆弧和两条径向直线围成的短路环，沿测量弧线所跨的角度等于各自节距的一半。除了按同心圆弧(周)分布、节距按圆心角计算外，其它情况与线性位移测量时完全相同，在此不再重复。
如将电源电压V CC经导通的驱动开关施加到发射绕组1.3的两端，在远小于回路时间常数(μs量级)的时间段内，将在发射绕组1.3中产生上升速度很快(10mA/ns量级)的线性时变驱动电流；该电流产生的线性时变磁场将在与其耦合的励磁板2上的两列励磁线圈中感应随时间线性增长的电涡流，各列励磁线圈中的电涡流在与其重叠的三相接收绕组区域产生以其节距为周期沿测量路径做周期变化的线性时变磁场，从而在与其重叠的三相接收绕组中感应不随时间变化(传感器不动时)、但以该三相接收绕组节距为周期随被测位置(收发板1和励磁板2的相对位置)做周期变化的三相电动势。采用分布绕组后，三相接收绕组线电压中的空间谐波分量已大幅衰减，因此，在推导线、相电压关系时可假定三相接收绕组的相电压也只含基波分量，通过选择合适的坐标原点，可设三相接收绕组的相电压如下：
Figure PCTCN2020116882-appb-000002
Figure PCTCN2020116882-appb-000003
式中：E m为每相电动势幅值，P为该三相接收绕组的节距，x为被测位移，A、B、C相电压依次滞后2π/3弧度(120°)。
据此可得如下线电压关系：
Figure PCTCN2020116882-appb-000004
可见，它们按所列顺序依次滞后π/3弧度。如以相同的时间间隔依次对6个线电压按所列的A—B、A—C、B—C、B—A、C—A、C—B顺序循环扫描采样即可合成以下离散时间正弦信号：
Figure PCTCN2020116882-appb-000005
式中：
Figure PCTCN2020116882-appb-000006
为线电压幅值。
显然它是对以下连续时间正弦信号以时间间隔T/6(T为时间周期)进行抽样的结果：
Figure PCTCN2020116882-appb-000007
上式已将被测位置在节距P内的空间相位2πx/P转换成连续时间正弦信号u(t)的初相位(含固定偏移π弧度)。
滤除离散时间正弦信号u s(n)的各次谐波即可还原出被抽样的连续时间正弦信号u(t)，测出其过零点与相位零点的时间差便可得到被测位置在该节距内的空间相位或位移。
实现以上设想的具体电路如图6所示，整个电路由一颗3V锂电池供电，按功能可分为中控、接口和测量三个单元。中控单元只包括低功耗微控制器13。接口单元包括按键输入、液晶(LCD)驱动电路、测量接口电路和电源变换电路。按键输入和液晶驱动电路 用于与用户交互；测量接口用于对测量事件的响应和处理；电源变换电路负责对微控制器13供电和产生电源电压一半的V CC/2用作模拟信号地AGND。余下电路即为测量单元，测量单元包括振荡器、分频电路3、由驱动和采样脉冲形成电路4和线电压扫描控制信号发生器5组成的信号发生器、由模拟开关组6(S 1—S 10)、采样保持电容C 1和C 2、差分放大器7、低通滤波器8、过零检测器9组成的模拟信号处理电路、由同步延时电路(由D型触发器FF 21、FF 22、FF 23组成)、加法计数器10、随机访问存储器11和12、同步捕捉电路(由D型触发器FF 31、FF 32、与门AG 31组成)组成的相位量化电路以及发射绕组驱动功率管T 1，模拟信号处理电路、随机访问存储器和同步捕捉电路各设两组形成两个并行处理通道；振荡器产生频率为f M的系统时钟，为分频电路3、开关电容低通滤波器8、12K进位加法计数器10、D型触发器FF 32提供输入时钟；分频电路3产生2路输出：K分频(K的取值由所需细分数决定)的S CLK用作驱动和采样脉冲形成电路4的触发时钟、2K分频的P CLK用作线电压扫描控制信号发生器5的输入时钟；驱动和采样脉冲形成电路4分别连接发射绕组驱动功率管T 1和模拟开关组6，线电压扫描控制信号发生器5分别连接模拟开关组6和同步延时电路，模拟开关组6、差分放大器7、低通滤波器8、过零检测器9、同步捕捉电路依次顺序相连，采样保持电容C 1和C 2分别连接在差分放大器7的两个输入端与模拟信号地之间；同步延时电路分别连接同步捕捉电路和12K进位加法计数器10，12K进位加法计数器10和同步捕捉电路又同时与随机访问存储器11和12相连；为降低功耗，测量单元可被使能(Enable)和禁能(Disable)，禁能方法可以是简单的断开电源或者是既停止系统时钟又关闭(Shutdown)模拟电路等。
收发板1上的三相接收绕组1.1和1.2均采用星形(Y)联结，两个中性点均和模拟信号地AGND相连，发射绕组1.3经驱动开关(NMOS功率管T 1)连接于电源两端。
每次测量时，微控制器13先使能测量单元、清零(Clear)初始化信号INI然后将其置壹(Set)使测量单元开始工作；完成测量后微控制器13禁能测量单元使其停止运行，籍此间歇工作降低电路功耗。
发射绕组1.3的电感量很小(不到1μH)，对其施加电源电压V CC后，绕组中的电流急剧上升(上升率在10mA/ns量级)，因此对功率管T 1的导通时间应精准控制，顾及功率管的开关特性及采样电路的暂态过程，其驱动信号TG的脉冲宽度宜在30ns左右。
驱动和采样脉冲形成电路4产生发射绕组1.3的驱动开关——NMOS功率管T 1的栅极驱动信号TG和模拟开关S 9及S 10(及第二节距三相接收绕组1.2处理通道中的2个对应开关)的采样和保存控制信号SAH。功率管T 1的输入电容较大(100pF量级)，因此驱 动信号TG需经缓冲器输出以增强其驱动能力。阻容延时在不做微调时很难实现如此短暂且有精度要求的窄脉冲，此外还受电源电压、温度等因素的影响明显，最佳方案是利用反相器传输延时形成批量分散性小、持续时间短、无需微调的驱动和采样脉冲，具体电路如图8所示。当输入信号(触发时钟)S CLK为低电平时，与非门G NA输入端2为低电平，输入端1为高电平(因奇数次反相)，G NA的输出Y为高电平；在触发时钟S CLK的上升沿，与非门G NA输入端2跃变为高电平，输入端1仍保持为高电平(因存在传输延时)，G NA的输出Y跃变为低电平；设每级反相器的传输延时为t pd(ns量级)，所用奇数个反相器的个数用(2N+1)表示，则经过(2N+1)t pd的延时后，与非门G NA输入端1跃变为低电平(因奇数次反相)，输出Y跃变为高电平并一直保持直至S CLK的上升沿再次到来(因S CLK变为低电平时G NA仍输出高电平)。因此，输入信号S CLK的每个上升沿触发与非门G NA输出一个负极性的窄脉冲，脉冲宽度为(2N+1)t pd。对与非门G NA的输出Y反相得到正极性的采样和保持控制信号SAH，对G NA的输出Y反相缓冲即得功率管T 1的栅极驱动信号TG。
对发射绕组1.3的每次驱动同时在三相接收绕组1.1和1.2中感应三相电动势，因而可对两个三相接收绕组的输出信号做并行处理，所用电路完全相同，故此以下讨论只针对第一节距三相接收绕组1.1展开。
模拟开关组6(S 1—S 10)、采样保持电容C 1和C 2、差分放大器7共同完成对第一节距三相接收绕组1.1的线电压循环扫描采样和放大，以合成表达式(d)所描述的离散时间正弦信号u s(n)。控制信号Q 0负责接通模拟开关S 1和S 4产生u 1A-u 1B(或其反相u 1B-u 1A)、控制信号Q 1负责接通模拟开关S 2和S 5产生u 1A-u 1C(或其反相u 1C-u 1A)、控制信号Q 2负责接通模拟开关S 3和S 6产生u 1B-u 1C(或其反相u 1C-u 1B)，控制信号Q 3负责单刀双投模拟开关S 7和S 8的通道切换决定是否反相，控制信号SAH负责接通模拟开关S 9和S 10对输入的两个相电压进行采样并将结果分别保存到采样保持电容C 1和C 2中，最后由差分放大器7实施减法运算和放大得到一个线电压的采样和零阶保持信号。可见将模拟开关组6的10个模拟开关组成一个三级开关串联的结构依次完成对相电压的选择、交换和采样，便无需对控制信号译码而使线路大为简化。只要使控制信号Q 0、Q 1、Q 2依次为高电平，Q 3对Q 0、Q 1、Q 2的每次循环发生翻转便可获得按A—B、A—C、B—C、B—A、C—A、C—B顺序的线电压循环扫描采样，合成表达式(d)所描述的离散时间正弦信号u s(n)。
产生上述Q 0、Q 1、Q 2和Q 3信号的线电压扫描控制信号发生器5的实现电路如图9所示，各信号波形参见图7。D型触发器FF 11、FF 12、FF 13构成循环移位的环形计数器，初始化信号INI将其预设为1、0、0，故任何时候有且只有一个为高电平；D型触发器FF 14 对Q 0的上升沿计数，因此其输出Q 3对Q 0、Q 1、Q 2的每次循环发生一次翻转；时钟信号P CLK来自分频电路3，是对触发时钟S CLK进行2分频的输出；驱动和采样脉冲形成电路4在触发时钟S CLK的每个上升沿输出一个驱动和一个采样脉冲，因此每个线电压被连续采样2次，藉以增加传感器相对运动时离散时间正弦信号u s(n)的样点数。P CLK对频率为f M的系统时钟2K分频，每个循环扫描周期(即Q 3周期)共6个节拍，包含2K×6＝12K个系统时钟周期，对应线电压在一个节距内变化一周，因此每个节距或空间2π弧度被细分为12K等份。
专利CN101949682B公开了一种绝对位置电容式位移传感器，其中包含对离散时间正弦信号进行低通滤波和过零检测后利用加法计数器测得被测位置在各波长内的位移及绝对定位被测位移的方法，在此引入其相关内容。
差分放大器7合成的离散时间正弦信号u s(n)经开关电容(为便于集成)低通滤波器8滤波后，还原出类似于表达式(e)所描述的连续时间正弦信号u r(t)，此时被测位移x在节距P 1内的空间相位2πx/P 1已被转换成u r(t)的初相位；过零检测器9将连续时间正弦信号u r(t)整形成方波信号U Z，其上升沿对应u r(t)从负到正的过零点、下降沿对应u r(t)从正到负的过零点，两者相位相差180°，在传感器不动时两者时间相差12K/2＝6K个频率为f M的系统时钟周期，因此两者均可用于测量空间相位或位移；连续时间正弦信号u r(t)的过零点相对相位零点的时间差与被测位置在节距P 1内的空间相位(可能包含固定偏移)成正比，因此可以利用方波信号U Z的上升沿或下降沿捕捉从相位零点开始计数、以系统时钟为计数脉冲的加法计数器的计数值得到被测位置在节距P 1内的空间相位或位移的量化编码(可能包含固定偏移)。由于测量电路间歇工作，捕捉计数前必须等待足够时间以使电路暂态过程足够衰减，为此设置由D型触发器FF 21、FF 22、FF 23组成的同步延时电路和由D型触发器FF 31、FF 32、与门AG 31组成的同步捕捉电路，其信号波形参见图7。同步延时电路、同步捕捉电路、12K进位(因每个节距被细分为12K等份)加法计数器10、随机访问存储器11和12共同完成相位量化任务。
初始化信号INI将D型触发器FF 21、FF 22、FF 23预设为1、1、0，FF 23输出的低电平C E将12K进位加法计数器10异步清零并阻止其计数；D型触发器FF 21、FF 22构成2位异步减法计数器，对来自线电压扫描控制信号发生器5的输出信号Q 3的上升沿进行减计数；当Q 3的第4个上升沿到达时，D型触发器FF 22的输出信号W 1产生正跳变从而将FF 23的输出信号C E置为1：12K进位加法计数器10从0开始计数、同步捕捉电路解除捕捉封锁；因此该同步延时电路产生4个Q 3周期的延时并将相位零点设置在Q 3的上升沿。
初始化信号INI将D型触发器FF 31和FF 32异步清0，在延时时间未到时，同步延时电路输出的低电平C E使D型触发器FF 31始终输出低电平的C S、与门AG 31始终输出低电平的C P、D型触发器FF 32始终输出低电平的【P 1上升沿捕捉】和高电平的【P 1下降沿捕捉】信号，从而封锁对12K进位加法计数器10的计数值捕捉；延时时间到达后，同步延时电路的输出信号C E被置为高电平、方波信号U Z在此之后的第一个上升沿将D型触发器FF 31的输出信号C S也置为高电平、与门AG 31的输出信号C P自此之后便和方波信号U Z相同；在12K进位加法计数器10不产生计数的系统时钟下降沿，D型触发器FF 32对与门AG 31的输出信号C P进行同步后输出同相位的【P 1上升沿捕捉】和反相位的【P 1下降沿捕捉】信号，分别在方波信号U Z的上升沿和下降沿以同步方式捕捉12K进位加法计数器10的计数值，将其分别保存在随机访问存储器11和12中，同时通知微控制器13捕捉事件的发生，微控制器13籍此读入该捕捉值并判断测量是否完成。从图7的波形图可以看出，上升沿总是先发生捕捉，下降沿滞后一段时间才发生捕捉，因此根据上升沿的捕捉值N 1r和下降沿的捕捉值N 1f便可推断传感器是否运动以及它的运动方向。
当两个节距的上升沿和下降沿都完成捕捉后，微控制器13禁能测量单元使其停止工作，然后通过软件计算被测位移。
与节距P 1对应的空间频率F 1＝1/P 1，与节距P 2对应的空间频率F 2＝1/P 2，其频率差F M＝F 2-F 1对应更大的空间周期——中节距P M：
Figure PCTCN2020116882-appb-000008
如欲P M＝m·P 1，式中m为整数，称为中节距波长比，则：
Figure PCTCN2020116882-appb-000009
为便于规模化生产，本发明优先选用较小的波长比m＝15，并取触发时钟S CLK的分频数K＝2 5＝32，则每个节距被细分为12K＝384份，当线性位移的分辨率为10μm时，P 1＝3.84mm，P 2＝3.6mm，P M＝15P 1＝16P 2＝57.6mm。
设被测位置的位移为x，它在节距P 1、P 2内的相位量化编码分别为N 1和N 2(量化编码可由上升沿捕捉值、下降沿捕捉值或两者的中点值求得)，则有：
Figure PCTCN2020116882-appb-000010
Figure PCTCN2020116882-appb-000011
Figure PCTCN2020116882-appb-000012
Figure PCTCN2020116882-appb-000013
Figure PCTCN2020116882-appb-000014
式中：λ为比例系数，mod为求模运算。
合理选择比例系数λ使P 1·λ/(2π)＝1，则被测位置在节距P 1内的位移x 1＝x mod P 1＝N 1，其步距为1，后称细位移x F(相应地称节距P 1为细节距P F)；被测位置在中节距P M内的位移x M＝x mod P M＝m·(N 2-N 1)，其步距随量程一起扩大了m倍，故称中位移x M；中位移x M是对两个不同节距的相位量化编码相减和放大的结果，这种用非累加方式确定位移的方法称为绝对定位；中节距P M只被扩大了m倍，尺度有限，为此对相邻两次测量的中位移增量Δx M进行累加获得没有量程限制、步距为m的总位移x T＝∑(Δx M)；由于中位移x M是通过两个不同节距绝对定位得到、总位移x T是依靠累加中位移增量获得，因此称该测量方法为混合定位。计算中位移增量Δx M时需结合传感器的运动方向，使其结果与运动方向一致，即正向运动产生正增量、反向运动产生负增量，因此必要时应对差值m·(ΔN 2-ΔN 1)进行加上或减去中节距P M(需要进行单位换算，下同)的修正，这是既用上升沿捕捉、又用下降沿捕捉的主要原因。
根据以下关系：
x T≈K M·P M+x M                 (i)
x M≈K F·P F+x F                 (j)
式中：K M为中节距个数，K F为细节距个数，两者均为整数。
即可求出总位移x T所包含的中节距整数个数K M、中位移x M所包含的细节距整数个数K F以及步距为1的被测位移x：
x＝K M·P M+K F·P F+x F＝(m·K M+K F)·P F+x F             (k)
本混合定位算法需要不断累加中位移增量，两次测量之间的传感器位移不能超过1个中节距范围，故存在最低测量频率要求。例如：P M＝57.6mm时，要获得2.5m/s的测量速度，测量频率不应低于44次/秒；因此本发明优先选用50次/秒的测量频率，每6次测量显示一次测量结果(每秒约显示8次)；考虑到刚开始移动时的速度较低，为进一步降低功耗，可将传感器静止和待机(静止并关闭显示)时的测量频率降低一半成为25 次/秒。每次测量约需5个循环扫描周期(Q 3周期，4个用于延时)共384×5＝1920个频率为f M的系统时钟周期，在频率f M＝1MHz时，完成测量约需2ms时间，因此测量电路间歇工作的占空比在传感器移动时为2/20＝0.1，静止和待机时为2/40＝0.05。
如将该使用2节距结构的传感器应用于指示表、小卡尺等短量程量具则可只使用中位移x M和细位移x F在中节距范围确定被测位移x而无需累加中位移增量Δx M扩展量程从而成为绝对定位的感应式位移传感器。它与已有绝对位置感应式位移传感器相比，具有以下优点：
1.发射绕组同时包围两个不同节距的三相接收绕组，一次驱动便可同时得到两组节距不同的传感信号，从而可并行完成被测位置在两个不同节距内的位移测量。这不仅节省一组驱动电路和一次发射绕组的驱动功耗，也增强了对传感器快速移动的跟踪能力；
2.分布式接收绕组对空间谐波的抑制能力很强，因而无需使用占用面积大、形状复杂的正弦形绕组，使接收绕组的布线密度增加、线圈数量增多、接收信号增强；
3.励磁线圈采用简单的短路环形状，比之互连的多耦合回路结构路径更短、回路电阻更小、感应的涡流更大、接收信号更强；比之矩形铜箔结构邻近磁场的干扰更小、精度更高；
4.利用传输延时形成的驱动脉冲在发射绕组激发持续时间很短的线性时变电流，在各接收绕组感应不随时间变化(传感器不动时)的电动势，大幅降低对采样电路的时序要求；将传感信号经三级串联的模拟开关和差分放大器合成为离散时间正弦信号后，通过低通滤波、过零检测、加法计数等简单电路完成对被测位置在两个不同节距内的空间相位或位移测量，因此测量电路易于实现和集成。
只在中节距范围绝对定位被测位移x时，需将中位移x M＝m·(N 2-N 1)映射到半闭区间[0，P M)以与细位移x F∈[0，P F)匹配，即当x M为负时将其加上中节距P M使其变正，然后由关系式(j)求出中位移x M所包含的细节距整数个数K F，最后按下式计算被测位移x：
x＝K F·P F+x F                 (l)
综上所述，本发明无需对传感器的结构和测量电路做任何变动仅通过软件算法便可实现绝对定位和混合定位两种测量方法，在使用2节距结构时，绝对定位功耗低但量程短(限于中节距范围)，混合定位没有量程限制但有最低测量频率要求。
为降低测量功耗，微控制器13采用中断驱动模式：即每次中断返回便立即进入休眠，直到有新的中断再次将其唤醒。完成测量需使用2个中断向量：定时器中断和捕捉中断。定时器中断用于发起测量，捕捉中断用于响应捕捉事件：【P 1上升沿捕捉】、【P 1下降沿捕捉】、【P 2上升沿捕捉】和【P 2下降沿捕捉】四个外部中断请求。总结前面的介绍，现将使用2节距结构时的混合定位和绝对定位两种算法的中断处理流程归纳如下：
Ⅰ、混合定位
定时器每20ms产生一次中断，发起最高频率为50次/秒的测量，传感器静止或待机时将测量频率减半；为及时响应捕捉事件，还开启(Enable)【P 1上升沿捕捉】、【P 1下降沿捕捉】、【P 2上升沿捕捉】和【P 2下降沿捕捉】4个外部中断请求，其处理流程如图10所示。
捕捉中断服务程序对测量过程和数据进行处理。为避免重复中断，读入该中断源的捕捉值后即关闭(Disable)该捕捉的中断请求；如果测量已全部完成(4个捕捉的中断请求均被关闭)：禁能测量单元使其停止运行、计算中位移x M＝m·(N 2-N 1)、判断传感器是否运动并设置状态标志以配合定时器中断实现动/静状态变频测量、累加中位移增量Δx M得到没有量程限制的总位移x T＝∑(Δx M)；需要显示测量结果时(距上次显示已是第6次测量)：由关系式(i)求出总位移x T所包含的中节距整数个数K M、由关系式(j)求出中位移x M所包含的细节距整数个数K F、根据公式(k)计算被测位移x及按用户要求显示测量结果。其流程图如图11所示。
Ⅱ、绝对定位
定时器每125ms产生一次中断，发起最高频率为8次/秒的测量，传感器静止或待机时同样将测量频率减半，因此其处理流程和混合定位时完全相同，如图10所示。
捕捉中断的处理流程与混合定位时相似，只是确定被测位移的方法有所不同且每次测量都显示测量结果，如图12所示。读入该中断源的捕捉值、关闭该捕捉的中断请求；如果测量已全部完成：禁能测量单元、计算中位移x M＝m·(N 2-N 1)、判断传感器是否运动并设置状态标志、将中位移x M映射到半闭区间[0，P M)、由关系式(j)求出中位移x M所包含的细节距整数个数K F、根据公式(l)计算被测位移x及按用户要求显示测量结果。
如果空间容许，本发明也可使用3节距结构，在测量线性位移时的传感器结构如图13所示。它由可沿测量轴线相对移动的两部分组成：收发板1和励磁板2。
收发板1上布置有三个节距不同的三相接收绕组1.1、1.2、1.3和两个发射绕组1.4、1.5。发射绕组1.4用两个近似闭合的矩形线圈1.4.1和1.4.2以同向串联的方式分别包围 第一节距三相接收绕组1.1和第二节距三相接收绕组1.2；发射绕组1.5用两个近似闭合的矩形线圈1.4.1和1.5.2以同向串联的方式分别包围第一节距三相接收绕组1.1和第三节距三相接收绕组1.3。三相接收绕组1.1、1.2和1.3均使用分布式绕组，其节距分别为P 1、P 2和P 3，均由3个依次相差120°的相绕组组成。
励磁板2上布置有3列沿测量轴线展开的励磁线圈2.1、2.2和2.3(为清晰起见，图中已省略位于接收绕组下方的励磁线圈)，它们分别与收发板1上的3个三相接收绕组1.1、1.2和1.3节距相等、中心线重合；3列励磁线圈的形状均采用圆角矩形短路环，沿测量轴线的含边宽度等于各自节距的一半。
发射绕组1.4和被其包围的第一节距三相接收绕组1.1及第二节距三相接收绕组1.2构成一个前面已详细说明的使用2节距结构的收发板部分，用于测量中节距内的位移；发射绕组1.5和被其包围的第一节距三相接收绕组1.1及第三节距三相接收绕组1.3构成另一个使用2节距结构的收发板部分，用于测量比中节距步距和范围更大的粗节距内的位移；两个使用2节距的结构共用矩形线圈1.4.1和被其包围的三相接收绕组1.1，因此3节距结构等同于两个2节距结构的组合，其收发板1和励磁板2均由使用2节距结构的线性位移传感器扩展而来，设计方法完全相同，在此不再重复。
类似于中节距P M的推导过程可得与表达式(f)相似的粗节距P C：
Figure PCTCN2020116882-appb-000015
如欲P C＝n·P M＝m·n·P 1，式中n为整数，称为粗节距波长比，则：
Figure PCTCN2020116882-appb-000016
本发明在使用2节距结构的优选参数基础上选用粗节距波长比n＝m＝15，则使用3节距结构的传感器参数：P 1＝3.84mm，P 2＝3.6mm，P 3＝3.823mm，P M＝15P 1＝57.6mm，P C＝225P 1＝864mm。
如果能分步并行测得被测位置在第一个2节距结构中的两个不同节距P 1、P 2内的相位量化编码N 11、N 12和并行测得被测位置在第二个2节距结构中的两个不同节距P 1、P 3内的相位量化编码N 21、N 23，则类似于表达式(h)的推导过程，被测位置在节距P 1内的位移x 1＝x mod P 1＝N 21(因节距P 1、P 3相隔较远，N 21的精度高于N 11)，其步距为1，后称细位移x F(相应地称节距P 1为细节距P F)；被测位置在中节距P M内的位移x M＝x mod P M＝m·(N 12-N 11)，其步距为m，后称中位移x M；被测位置在粗节距P C内的位 移x C＝x mod P C＝m·n·(N 23-N 21)，其步距为m·n，后称粗位移x C。得到粗、中、细位移x C、x M和x F之后，便可按以下步骤计算被测位移x：将粗位移x C映射到半闭区间[0，P C)以与细位移x F∈[0，P F)匹配、用粗位移x C代替总位移x T由关系式(i)求出粗位移x C所包含的中节距整数个数K M、由关系式(j)求出中位移x M所包含的细节距整数个数K F、根据公式(k)计算被测位置在粗节距P C内的绝对位移x a；一般应用时该绝对位移即为待求的被测位移x＝x a，但在大量程(超出粗节距P C范围)应用时则可通过累加相邻两次测量的绝对位移增量Δx a得到没有量程限制的被测位移x＝∑(Δx a)。粗节距数值较大，即使累加绝对位移增量也无需提高测量频率。
实现以上分步测量的具体电路如图14所示，它在图6所示使用2节距结构的测量电路上仅仅增加一个NMOS功率管T 2和4个多路模拟开关S 11—S 14(含控制信号【节距选择】)，因此工作原理完全相同。微控制器13在使能测量单元后，输出低电平的【节距选择】信号：将驱动信号TG的输出端子连接到驱动功率管T 1的栅极使发射绕组1.4(由线圈1.4.1和1.4.2组成)接受驱动、将第二节距三相接收绕组1.2的输出端子连接到第二处理通道，然后清零初始化信号INI并将其置壹发起第一次测量，并行测得被测位置在第一节距P 1和第二节距P 2内的相位量化编码N 11和N 12；接着微控制器13输出高电平的【节距选择】信号：将驱动信号TG的输出端子改接到驱动功率管T 2的栅极使发射绕组1.5(由线圈1.4.1和1.5.2组成)接受驱动、将第三节距三相接收绕组1.3的输出端子连接到第二处理通道，再次清零初始化信号INI并将其置壹发起第二次测量，并行测得被测位置在第一节距P 1和第三节距P 3内的相位量化编码N 21和N 23；此后微控制器禁能测量单元、利用绝对定位算法计算被测位置在粗节距内的绝对位移x a、根据需要累加相邻两次测量的绝对位移增量Δx a扩展量程。
和使用2节距结构时一样，微控制器13也采用中断驱动模式，完成测量需使用2个中断向量：定时器中断和捕捉中断。
定时器每125ms产生一次中断，发起最高频率为8次/秒的测量，传感器静止或待机时将测量频率减半，其处理流程与使用2节距结构时相比，多出一步“清零【节距选择】信号”，如图15所示。
捕捉中断的处理流程如图16所示。读入该中断源的捕捉值、关闭该捕捉的中断请求；如果完成的是第一次测量：置壹【节距选择】信号、清零初始化信号INI、清除各捕捉中断请求标志、重新打开各捕捉中断请求、置壹初始化信号INI后发起第二次测量；如果完成的是第二次测量：禁能测量单元、计算中位移x M＝m·(N 12-N 11)、判断传感器是否 移动并设置状态标志、计算粗位移x C＝m·n·(N 23-N 21)、将粗位移x C映射到半闭区间[0，P C)、用粗位移x C代替总位移x T由关系式(i)求出粗位移x C所包含的中节距整数个数K M、由关系式(j)求出中位移x M所包含的细节距整数个数K F、根据公式(k)计算被测位置在粗节距P C内的绝对位移x a、如需扩展量程则累加绝对位移增量Δx a、最后按用户要求显示测量结果。
也可使用3节距结构测量角位移，但因角位移仅需在360°范围内绝对定位，一般情况使用2节距结构便已满足要求，故此不再说明。
上述实施例，仅为对本发明的目的、技术方案和有益效果作进一步详细说明的具体个例，本发明并非限定于此。凡在本发明公开的范围之内所做的任何修改、等同替换、改进等，均包含在本发明的保护范围之内。

Claims (10)

  1. 混合定位电磁感应式位移传感器，其特征在于所述传感器包括有可沿测量路径相对运动的收发板(1)和励磁板(2)；
    所述的收发板(1)上布置有测量电路以及沿测量路径展开的至少一个发射绕组，每个发射绕组用两个近似闭合的矩形线圈以同向串联的方式分别包围第一节距三相接收绕组和第二节距三相接收绕组，每个三相接收绕组包含3个结构相同、依次相差120°的相绕组，发射绕组和三相接收绕组均与所述测量电路相连；所述测量电路包括有中控单元、接口单元和测量单元；中控单元包含微控制器(13)；接口单元包括有与微控制器(13)连接的按键输入电路、液晶驱动电路、测量接口电路和电源变换电路；
    所述的励磁板(2)上布置有沿测量路径展开的与收发板(1)上三相接收绕组数量相等的至少两列励磁线圈，各列励磁线圈分别与收发板(1)上对应的三相接收绕组节距相等、中心线重合，沿测量路径的尺寸为各自节距的一半。
  2. 根据权利要求1所述的混合定位电磁感应式位移传感器，其特征在于：所述的测量单元包括振荡器、分频电路、由驱动和采样脉冲形成电路和线电压扫描控制信号发生器组成的信号发生器、由模拟开关组(6)、采样保持电容、差分放大器、低通滤波器、过零检测器组成的模拟信号处理电路、由同步延时电路、加法计数器、随机访问存储器和同步捕捉电路组成的相位量化电路以及发射绕组驱动功率管，模拟信号处理电路、随机访问存储器和同步捕捉电路各设两组形成两个并行处理通道；
    振荡器直接或经分频电路为驱动和采样脉冲形成电路、线电压扫描控制信号发生器、低通滤波器、同步捕捉电路及加法计数器提供输入时钟；驱动和采样脉冲形成电路与模拟开关组连接的同时还直接或经多路开关与发射绕组驱动功率管相连，线电压扫描控制信号发生器分别连接模拟开关组和同步延时电路；模拟开关组、差分放大器、低通滤波器、过零检测器、同步捕捉电路依次顺序相连，采样保持电容连接在差分放大器的输入端与模拟信号地之间；同步延时电路分别连接同步捕捉电路和加法计数器，加法计数器和同步捕捉电路又同时与随机访问存储器相连；各发射绕组均经各自的驱动功率管和电源连接，各三相接收绕组均按星型联结且中性点和模拟信号地相连。
  3. 根据权利要求1所述的混合定位电磁感应式位移传感器，其特征在于：每个发射绕组均用两个近似闭合的线圈以同向串联的方式分别包围两个节距不同的三相接收绕组；所有励磁线圈均采用短路环的形状；每相接收绕组均由至少为2的M个结构相同但空间 依次移相60°/M的子绕组串联而构成一个分布绕组，构成分布绕组的各子绕组本身或者为分布绕组；每次都并行测量被测位置在被同一个发射绕组所包围的两个不同节距内的空间相位或位移；测得被测位置在所需不同节距内的空间相位或位移后微控制器(13)禁能测量单元使其停止运行、利用混合定位或绝对定位算法计算被测位移。
  4. 根据权利要求1所述的混合定位电磁感应式位移传感器，其特征在于：收发板(1)和励磁板(2)可沿测量轴线相对移动；收发板(1)上布置有沿测量轴线展开的一个发射绕组(1.3)和两个节距不同的三相接收绕组(1.1、1.2)；发射绕组(1.3)用两个近似闭合的矩形线圈(1.3.1、1.3.2)以同向串联的方式分别包围节距为P 1的三相接收绕组(1.1)和节距为P 2的三相接收绕组(1.2)；励磁板(2)上布置有沿测量轴线展开的两列励磁线圈(2.1、2.2)，分别与收发板(1)上的两个三相接收绕组(1.1、1.2)节距相同，中心线重合，励磁线圈的形状采用圆角矩形短路环；
    将节距P 2和P 1的空间频率差作为中节距空间频率F M＝1/P 2-1/P 1，得到波长比为m的中节距P M＝1/F M＝P 1·P 2/(P 1-P 2)＝m·P 1＝(m+1)·P 2。
  5. 根据权利要求1所述的混合定位电磁感应式位移传感器，其特征在于：收发板(1)和励磁板(2)可围绕转轴相对转动，节距按角度计算；收发板(1)上布置有沿同心圆弧展开的一个发射绕组(1.3)和两个节距不同的三相接收绕组(1.1、1.2)；发射绕组(1.3)用两个近似闭合的同心圆弧形线圈(1.3.1、1.3.2)以同向串联的方式分别包围节距为P 1的三相接收绕组(1.1)和节距为P 2的三相接收绕组(1.2)；励磁板(2)上布置有沿同心圆周展开的两列励磁线圈(2.1、2.2)，分别与收发板(1)上的两个三相接收绕组(1.1、1.2)节距相同，中心线重合，励磁线圈的形状采用由两条同心圆弧和两条径向直线围成的短路环；
    将节距P 2和P 1的空间频率差作为中节距空间频率F M＝1/P 2-1/P 1，得到波长比为m的中节距P M＝1/F M＝P 1·P 2/(P 1-P 2)＝m·P 1＝(m+1)·P 2。
  6. 根据权利要求1所述的混合定位电磁感应式位移传感器，其特征在于：收发板(1)和励磁板(2)可沿测量轴线相对移动；收发板(1)上布置有沿测量轴线展开的两个发射绕组(1.4、1.5)和三个节距不同的三相接收绕组(1.1、1.2、1.3)，第一个发射绕组(1.4)用两个近似闭合的矩形线圈(1.4.1、1.4.2)以同向串联的方式分别包围节距为P 1的三相接收绕组(1.1)和节距为P 2的三相接收绕组(1.2)，第二个发射绕组(1.5)用两个近似闭合的矩形线圈(1.4.1、1.5.2)以同向串联的方式分别包围节距为P 1的三相接收绕组(1.1)和节距为P 3的三相接收绕组(1.3)；励磁板(2)上布置有沿测量轴线展开 的三列励磁线圈(2.1、2.2、2.3)，分别与收发板(1)上的三个三相接收绕组(1.1、1.2、1.3)节距相同，中心线重合，励磁线圈的形状采用圆角矩形短路环；
    将节距P 2和P 1的空间频率差作为中节距空间频率F M＝1/P 2-1/P 1，得到波长比为m的中节距P M＝1/F M＝P 1·P 2/(P 1-P 2)＝m·P 1＝(m+1)·P 2；将节距P 3和P 1的空间频率差作为粗节距空间频率F C＝1/P 3-1/P 1，得到波长比为n的粗节距P C＝1/F C＝P 1·P 3/(P 1-P 3)＝m·n·P 1＝(m·n+1)·P 3。
  7. 根据权利要求2所述的混合定位电磁感应式位移传感器，其特征在于：
    所述驱动和采样脉冲形成电路包括有级联的奇数个反相器和一个与非门；所述分频电路输出的第一个时钟信号作为所述驱动和采样脉冲形成电路的触发脉冲分别与第一个反相器的输入端子和与非门的一个输入端子相连，级联的奇数个反相器的输出和与非门的另一个输入端子相连，在触发脉冲的每个上升沿，与非门输出一个负极性的窄脉冲，其宽度等于级联的奇数个反相器的总传输延时；对与非门(G NA)的输出反相得到正极性的采样和保持控制信号，对与非门的输出反相缓冲得到正极性的驱动信号；
    所述线电压扫描控制信号发生器由4个D型触发器组成；所述分频电路输出的第二个时钟信号同时和这4个D型触发器的时钟端子相连，所述微控制器输出的初始化信号将这4个D型触发器分别预设为1、0、0、1；前3个D型触发器构成一个循环移位的环形计数器，其输出的3个信号轮流为高电平；第4个D型触发器对第1个D型触发器的输出进行计数，所述环形计数器的每次循环使其输出(Q 3)发生一次翻转。
  8. 根据权利要求2所述的混合定位电磁感应式位移传感器，其特征在于：
    所述模拟开关组组成一个三级开关串联的结构，对输入的三相电压依次完成相电压的选择、交换和采样：第一级开关被配成{S 1、S 4}、{S 2、S 5}和{S 3、S 6}的三对，分别选择相电压对{u A、u B}、{u A、u C}和{u B、u C}；第二级开关根据需要交换第一级开关所选择的相电压对的顺序；第三级开关对第二级开关输出的一对相电压进行采样并将结果分别保存到所述的采样保持电容之中；
    所述差分放大器对保存在采样保持电容中的两个相电压采样实施减法运算和放大得到与之对应的一个线电压的采样和零阶保持信号；在所述信号发生器的输出信号控制下，依次获得按A—B、A—C、B—C、B—A、C—A、C—B顺序的线电压循环扫描采样，将输入的所述三相接收绕组不随时间变化，但随被测位置做周期变化的三相电压合成为一个离散时间正弦信号，被测位置在所述三相接收绕组节距内的空间相位被转换成该离散时间正弦信号的初相位。
  9. 根据权利要求2所述的混合定位电磁感应式位移传感器，其特征在于：
    所述同步延时电路由3个D型触发器组成；所述线电压扫描控制信号发生器的最后一个输出信号与第一个D型触发器的时钟端相连作为同步延时电路的输入时钟，所述微控制器输出的初始化信号将这3个D型触发器分别预设为1、1、0，使输出信号为低电平；前2个D型触发器构成一个2位异步减法计数器，对输入时钟的上升沿进行减计数；当第4个输入时钟的上升沿到达时，使输出信号跃变为高电平；
    所述同步延时电路的输出信号跃变为高电平之后，所述加法计数器从0开始计数、所述同步捕捉电路解除捕捉封锁；
    所述同步捕捉电路由2个D型触发器和一个与门组成；所述微控制器输出的初始化信号将2个D型触发器异步清零，在延时时间未到时封锁对所述加法计数器计数值的捕捉；延时时间到达后，所述同步延时电路的输出信号跃变为高电平、所述模拟信号处理电路输出的方波信号在此之后的第一个上升沿将第一个D型触发器的输出信号也置为高电平、与门的输出信号自此之后便和所述方波信号相同；第二个D型触发器将与门的输出信号和系统时钟下降沿同步后分别在所述方波信号的上升沿和下降沿以同步方式捕捉加法计数器的计数值，将其分别保存在两个随机访问存储器中，由此得到被测位置在与该处理通道相连的所述三相接收绕组节距内的空间相位或位移的量化编码。
  10. 根据权利要求1所述的混合定位电磁感应式位移传感器，其特征在于：
    并行测得被测位置在被发射绕组(1.3)包围的两个不同节距P 1和P 2内的相位量化编码N 1和N 2后，则细位移x F＝N 1、中位移x M＝m·(N 2-N 1)，然后累加相邻两次测量的中位移增量Δx M得到总位移x T＝∑(Δx M)、根据关系x T≈K M·P M+x M求出总位移x T所包含的中节距整数个数K M、根据关系x M≈K F·P F+x F求出中位移x M所包含的细节距整数个数K F、然后根据公式x＝(m·K M+K F)·P F+x F得到没有量程限制的被测位移x；或使用以下绝对定位算法：将中位移x M映射到半闭区间[0，P M)、根据关系x M≈K F·P F+x F求出中位移x M所包含的细节距整数个数K F、根据公式x＝K F·P F+x F得到在中节距范围绝对定位的被测位移x；
    并行测得被测位置在第一个发射绕组(1.4)所包围的两个不同节距P 1和P 2内的相位量化编码N 11和N 12以及并行测得被测位置在第二个发射绕组(1.5)所包围的两个不同节距P 1和P 3内的相位量化编码N 21和N 23之后，则细位移x F＝N 21、中位移x M＝m·(N 12-N 11)、粗位移x C＝m·n·(N 23-N 21)，将粗位移x C映射到半闭区间[0，P C)、根据关系 x C≈K M·P M+x M求出粗位移x C所包含中节距P M的整数个数K M、根据关系x M≈K F·P F+x F求出中位移x M所包含细节距P F的整数个数K F、根据公式x a＝(m·K M+K F)·P F+x F得到在粗节距范围的绝对位移x a、如不扩展量程此即被测位移x＝x a、否则通过累加相邻两次测量的绝对位移增量Δx a得到没有量程限制的被测位移x＝∑(Δx a)。
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