WO2019019555A1 - 一种基于双环预测控制的切换型控制方法 - Google Patents

一种基于双环预测控制的切换型控制方法 Download PDF

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WO2019019555A1
WO2019019555A1 PCT/CN2017/120375 CN2017120375W WO2019019555A1 WO 2019019555 A1 WO2019019555 A1 WO 2019019555A1 CN 2017120375 W CN2017120375 W CN 2017120375W WO 2019019555 A1 WO2019019555 A1 WO 2019019555A1
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control
value
sampling
state
variable
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PCT/CN2017/120375
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French (fr)
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杜贵平
黎嘉健
柳志飞
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华南理工大学
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Priority to JP2019548986A priority Critical patent/JP6866985B2/ja
Priority to US16/089,383 priority patent/US11012000B2/en
Publication of WO2019019555A1 publication Critical patent/WO2019019555A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53873—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with digital control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H02M1/123—Suppression of common mode voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H02M1/126—Arrangements for reducing harmonics from ac input or output using passive filters

Definitions

  • the invention relates to power electronic converter and industrial control technology, in particular to a switching type control method based on double loop prediction control, belonging to the technical field of power electronic converter.
  • the outer ring mainly has Proportional-Integral (PI) control and synovial control, but its dynamic performance is not high.
  • PI Proportional-Integral
  • the system (system refers to the implementation of the control method, generally the power electronic converter, in the present invention refers to the single-phase voltage inverter) performance is closely related to the outer loop control and inner loop control, how to pass The overall performance of the outer loop control lift system is an area worthy of further study.
  • the currently used predictive control methods are deadbeat control and limited control set model predictive control.
  • the deadbeat control can realize the tracking of the reference quantity by the controlled quantity in one control period, has excellent steady state performance, and can realize a fixed switching frequency, but the presence of the modulation unit limits the dynamic performance of the deadbeat control.
  • the finite control set model predictive control directly utilizes the discrete characteristics of the converter and the limited switching state. It has the advantages of no modulator, fast dynamic response and easy increase of nonlinear constraints. However, the switching frequency is not fixed, which makes the filter design difficult and control performance. Subject to certain restrictions. How to combine the advantages of the two control methods to achieve reasonable switching of the two control strategies has great practical significance.
  • the present invention aims to provide a switching type control method based on dual loop predictive control.
  • the method is based on double-loop predictive control.
  • the outer loop control adopts deadbeat control to improve the overall performance of the system.
  • the inner loop control adopts a switching control method: when the system is in steady state, the deadbeat control is used to ensure the stability of the system. State accuracy and achieve a fixed switching frequency, when the system is in a transient state, switching to a finite control set model predictive control ensures that the system transitions quickly to steady state.
  • the object of the present invention can be achieved by one of the following technical solutions.
  • a switching control method based on dual-loop predictive control the main steps are as follows:
  • (S5) According to the judgment condition of (S4), when the system is judged to be in a steady state, the inner loop adopts deadbeat control; when the judgment system is in a transient state, the inner loop adopts a finite control set model predictive control.
  • V 0 (k+1), i L (k+1) respectively represent the output voltage value and the filtered inductor current value at the k+1 sampling time as the state variables of the system;
  • V 0 (k), i L (k) The output voltage value and the filter inductor current value of the inverter k sampling time are respectively indicated as the state variables of the system;
  • V in (k) represents the AC side voltage value of the inverter bridge at the time of k sampling, as the disturbance variable of the system;
  • i 0 ( k) represents the output current value at the time of k sampling as the disturbance variable of the system;
  • y c (k) represents the value of the controlled output variable at the time of k sampling;
  • L and C respectively represent the filter inductance value of the single-phase voltage type inverter and Filter capacitor value;
  • T is the sampling period of the system.
  • the outer loop control adopts the deadbeat control to improve the overall performance of the system;
  • a switching function J sw (k)
  • , J sw (k) represents the k-sampling time switching function value
  • V 0 (k) represents the output voltage value at the k sampling time
  • V r (k) represents the reference corresponding to the k sampling time V 0 (k) Voltage value
  • V 0 (k-1) represents the output voltage value at the k-1 sampling time
  • V r (k-1) represents the reference voltage value corresponding to the k-1 sampling time V 0 (k-1);
  • the point threshold is e max , which compares J sw (k) with e max ; when J sw (k)>e max , the system is judged to be in a transient state; when J sw (k) ⁇ e max , the system is judged to be in a steady state.
  • the inner loop control adopts the deadbeat control to reduce the steady state error of the system and realize the fixed switching frequency; In the transient state, the inner loop control uses a finite control set model predictive control to ensure that the system transitions quickly to steady state.
  • the outer ring adopts the deadbeat control to greatly improve the overall performance of the system
  • the fixed switching frequency can be achieved when the system is in steady state.
  • FIG. 1 is a schematic diagram of a switching type control method based on dual loop predictive control according to the present invention.
  • FIG. 2 is a rendering diagram of a steady state output voltage waveform simulated by MATLAB to which the present invention is applied.
  • Fig. 3 is a diagram showing the effect of the steady state output voltage THD of the MATLAB to which the present invention is applied.
  • FIG. 4 is a diagram showing the effect of output voltage dynamic response when the MATLAB simulation reference voltage is applied to the present invention.
  • FIG. 1 a schematic diagram of a switching type control method based on dual-loop predictive control
  • d[]/dt represents the differential value of the state variable
  • L and C respectively represent the filter inductance value and the filter capacitance value of the single-phase voltage type inverter
  • V 0 (k), i L (k) respectively represent the inverse
  • the output voltage value and the filter inductor current value of the transformer k sampling time are used as the state variables of the system
  • V in (k) represents the AC side voltage value of the inverter bridge at the time of k sampling, as the disturbance variable of the system
  • i 0 (k) represents The output current value at the time of k sampling is used as the disturbance variable of the system
  • y c (k) represents the value of the controlled output variable at the time of k sampling.
  • V 0 (k+1), i L (k+1) represent the output voltage value and the filtered inductor current value at the time of k+1 sampling, respectively, as the state variables of the system;
  • V 0 (k), i L ( k) respectively represent the output voltage value and the filter inductor current value of the sampling time of the inverter k as the state variable of the system;
  • V in (k) represents the voltage value of the AC side of the inverter bridge at the time of k sampling, as the disturbance variable of the system;
  • i 0 (k) represents the output current value at the k sampling time as the system's interference variable;
  • y c (k) represents the controlled output variable value at the k sampling time;
  • L and C respectively represent the filter inductance of the single-phase voltage type inverter Value and filter capacitor value;
  • T is the sampling period of the system.
  • V r (k+1)-V 0 (k+1) 0 (4)
  • V 0 (k+1) represents the output voltage value at the k+1 sampling instant
  • V r (k+1) represents the reference voltage value corresponding to the k+1 sampling instant V 0 (k+1).
  • control rate of the outer loop control can be obtained by combining equations (3) and (4):
  • i r (k) represents the reference current value of the filtered inductor current i L (k) at k sampling time
  • V r (k+1) represents the reference voltage value corresponding to the k+1 sampling instant
  • V 0 (k) represents the output voltage value at the k sampling time
  • i 0 (k) represents the output current value at the k sampling time
  • C represents the filter capacitance value
  • T represents the sampling period of the system.
  • the current value calculated by the equation (5) is input as a reference signal to the inner loop controller.
  • J sw (k)
  • J sw (k) represents the switching function value at the k sampling time
  • V 0 (k) represents the output voltage value at the k sampling time
  • V r (k) represents the reference voltage value corresponding to the k sampling time
  • V 0 (k-1) represents the output voltage value at the k-1 sampling time
  • V r (k-1) represents the reference voltage value corresponding to the k-1 sampling instant V 0 (k-1).
  • V in (k) has another expression:
  • V in (k) d * V dc (8)
  • V in (k) represents the AC bridge voltage value of the inverter bridge of the inverter at the time of k sampling
  • V dc represents the DC side voltage value
  • d * represents the modulation signal.
  • i L (k+1) represents the filtered inductor current value at the k+1 sampling instant
  • i r (k+1) represents the reference current value corresponding to the k+1 sampling instant i L (k+1).
  • the modulation signal is input to the modulation unit for comparison with the triangular wave, and the switch combination is calculated and applied to the inverter to control the output voltage.
  • Power electronic converters achieve control objectives by controlling the turn-on and turn-off of controllable switching transistors, and each switch has only two states: on and off, and all switching functions are combined by these two states. Define the switch status as follows:
  • V in (k) The relationship between the alternative voltage vector V in (k) and the switch combination is shown in Table 1 (V dc is the DC side voltage value of the inverter).
  • i r (k+1) is the reference current value at time k+1
  • i p (k+1) represents the predicted current value at time k+1
  • J represents the objective function value
  • FIG. 2 is a rendering diagram of a steady state output voltage waveform simulated by the MATLAB to which the present invention is applied (the abscissa indicates time and the ordinate indicates output voltage value).
  • Fig. 3 is a diagram showing the steady state output voltage THD effect of the MATLAB to which the present invention is applied (the abscissa indicates the frequency value, and the ordinate indicates the voltage value at the corresponding frequency value after Fourier decomposition).
  • FIG. 4 is a diagram showing the dynamic response of the output voltage when the MATLAB simulation reference voltage is applied by the present invention (the abscissa indicates time and the ordinate indicates the output voltage value; the solid curve and the dashed curve represent the reference voltage and the actual output voltage, respectively, except for the middle small Part, the rest are very close).
  • the output voltage waveform is good at steady state, and the voltage distortion rate is low.
  • the finite control set model predictive control makes the system transition to a steady state more quickly, and has good dynamic performance.

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Abstract

一种基于双环预测控制的切换型控制方法，属于电力电子变流技术、工业控制领域。该方法以双环预测控制为基础，外环控制采用无差拍控制。内环控制采用一种切换型控制方法，其中，当系统处于稳态时采用无差拍控制保证系统的稳态精度并实现固定开关频率；当系统处于暂态时切换至有限控制集模型预测控制保证系统快速过渡至稳态。该方法兼备无差拍和有限控制集模型预测两种控制的各自优点，而且外环采用无差拍控制可以提升系统的总体性能，满足电力电子系统的工作要求。

Description

一种基于双环预测控制的切换型控制方法 技术领域
本发明涉及电力电子变流和工业控制技术，尤其涉及一种基于双环预测控制的切换型控制方法，属于电力电子变流技术领域。
背景技术
随着碳化硅等宽禁带功率器件的逐步成熟和推广应用，功率变换器的开关频率在不断提升，与之对应的控制方法成为研究主流。传统的双环控制中，外环主要有比例积分(Proportional-Integral,PI)控制和滑膜控制，但是其动态性能不高。系统(系统是指控制方法的实施对象，一般是电力电子变换器，在本发明当中指的是单相电压型逆变器)的性能与外环控制和内环控制都有紧密联系，如何通过外环控制提升系统的整体性能是值得深入研究的方面。
目前常用的预测控制方法是无差拍控制和有限控制集模型预测控制。无差拍控制可以实现被控量在一个控制周期内对参考量的跟踪，具有优良的稳态性能，可以实现固定的开关频率，但是调制单元的存在使无差拍控制的动态性能受限。有限控制集模型预测控制直接利用变换器的离散特性和开关状态有限的特性，具有无需调制器，动态响应快和易增加非线性约束等优点，但开关频率不固定造成滤波器设计困难，控制性能受到了一定的限制。如何将两种控制方法的优点结合在一起，实现两种控制策略的合理切换，具有很大的现实意义。
发明内容
针对现有控制策略的不足，本发明目的在于提供一种基于双环预测控制 的切换型控制方法。该方法以双环预测控制为基础，外环控制采用无差拍控制，提升系统的总体性能；内环控制采用一种切换型控制方法：当系统处于稳态时采用无差拍控制保证系统的稳态精度并实现固定开关频率，当系统处于暂态时切换至有限控制集模型预测控制保证系统快速过渡至稳态。
本发明的目的可以通过以下技术方案之一来实现。
一种基于双环预测控制的切换型控制方法，主要步骤如下：
(S1)列出系统(本发明所述系统是指控制方法的实施对象，一般是电力电子变换器)离散时刻的状态方程，并进行离散化；
(S2)测量系统的状态变量，控制输入变量，被控输出变量和干扰变量；
(S3)电压外环控制采用无差拍控制，计算电流内环参考值；
(S4)定义一种切换函数，判断系统所处状态；
(S5)根据(S4)的判断条件，当判断系统处于稳态时，内环采用无差拍控制；当判断系统处于暂态时，内环采用有限控制集模型预测控制。
进一步地，在(S1)中，根据基尔霍夫电压定律(Kirchhoff Voltage Law,KVL)和基尔霍夫电流定律(Kirchhoff Current Law,KCL)列出系统离散时刻的状态方程：
Figure PCTCN2017120375-appb-000001
d[]/dt表示状态变量的微分值；L、C分别表示单相电压型逆变器的滤波电感值和滤波电容值；V 0(k)、i L(k)分别表示逆变器k采样时刻的输出电压值和滤波电感电流值，作为系统的状态变量；V in(k)表示k采样时刻逆变桥交流侧电压值，作为系统的干扰变 量；i 0(k)表示k采样时刻的输出电流值,作为系统的干扰变量；y c(k)表示k采样时刻的被控输出变量值；设定系统的采样周期为T，根据前进欧拉法对该状态方程进行离散化，可得：
Figure PCTCN2017120375-appb-000002
(V 0(k+1)、i L(k+1)分别表示k+1采样时刻的输出电压值和滤波电感电流值，作为系统的状态变量；V 0(k)、i L(k)分别表示逆变器k采样时刻的输出电压值和滤波电感电流值，作为系统的状态变量；V in(k)表示k采样时刻逆变桥交流侧电压值，作为系统的干扰变量；i 0(k)表示k采样时刻的输出电流值,作为系统的干扰变量；y c(k)表示k采样时刻的被控输出变量值；L、C分别表示单相电压型逆变器的滤波电感值和滤波电容值；T是系统的采样周期。
进一步地，在(S2)中，测量系统的状态变量V 0(k)、i L(k)，控制输入变量V r(k)，被控输出变量y c(k)和干扰变量V in(k)、i 0(k)。
进一步地，在(S3)中，外环控制采用无差拍控制提升系统的总体性能；根据(S1)得到系统离散KCL方程：
Figure PCTCN2017120375-appb-000003
结合无差拍控制原理：V r(k+1)-V 0(k+1)＝0，V 0(k+1)表示k+1采样时刻的输出电压值；V r(k+1)表示k+1采样时刻V 0(k+1)对应的参考电压值，可得电流内环控制参考值的计算式：
Figure PCTCN2017120375-appb-000004
i r(k)表示k采样时刻滤波电感电流i L(k)的参考电流值；V r(k+1)表示k+1采样时刻V 0(k+1)对应的参考电压值；V 0(k)表示k采样时刻的输出电压值；i 0(k)表示k采样时刻的输出电流值；C表示滤波电 容值；T表示系统的采样周期。
进一步地，在(S4)中，定义一种切换函数J sw(k)＝|(V r(k)-V 0(k)) 2-(V r(k-1)-V 0(k-1)) 2|，J sw(k)表示k采样时刻切换函数值；V 0(k)表示k采样时刻的输出电压值；V r(k)表示k采样时刻V 0(k)对应的参考电压值；V 0(k-1)表示k-1采样时刻的输出电压值；V r(k-1)表示k-1采样时刻V 0(k-1)对应的的参考电压值；设切换点阈值为e max，将J sw(k)与e max进行对比；当J sw(k)>e max，判断系统处于暂态；当J sw(k)<e max，判断系统处于稳态。
进一步地，在(S5)中，根据(S4)的判断条件，当判断系统处于稳态时，内环控制采用无差拍控制，减小系统的稳态误差并实现固定开关频率；当判断系统处于暂态时，内环控制采用有限控制集模型预测控制保证系统快速过渡至稳态。
与现有技术相比，本发明的有益效果是：
1.外环采用无差拍控制大大提升系统的整体性能；
2.当系统处于稳态时，可以减少系统的稳态误差；
3.当系统处于暂态时，可以保证系统可以从暂态快速过渡至稳态；
4.系统处于稳态时可以实现固定开关频率。
附图说明
图1是本发明的一种基于双环预测控制的切换型控制方法示意图。
图2是应用本发明的MATLAB仿真稳态输出电压波形效果图。
图3是应用本发明的MATLAB仿真稳态输出电压THD效果图。
图4是应用本发明的MATLAB仿真参考电压变化时输出电压动态响应效果图。
具体实施方式
以下结合附图和实例对本发明的实施作进一步说明，但本发明的实施和保护不限于此。
如图1所示，一种基于双环预测控制的切换型控制方法示意图，
下面以单相电压型逆变器为例进行说明。其具体步骤如下。
(S1)根据基尔霍夫电压定律(Kirchhoff Voltage Law,KVL)和基尔霍夫电流定律(Kirchhoff Current Law,KCL)列出系统(本发明所述系统是指控制方法的实施对象，一般是电力电子变换器，在本实例当中指的是单相电压型逆变器)离散时刻的状态方程：
Figure PCTCN2017120375-appb-000005
式中：d[]/dt表示状态变量的微分值；L、C分别表示单相电压型逆变器的滤波电感值和滤波电容值；V 0(k)、i L(k)分别表示逆变器k采样时刻的输出电压值和滤波电感电流值，作为系统的状态变量；V in(k)表示k采样时刻逆变桥交流侧电压值，作为系统的干扰变量；i 0(k)表示k采样时刻的输出电流值,作为系统的干扰变量；y c(k)表示k采样时刻的被控输出变量值。
设系统的采样周期为T，根据前进欧拉法，对状态方程中微分项进行离散化，得系统离散化的状态方程：
Figure PCTCN2017120375-appb-000006
式中：V 0(k+1)、i L(k+1)分别表示k+1采样时刻的输出电压值和滤波电感电流值，作为系统的状态变量；V 0(k)、i L(k)分别表示逆变器k采样时刻的输出电压值和滤波电感电流值，作为系统的状态变量；V in(k)表示k采样时刻逆变桥交流侧电压值，作为系统的干扰变量；i 0(k)表示k采样时刻的输出电流值,作为系统的干扰变量；y c(k)表示k采样时刻的被控输出变量值；L、C分别表示单相电压型逆变器的滤波电感值和滤波电容值；T是系统的采样周期。
(S2)测量系统的状态变量V 0(k)、i L(k)，控制输入变量V r(k)，被控输出变量y c(k)和干扰变量V in(k)、i 0(k)。
(S3)为提高系统的整体性能，外环采用无差拍控制。根据(S1)的离散化状态方程，可以得到系统的KCL方程：
Figure PCTCN2017120375-appb-000007
根据无差拍控制原理：
V r(k+1)-V 0(k+1)＝0                 (4)
式中：V 0(k+1)表示k+1采样时刻的输出电压值；V r(k+1)表示k+1采样时刻V 0(k+1)对应的参考电压值。
结合式(3)和(4)可以得到外环控制的控制率：
Figure PCTCN2017120375-appb-000008
式中：i r(k)表示k采样时刻滤波电感电流i L(k)的参考电流值；V r(k+1)表示k+1 采样时刻V 0(k+1)对应的参考电压值；V 0(k)表示k采样时刻的输出电压值；i 0(k)表示k采样时刻的输出电流值；C表示滤波电容值；T表示系统的采样周期。
将式(5)计算得到的电流值作为参考信号输入至内环控制器。
(S4)为增强切换点的灵敏性和有效性，我们定义以下切换函数：
J sw(k)＝|(V r(k)-V 0(k)) 2-(V r(k-1)-V 0(k-1)) 2|         (6)
式中：J sw(k)表示k采样时刻的切换函数值；V 0(k)表示k采样时刻的输出电压值；V r(k)表示k采样时刻V 0(k)对应的参考电压值；V 0(k-1)表示k-1采样时刻的输出电压值；V r(k-1)表示k-1采样时刻V 0(k-1)对应的的参考电压值。
根据系统的要求，设切换点阈值为e max，将J sw(k)与e max进行对比。当J sw(k)>e max，判断系统处于暂态；当J sw(k)<e max，判断系统处于稳态。
(S5)根据(S4)的判断结果，选择适用不同情况的内环控制方法。
当J sw(k)<e max，判断系统已经进入稳态，电流内环控制采用无差拍控制。此时，根据式(2)可以得到系统离散形式KVL方程：
Figure PCTCN2017120375-appb-000009
V in(k)具有另外一种表达式：
V in(k)＝d *V dc                   (8)
式中：V in(k)表示k采样时刻逆变器的逆变桥交流侧电压值；V dc表示直流侧电压值；d *表示调制信号。
根据无差拍控制控制原理：
i r(k+1)-i L(k+1)＝0                  (9)
式中：i L(k+1)表示k+1采样时刻的滤波电感电流值；i r(k+1)表示k+1采样时刻i L(k+1)对应的参考电流值。
结合式(7)、(8)和(9)可以得到电流内环控制的控制率，即得调制信号：
Figure PCTCN2017120375-appb-000010
将调制信号输入至调制单元与三角波进行比较，计算开关组合并作用于逆变器，进而对输出电压进行控制。
当J sw(k)>e max时，判断系统处于暂态，此时电流内环控制切换至有限控制集模型预测控制。此时，系统离散化KVL方程如式(7)所示，改写式(7)可以得到预测电流计算式：
Figure PCTCN2017120375-appb-000011
电力电子变换器通过控制可控开关管的开通和关断来实现控制目标，而每个开关只有两个状态：开和关，所有的开关函数都由这两种状态组合。定义开关状态如下：
Figure PCTCN2017120375-appb-000012
备选电压矢量V in(k)和开关组合的关系如表1所示(V dc是逆变器的直流侧电压值)。
表1 备选电压V in(k)与开关组合的关系
Figure PCTCN2017120375-appb-000013
将备选电压矢量带入式(11)，得到不同的预测电流值。定义一个目标函数如下：
J＝|i r(k+1)-i p(k+1)|                 (13)
式中：i r(k+1)是k+1时刻的参考电流值，i p(k+1)表示k+1时刻的预测电流值；J表示目标函数值。
将不同的预测电流值代入式(13)，把使J值最小所对应的开关组合作用于逆变器。
如图2、图3和图4所示，是应用本发明的MATLAB仿真效果图。图2是应用本发明的MATLAB仿真稳态输出电压波形效果图(横坐标表示时间，纵坐标表示输出电压值)。图3是应用本发明的MATLAB仿真稳态输出电压THD效果图(横坐标表示频率值，纵坐标表示傅立叶分解后对应频率值下的电压值)。图4是应用本发明的MATLAB仿真参考电压变化时输出电压动态响应效果图(横坐标表示时间，纵坐标表示输出电压值；实线曲线和虚线曲线分别代表参考电压和实际输出电压，除中间小部分，其余都是非常接近)。
具体仿真参数如表2所示。
表2 仿真参数
Figure PCTCN2017120375-appb-000014
将上述算法通过C语言编写到MATLAB的FUNTION模块，将采样的变量值输入到FUNTION模块，经过计算输出当前时刻的开关组合并作用于开关变换器。
如图2和图3所示，稳态时输出电压波形好，电压畸变率低。根据图4描述，当参考电压值发生变化时，有限控制集模型预测控制使系统更快过渡至稳态，具有很好的动态性能。
本领域技术人员可以在不违背本发明的原理和实质的前提下对本具体实施例做出各种修改或补充或者采用类似的方式替代，但是这些改动均落入本 发明的保护范围。因此本发明技术范围不局限于上述实施例。

Claims (6)

  1. 一种基于双环预测控制的切换型控制方法，其特征在于，包括如下步骤：(S1)列出系统离散时刻的状态方程，并进行离散化；(S2)测量系统的状态变量，控制输入变量，被控输出变量和干扰变量；(S3)电压外环控制采用无差拍控制，计算电流内环参考值；(S4)定义一种切换函数，判断系统所处状态；(S5)根据(S4)的判断条件，当判断系统处于稳态时，内环采用无差拍控制；当判断系统处于暂态时，内环采用有限控制集模型预测控制。
  2. 根据权利要求1所述的一种基于双环预测控制的切换型控制方法，其特征在于：在(S1)中，根据基尔霍夫电压定律(Kirchhoff Voltage Law,KVL)和基尔霍夫电流定律(Kirchhoff Current Law,KCL)列出系统离散时刻的状态方程：
    Figure PCTCN2017120375-appb-100001
    d[]/dt表示状态变量的微分值；L、C分别表示单相电压型逆变器的滤波电感值和滤波电容值；V 0(k)、i L(k)分别表示逆变器k采样时刻的输出电压值和滤波电感电流值，作为系统的状态变量；V in(k)表示k采样时刻逆变桥交流侧电压值，作为系统的干扰变量；i 0(k)表示k采样时刻的输出电流值,作为系统的干扰变量；y c(k)表示k采样时刻的被控输出变量值；设定系统的采样周期为T，根据前进欧拉法对该状态方程进行离散化，可得：
    Figure PCTCN2017120375-appb-100002
    V 0(k+1)、i L(k+1)分别表示k+1采样时刻的输出电压值和滤波电感电流值，作为系统的状态变量；V 0(k)、i L(k)分别表示逆变器k采样时刻的输出电压值和滤波电感电流值，作为系统的状态变量；V in(k)表示k采样时刻逆变桥交流侧电压值，作为系统的干扰变量；i 0(k)表示k采样时刻的输出电流值,作为系统的干扰变量；y c(k)表示k采样时刻的被控输出变量值；L、C分别表示单相电压型逆变器的滤波电感值和滤波电容值；T是系统的采样周期。
  3. 根据权利要求1所述的一种基于双环预测控制的切换型控制方法，其特征在于：在(S2)中，测量系统的状态变量V 0(k)、i L(k)，控制输入变量V r(k)，被控输出变量y c(k)和干扰变量V in(k)、i 0(k)。
  4. 根据权利要求1所述的一种基于双环预测控制的切换型控制方法，其特征在于：在(S3)中，外环控制采用无差拍控制提升系统的总体性能；根据(S1)得到系统离散KCL方程：
    Figure PCTCN2017120375-appb-100003
    结合无差拍控制原理：V r(k+1)-V 0(k+1)＝0，V 0(k+1)表示k+1采样时刻的输出电压值；V r(k+1)表示k+1采样时刻V 0(k+1)对应的参考电压值，可得电流内环控制参考值的计算式：
    Figure PCTCN2017120375-appb-100004
    i r(k)表示k采样时刻滤波电感电流i L(k)的参考电流值；V r(k+1)表示k+1采样时刻V 0(k+1)对应的参考电压值；V 0(k)表示k采样时刻的输出电压值；i 0(k)表示k采样时刻的输出电流值；C表示滤波电容值；T表示系统的采样周期。
  5. 根据权利要求1所述的一种基于双环预测控制的切换型控制方法，其特征在于：在(S4)中，定义一种切换函数J sw(k)＝|(V r(k)-V 0(k)) 2-(V r(k-1)-V 0(k-1)) 2|，J sw(k)表示k采样时刻切换函数值；V 0(k)表示k采样时刻的输出电压值；V r(k)表示k采样时刻V 0(k)对应的参考电压值；V 0(k-1)表示k-1采样时刻的输出电压值；V r(k-1)表示k-1采样时刻V 0(k-1)对应的的参考电压值；设切换点阈值为e max，将J sw(k)与e max进行对比；当J sw(k)>e max，判断系统处于暂态；当J sw(k)<e max，判断系统处于稳态。
  6. 根据权利要求1所述的一种基于双环预测控制的切换型控制方法，其特征在于：在(S5)中，根据(S4)的判断条件，当判断系统处于稳态时，内环控制采用无差拍控制，减小系统的稳态误差并实现固定开关频率；当判断系统处于暂态时，内环控制采用有限控制集模型预测控制保证系统快速过渡至稳态。
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