WO2018087960A1 - Dispositif d'alimentation électrique à compensation de facteur de puissance et dispositif d'éclairage à del - Google Patents

Dispositif d'alimentation électrique à compensation de facteur de puissance et dispositif d'éclairage à del Download PDF

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Publication number
WO2018087960A1
WO2018087960A1 PCT/JP2017/025760 JP2017025760W WO2018087960A1 WO 2018087960 A1 WO2018087960 A1 WO 2018087960A1 JP 2017025760 W JP2017025760 W JP 2017025760W WO 2018087960 A1 WO2018087960 A1 WO 2018087960A1
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Prior art keywords
power supply
input voltage
input
phase
current
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PCT/JP2017/025760
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English (en)
Japanese (ja)
Inventor
義章 石黒
章太 渡辺
友一 坂下
前田　貴史
陽 山上
達也 平山
Original Assignee
三菱電機株式会社
三菱電機照明株式会社
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Application filed by 三菱電機株式会社, 三菱電機照明株式会社 filed Critical 三菱電機株式会社
Priority to JP2018550024A priority Critical patent/JP6599024B2/ja
Publication of WO2018087960A1 publication Critical patent/WO2018087960A1/fr

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only

Definitions

  • the present invention relates to a power factor compensation power supply device and an LED lighting device that have a power factor correction (PFC) function and convert AC power into DC power.
  • PFC power factor correction
  • a power factor correction circuit comprising a DC / DC converter connected at the subsequent stage of a full-wave rectifier circuit is detected, and phase information of the input voltage after full-wave rectification is detected with high accuracy.
  • a switching power supply device has been proposed in which the ON width of the switching elements constituting the DC / DC converter is widened to prevent the power factor from being lowered when it is low (see, for example, Patent Document 1 below).
  • an input filter is provided for the purpose of suppressing switching noise generated by switching on / off of the switching element to the power system on the input side. Due to the influence of the above, there is a problem that a phase difference which becomes a leading phase with respect to the input voltage occurs in the input current, and the power factor deteriorates.
  • the present invention has been made to solve the above-described problem, and is a power factor compensating power source capable of achieving a high power factor by controlling the phase difference between the input current and the input voltage to be zero.
  • An object is to provide a device and an LED lighting device.
  • a power factor compensation power supply device includes a power supply main circuit unit and a power supply control unit that controls the power supply main circuit unit,
  • the power supply main circuit section includes a full-wave rectifier circuit that full-wave rectifies an AC voltage of an AC power supply, an inductance element, and a switching element, and an output voltage that targets an input voltage obtained by the full-wave rectifier circuit.
  • An input filter configured to include an input capacitor for suppressing outflow of a noise component due to the switching element to the AC power supply, an input voltage detection unit that detects the input voltage, and the output An output voltage detection unit for detecting the voltage
  • the power supply control unit adjusts the phase of the input voltage from the signal detected by the input voltage detection unit with respect to a reference on-time of the switching element determined based on the signal detected by the output voltage detection unit.
  • the output voltage is controlled to a desired voltage by turning on and off the switching element using the on-time of the switching element that is detected and corrected by multiplying the correction signal generated based on the phase.
  • power factor compensation control is performed on the input current from the AC power source.
  • the LED lighting device includes a power supply main circuit unit and a power supply control unit that controls the power supply main circuit unit, and an LED module is connected to an output side of the power supply main circuit unit,
  • the power supply main circuit section includes a full-wave rectifier circuit that full-wave rectifies an AC voltage of an AC power supply, an inductance element, and a switching element, and an output current that targets an input current obtained by the full-wave rectifier circuit.
  • An input filter configured to include an input capacitor for suppressing the outflow of noise components from the switching element to the AC power supply, and an input for detecting an input voltage obtained by the full-wave rectifier circuit A voltage detection unit, and an output current detection unit for detecting the output current
  • the power supply control unit adjusts the phase of the input voltage from the signal detected by the input voltage detection unit with respect to the reference on-time of the switching element determined based on the signal detected by the output current detection unit.
  • the output current is controlled to a desired current by turning on and off the switching element using the on-time of the switching element that is detected and corrected by multiplying the correction signal generated based on the phase.
  • power factor compensation control is performed on the input current from the AC power source.
  • the LED lighting device includes the power factor compensation power supply device having the above-described configuration, and an LED module is connected to the output side of the power supply main circuit unit, and between the converter and the LED module. Is provided with an LED current adjusting circuit for adjusting the current flowing through the LED module to a target current.
  • the phase information of the input voltage is detected, and the ON width of the switching element is adjusted as needed with the correction amount corresponding to the detected input voltage phase. Correction is performed to thereby bring the phase difference between the input current and the input voltage close to zero, so that a high power factor can be achieved.
  • FIG. 1 It is a block diagram which shows the structure of the power factor compensation power supply apparatus in Embodiment 1 of this invention. It is explanatory drawing regarding operation
  • Embodiment 2 of this invention it is explanatory drawing regarding the corrected amount about the case where an alternating current input has a phase advance with respect to an alternating current input voltage, and a case where there exists a phase delay. It is explanatory drawing which shows the change tendency of the correction amount with respect to the input voltage phase for introduction of the approximate expression which calculates correction amount in Embodiment 2 of this invention.
  • FIG. 5 is an explanatory diagram of how to determine a phase reference related to a pulsating voltage obtained by a full-wave rectifier circuit in the first to third embodiments of the present invention.
  • FIG. 1 is a block diagram showing a configuration of a power factor compensating power supply apparatus according to Embodiment 1 of the present invention.
  • a power factor compensation power supply device 1 includes a power supply main circuit unit 2 and a power supply control unit 3.
  • the power supply main circuit unit 2 is mainly composed of a full-wave rectifier circuit 14, an input filter 21, a DC / DC converter (hereinafter simply referred to as a converter) 4, and an output capacitor 15.
  • the full-wave rectifier circuit 14 is configured with a diode bridge to obtain a full-wave rectified voltage
  • the input filter 21 is for suppressing conduction noise generated by switching to the AC power supply 5 by a switching element 7 described later, and includes an input coil 20 and input capacitors 6 and 6a.
  • the converter 4 adjusts the pulsating voltage
  • the output capacitor 15 is for smoothing the pulsation of the output voltage of the converter 4 to obtain a DC output voltage vo.
  • a load 8 is connected to the DC power output side of the power supply main circuit unit 2.
  • the power main circuit unit 2 is provided with a zero current detection unit 16, an input voltage detection unit 10, and an output voltage detection unit 9.
  • the input voltage detection unit 10 detects the magnitude of the pulsating voltage
  • two or more divided voltages connected in series It consists of resistance.
  • the output voltage detection unit 9 detects the magnitude of the DC output voltage vo as an output voltage detection value vosen. Although not shown here, for example, two or more voltage dividing resistors connected in series are used. Consists of. The contents of current detection by the zero current detection unit 16 will be described later.
  • the converter 4 is composed of a boost chopper circuit here. That is, in the converter 4, for example, a connection point between one end of the full-wave rectifier circuit 14 and one end of the input capacitor 6 is connected to one end of the reactor 18 that is an inductance element, and the other end of the reactor 18 is connected to the anode of the diode 17.
  • the cathode of the diode 17 is connected to a connection point between one end of the output capacitor 15 and one end of the load 8.
  • the connection point between the reactor 18 and the diode 17 is connected to one end of the switching element 7 (drain terminal in the case of an FET element), and the other end (source terminal in the case of FET element) of the switching element 7 is connected to the output capacitor 15. It is connected to a connection point between the end and the other end of the load 8.
  • the switching element 7 may be an FET (Field Effect Transistor) element, an IGBT (Insulated Gate Bipolar Transistor) element, or the like that is driven by the element drive signal Vg generated by the power supply control unit 3. Further, the above diode 17 can be changed to a switching element 17 such as an FET element or an IGBT element, and a synchronous rectification system can be employed in which the switching elements 7 and 17 are operated with reverse logic.
  • FET Field Effect Transistor
  • IGBT Insulated Gate Bipolar Transistor
  • the zero current detection unit 16 is for detecting the current iL flowing through the reactor 18 in order to determine the turn-on timing of the switching element 7.
  • a current detection resistor is provided between both ends of the current detection resistor. Is detected as a voltage conversion value iLsen corresponding to the reactor current iL.
  • the zero current detection unit 16 is not limited to the method of detecting on the low potential side as shown in FIG. 1, but a voltage conversion corresponding to the reactor current iL on the high potential side by connecting a current detection resistor in series with the reactor 18. A method of detecting the value iLsen can also be adopted. In addition, an auxiliary winding having a reverse polarity may be provided in the reactor 18 without using a current detection resistor, and a voltage obtained from the auxiliary winding may be detected as the voltage conversion value iLsen.
  • the power supply control unit 3 includes an input voltage phase detection unit 12, an output voltage control unit 11, and a switch control unit 13.
  • the input voltage phase detector 12 determines the phase binphase of the pulsating voltage
  • a difference is obtained from vosen and a preset target voltage value voref, and a control calculation is performed to determine an on-time (hereinafter referred to as a reference on-time) ton1 as a reference of the switching element 7.
  • the on-time ton2 of the element drive signal Vg to be determined is determined. Further, the switch control unit 13 determines the ON timing of the element drive signal Vg according to the voltage conversion value iLsen corresponding to the reactor current iL detected by the zero current detection unit 16.
  • the input voltage phase ⁇ is expressed with reference to the bottom of the pulsating voltage
  • the power supply control unit 3 may be a general digital control circuit (including a circuit using software having the same function) that does not use an IC, and some of the components are digital control circuits. Further, an analog control circuit that does not use a digital control circuit may be used. In the first embodiment, a configuration in the case of a digital control circuit using a microcomputer will be described.
  • the output voltage control unit 11 is the frequency of the pulsating voltage
  • after full-wave rectification when the input voltage phase ⁇ ( vinphase) is 0, ⁇ , that is, when the commercial frequency is 50 Hz (cycle 20 ms).
  • the reference on-time ton1 is determined at an interval of 100 Hz (period 10 ms).
  • the converter 4 is a step-up chopper circuit
  • the change ⁇ iL of the reactor current iL flowing through the reactor 18 having the capacitance L in the on period ton of the switching element 7 is pulsed as shown in the following equation (1). Since the relationship is proportional to the current voltage
  • the reference on-time is set to ton1a for one period Ta of the pulsating voltage
  • the power factor improvement control can be performed by utilizing the property that the average current iLmean per unit time of the reactor current iL can be controlled in the same phase as the pulsating voltage
  • are detected by the input voltage phase detection unit 12, and the detection timing method will be described later.
  • the output voltage control unit 11 calculates the difference between the output voltage detection value vosen and the target voltage value voref, and determines the reference on-time ton1 for the switching element 7 so that the difference becomes zero.
  • the reference on-time ton1 is determined by classical control such as proportional-integral (PI) control and proportional-integral-derivative (PID) control, or modern control such as H ⁇ (H-infinity) control.
  • the present invention is not limited to this method.
  • the control calculation may be performed every time the switching element 7 is turned on.
  • the high-frequency component of the output voltage detection value vosen is sufficiently attenuated by adjusting an RC filter (not shown) provided in the output voltage detection unit 9, or a control coefficient (gain) used for the control calculation is sufficient.
  • the effect equivalent to the method mentioned above can be acquired by making it small.
  • the input voltage phase detector 12 detects the phase (0 to ⁇ ) of the pulsating voltage
  • the bottom detection timing is set to “zero”, the count-up process is performed for each sampling of the AD converter, etc., and the sampling frequency fsamp, the commercial frequency fcom of the AC power supply 5 and the count-up number n are used as follows.
  • the input voltage phase ⁇ is determined using the following equation (3) based on the assumption of one cycle delay that is an average delay value: Good.
  • the switch control unit 13 which is a characteristic part of this embodiment, performs the reference on-time ton1 according to the correction amount adjustment equation shown in the following equation (8) according to the input voltage phase ⁇ calculated by the above-described method.
  • a correction amount Kphase as a correction signal for is determined.
  • the on-time ton2 of the element drive signal Vg for the switching element 7 is calculated as needed by the correction amount Kphase with respect to the reference on-time ton1 obtained by the output voltage control unit 11 using the following equation (4). Is finally determined.
  • Ton2 ton1 ⁇ Kphase (4)
  • the element drive signal Vg is turned off after the switching element 7 is kept on for the on time ton2 determined by the above-described equation (4) by the element drive signal Vg.
  • the switching element 7 is turned off and the current iL flowing through the reactor 18 gradually decreases, and accordingly, the voltage conversion value iLsen detected by the zero current detection unit 16 also decreases.
  • the element drive signal Vg is switched on again, and the equation (4) The on state is maintained for the determined on time ton2.
  • the comparison of the voltage conversion value iLsen with the threshold voltage iLth uses a comparator here, but the present invention is not limited to this.
  • the on-threshold voltage of the switching element 7 is configured as a substitute for iLth, so that the timing when the voltage conversion value iLsen falls below iLth can be detected.
  • the threshold voltage iLth can be adjusted so as to perform a critical operation in consideration of an element delay of a comparator or the like used for comparison of the capacitance or voltage level included in the zero current detection unit 16 in advance. preferable.
  • FIG. 4A shows an example in which the voltage conversion value iLsen is obtained by configuring the zero current detection unit 16 with a current detection resistor as shown in FIG.
  • FIG. 4A shows an example in which the voltage conversion value iLsen is obtained by configuring the zero current detection unit 16 with a current detection resistor as shown in FIG.
  • an auxiliary winding having a reverse polarity is provided in the reactor 18 as described above without using the current detection resistor and the voltage obtained from the auxiliary winding is obtained as the voltage conversion value iLsen
  • FIG. The waveform shown in FIG. Even in this case, the on-switching timing of the element drive signal Vg for the switching element 7 can be determined by the same method from the comparison between the voltage conversion value iLsen and the threshold voltage iLth by the comparator.
  • the reference on-time ton1 is determined at an interval of one cycle of the pulsating voltage
  • a correction operation is performed by the equation (4) to turn on the on-time ton2 of the element driving signal Vg. It is preferable that the timing of determining is determined every time the element drive signal Vg is output, that is, in the output cycle of the element drive signal Vg.
  • FIG. 5A shows an equivalent circuit of the input filter 21 and the converter 4.
  • the current flowing through the converter 4 is expressed by a current source I * (hereinafter, * indicates a vector)
  • the full-wave rectifier circuit 14 is omitted for simplification
  • the input filter 21 is
  • the constituting input coil 20 is a reactor L
  • the input capacitors 6 and 6a are represented as a combined capacitance C.
  • the input coil 20 shown in FIG. 5A is a hybrid choke coil that simultaneously reduces common mode noise and normal mode noise.
  • the input coil 20 is not limited to this configuration.
  • FIG. 6A is an equivalent circuit diagram of the input coil 20 that is the hybrid choke coil shown in FIG.
  • FIG. 6B is a circuit diagram showing a configuration of an input coil 20a having a configuration different from that of the input coil 20 shown in FIG.
  • FIG. 6C is a circuit diagram illustrating a configuration of an input coil 20b having a configuration different from that of the input coil 20 illustrated in FIG.
  • the input coil 20 is not limited to the hybrid choke coil shown in the input coil 20 in FIG. 6A, and for example, a normal mode choke coil may be used as shown in the input coil 20a in FIG. 6B. Alternatively, a common mode choke coil may be used as shown in the input coil 20b of FIG.
  • FIG. 5B shows a vector diagram in the case where no control for canceling is performed.
  • Ip * indicates the effective current
  • the phase of the current I * flowing through the converter 4 is corrected (delayed) by ⁇ , thereby controlling the AC input current iac * only for the effective current Ip * component.
  • the effective current Ip * can be obtained by the following equation (6) from the law of conservation of energy.
  • the on-time ton2 of the element drive signal Vg for the switching element 7 is finally determined by performing a correction operation on the reference on-time ton1 as needed with the correction amount Kphase according to the above equation (4).
  • the reference on-time ton1 is determined by an interval of one cycle of the pulsating voltage
  • , but the correction amount Kphase is a function of the input voltage phase ⁇ ( vinphase), and therefore, the element drive
  • the determination of the on time ton2 of the signal Vg is performed every time the element drive signal Vg is output within one cycle of the pulsating voltage
  • FIG. 7 shows a simulation result of a reference example to which this embodiment is not applied
  • FIG. 8 shows a simulation result to which this embodiment is applied.
  • the ON time ton2 of the element drive signal Vg, which is a multiplication value, and the waveforms of the AC input voltage vac and the AC input current iac are shown.
  • the lower limit is set to “0”
  • the upper limit is set to “2”.
  • the waveforms of the AC input voltage vac and the AC input current iac at the bottom stage are for comparing the phase difference between them, and the AC input current iac is a value multiplied by 600 for easy comparison. Is displayed.
  • the correction amount Kphase is always “1” for convenience, and therefore, the on-time ton2 of the element drive signal Vg outputs the reference on-time ton1 as it is.
  • the on-time ton2 of the element drive signal Vg is compared to the reference on-time ton1 so as to correct the phase difference ⁇ between the AC input voltage vac and the AC input current iac. Has changed greatly.
  • the lead phase can be confirmed in the AC input current iac with respect to the AC input voltage vac in FIG. 7 showing the reference example. 8, it can be confirmed that the phase difference ⁇ between the AC input voltage vac and the AC input current iac is substantially zero.
  • phase detection means of alternating current input current iac was further added to the structure of FIG.
  • the phase difference ⁇ may be obtained from the phase of both the measured AC input voltage vac and the AC input current iac, and the phase difference ⁇ may be obtained in advance by simulation or experiment.
  • the on / off control of the switching element 7 using the on time ton2 obtained by multiplying the reference correction time Kphase by the reference on time ton1 allows the AC input to the AC input voltage vac by the input capacitors 6 and 6a constituting the input filter 21. Since the influence of the phase advance of the current iac can be suppressed, the AC input current iac and the AC input voltage vac can have the same phase and the same waveform, and the power factor can be further improved as compared with the conventional case.
  • the equation (8) for determining the correction amount Kphase includes sin calculation and division, and it is difficult to realize the equation (8) by a CPU such as a microcomputer.
  • the switch control unit 13 can approximate the calculation of the equation (8) to determine the correction amount Kphase, and is intended to facilitate the mounting of a microcomputer or the like. It is said.
  • equation (8) When the input voltage phase ⁇ is displayed as a frequency (the period of the pulsating voltage
  • a solid line indicates a power factor “1”, that is, a waveform of an ideal AC input current iacref in the case of the same phase as the AC input voltage, and a broken line indicates an ideal current waveform iacref.
  • the waveform of the AC input current iaclag delayed by 10 degrees is shown.
  • FIG. 10A shows the result of calculating the correction amount Kphase by dividing the broken line AC input current iaclead in FIG. 9A by the ideal AC input current iacref in FIG. 9A based on the equation (8).
  • FIG. 10B based on the equation (8), the correction amount Kphase was obtained by dividing the broken line AC input current iaclag in FIG. 9B by the solid line ideal AC input current iacref. Results are shown.
  • the correction amount Kphase when the correction amount Kphase indicated by each solid line is approximated to a simple linear function is indicated by the broken line in FIG.
  • the correction amount Kphase is a sawtooth signal having a period of 180 degrees by the respective linear functions shown by the broken lines in FIG. 11, and the expression (8), which requires processing time, is expressed as the following expression (9): It can be confirmed that it can be approximated to a linear function expression in which the correction amount Kphase simply increases in accordance with the input voltage phase ⁇ .
  • the slope A and intercept B of the linear function equation represented by equation (9) can be set under the following conditions.
  • the slope A is set to a “positive” value.
  • the change in the correction amount Kphase with respect to the change in the input voltage phase ⁇ is increased, that is, the slope A is set to “large”.
  • Equation (7) the condition that the phase difference ⁇ of the AC input current iac with respect to the AC input voltage vac is increased will be considered from the equation (7).
  • phase difference ⁇ increases as the output power Po decreases from FIG. 12A
  • the phase difference ⁇ increases as the AC input voltage vac increases from FIG. 12B
  • the AC input voltage frequency f from FIG. The larger the value, the larger the phase difference ⁇ .
  • phase difference ⁇ increases according to changes such that the output power Po to the load 8 decreases, the AC input voltage vac increases, and the AC input voltage frequency f increases. Accordingly, the input voltage phase is increased accordingly.
  • a change in the correction amount Kphase with respect to the change in ⁇ is set to be large, that is, the slope A of the equation (9) is set to be large.
  • correction amount Kphase may be determined by reading the values of the slope A and the intercept B determined in advance for each combination of various parameters from the table memory, and the correction amount Kphase calculated for each combination of various parameters in advance. The value may be read from the table memory.
  • a means capable of detecting the power factor and the AC input current iac is added to the configuration of FIG. 1, and the pulsating voltage
  • These values A and B can be adjusted so as to further improve the power factor by performing control to change the slope A and the intercept B for each cycle.
  • the effect was confirmed by simulation using a circuit simulation of Myway Plus.
  • FIG. 13 shows a simulation result of a reference example to which the present invention is not applied
  • FIG. 14 shows a simulation result of the second embodiment
  • FIG. 15 shows a simulation result of the first embodiment.
  • the upper stage shows the correction amount Kphase corresponding to the input voltage phase ⁇
  • the lower stage shows the waveforms of the AC input voltage vac and the AC input current iac.
  • the correction amount Kphase is set such that the lower limit is “0” and the upper limit is “2”.
  • the waveforms of the lowest AC input voltage vac and the AC input current iac are for comparing the phase difference between the two, and the AC input voltage vin is easy to compare. Indicates a value multiplied by 1/1000.
  • FIG. 16 shows a simulation result of a reference example to which the present invention is not applied
  • FIG. 17 shows a simulation result of the second embodiment
  • FIG. 18 shows a simulation result of the first embodiment.
  • the upper stage shows the correction amount Kphase corresponding to the input voltage phase ⁇
  • the lower stage shows the waveforms of the AC input voltage vac and the AC input current iac.
  • the correction amount Kphase has a lower limit set to “0” and an upper limit set to “2”.
  • the waveforms of the lowest AC input voltage vac and the AC input current iac are for comparing the phase difference between them, and the AC input voltage vin is easy to compare. Indicates a value multiplied by 1/1000.
  • the power factor is 0.673 in FIG. 16, 0.783 in FIG. 17, and 0.819 in FIG. 18.
  • the approximation formula (9) in the second embodiment It can be confirmed that there is a power factor improvement effect larger than that of the example, and that an effect of a level equivalent to that of the expression (8) shown in the first embodiment can be obtained.
  • the correction amount Kphase is approximated to a simple linear function equation of the input voltage phase ⁇ shown in the equation (9), so that the correction amount Kphase is generated as a sawtooth signal having a cycle of 180 degrees.
  • the present invention is not limited to this, and the calculation of the correction amount Kphase can be determined as another function. This will be described next.
  • FIG. 19 shows a simulation waveform when the input voltage phase ⁇ detected by the input voltage phase detector 12 is delayed by 18 degrees with respect to the actual phase of the pulsating voltage
  • the upper part shows the relationship between the pulsating voltage
  • the lower part shows the relationship between the AC input voltage vac and the AC input current iac (in order to make the waveforms easier to compare, the AC input voltage vac is displayed at 1/1000 times).
  • the detected input voltage phase ⁇ is delayed by 18 degrees with respect to the phase of the actual pulsating voltage
  • the correction amount Kphase is not a sawtooth signal, but a period of 180 degrees inclined to the left and right with respect to the peak in accordance with the change of the input voltage phase ⁇ .
  • can be suppressed, and the input voltage phase detection can be performed. It becomes possible to suppress the power factor deterioration due to the phase delay when the input voltage phase ⁇ is detected by the unit 12.
  • the method of determining the correction amount Kphase according to the input voltage phase ⁇ is not limited to a sawtooth signal, but can be given by a function that is relatively easy to perform arithmetic processing such as a triangular wave as shown in FIGS. it can.
  • the reference on-time ton1 is multiplied by a correction amount Kphase determined by a function that is relatively easy to calculate, such as a sawtooth waveform obtained by equation (9) or a triangular waveform as shown in FIG.
  • Kphase determined by a function that is relatively easy to calculate, such as a sawtooth waveform obtained by equation (9) or a triangular waveform as shown in FIG.
  • FIG. FIG. 22 is a block diagram showing the configuration of the LED lighting device according to the third embodiment, and the same reference numerals are given to the components corresponding to those in FIG.
  • the feature of the LED lighting device 27 of the third embodiment is the case where the load 8 is an LED module 22 in which a plurality of LEDs 23 are connected in cascade with respect to the configuration shown in FIG.
  • the difference from the first embodiment is that the power supply control unit 3 includes an output current detection unit 24 instead of the output voltage detection unit 9 and an output current control unit 26 instead of the output voltage control unit 11. It is changed so that the target current value ioreef is set.
  • Other configurations are the same as those shown in FIG.
  • the output current detection unit 24 installs a current detection resistor (not shown), for example, and detects a potential difference generated between both ends of the current detection resistor as a voltage conversion value iosen corresponding to the output current flowing through the LED module 22.
  • a current detection resistor not shown
  • the connection configuration of the LED modules 22 is here that all the LEDs 23 are connected in series, but a plurality of LEDs 23 may be connected in series or in series-parallel.
  • the light source is the LED 23 here, it is not limited to this and can be changed to an organic EL or a laser diode.
  • the LED 23 is usually suitable for current control because of its VI characteristics, and therefore, in FIG. 21, the control is changed from output voltage control to output current control.
  • the current flowing through the LED module 22 can be controlled by performing the same control as in the first and second embodiments.
  • the dimming function can be realized if the target current value ioref inputted from the outside is variable.
  • the output voltage vo does not change even when the current flowing through the LED module 22 is changed by the dimming function.
  • the output power Po is one parameter for determining the correction amount Kphase. Ideally, it is necessary to calculate the power by detecting both the output voltage vo and the output current io. .
  • the load 8 becomes the LED module 22 as shown in FIG. 22, so that the output current io is detected as the voltage conversion value iosen by the output current detector 24 without detecting the output voltage vo.
  • the correction amount Kphase can be adjusted.
  • the dimming signal is normally input to the microcomputer from the outside to perform dimming control, the optimum correction amount Kphase after dimming can be obtained in advance even when the dimming signal is varied. It is possible to perform power factor correction control with a fast response speed against fluctuations.
  • the LED lighting device 27 having the configuration shown in FIG. 22 converts the AC input into the DC output by the single-stage converter 4, but in this way, the commercial frequency ripple of the AC power source 5 is removed on the output side. There is a concern that the LED module 22 is visually observed as flickering of light.
  • FIG. 23 shows a configuration in the case of using a two-stage converter.
  • the LED lighting device 27 having the configuration shown in FIG. Is provided with an LED current adjustment circuit 25 (a detailed control configuration is omitted) as an output current adjustment DC / DC converter, and an output current for detecting a voltage conversion value iosen corresponding to the current flowing through the LED module 22
  • the detection unit 24 is provided, and the power supply control unit 3 is provided with an output current control unit 26 in addition to the output voltage control unit 11.
  • the output current control unit 26 uses the LED current adjustment circuit 25 so that the output current io becomes constant according to the target current value ioreef inputted from the outside and the voltage conversion value iosen detected by the output current detection unit 24.
  • the output voltage control unit 11 sets the target voltage value voref to a value that is equal to or higher than the voltage of the LED module 22, and controls the converter 4 in the same manner as described above. .
  • the output voltage vo is constant regardless of the dimming degree
  • the output voltage of the converter 4 is generally controlled to a constant value regardless of the dimming degree. Advantages similar to those of the configuration of FIG. 22 can be obtained.
  • the output current detection unit 24 is provided in the power supply main circuit unit 2 to detect the output current io as the voltage conversion value iosen.
  • the switching element 7 is on / off controlled using an on-time ton2 obtained by multiplying the amount Kphase by a reference on-time ton1.
  • the influence of the phase advance of the AC input current iac on the AC input voltage vac by the input capacitors 6 and 6a can be corrected, so that the AC input current iac and the AC input voltage vac can have the same phase and waveform, thereby improving the power factor. can do.
  • the power supply control unit 3 is preferably housed in a single control IC package.
  • the present invention is not limited to the configurations of the above-described first to third embodiments, and each of the first to third embodiments can be freely combined without departing from the spirit of the present invention.
  • the configurations of the first to third embodiments can be modified or omitted as appropriate.

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)
  • Rectifiers (AREA)
  • Circuit Arrangement For Electric Light Sources In General (AREA)

Abstract

L'invention concerne un dispositif d'alimentation électrique à compensation de facteur de puissance (1) pourvu : d'une unité de circuit principal d'alimentation électrique (2) équipée d'un convertisseur (4) comportant un élément de commutation (7) ; et d'une unité de commande d'alimentation électrique (3) servant à commander l'unité de circuit principal d'alimentation électrique (2). L'unité de commande d'alimentation électrique (3) génère une quantité de correction (Kphase) conformément à la phase détectée d'une tension d'entrée dans l'unité de circuit principal d'alimentation électrique (2), génère un temps d'activation (ton2) obtenu par correction d'un temps d'activation de référence (ton1) de l'élément de commutation (7) à l'aide de la quantité de correction (Kphase), et effectue une commande telle que la différence de phase entre un courant d'entrée et la tension d'entrée devient nulle.
PCT/JP2017/025760 2016-11-08 2017-07-14 Dispositif d'alimentation électrique à compensation de facteur de puissance et dispositif d'éclairage à del WO2018087960A1 (fr)

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WO2020012803A1 (fr) * 2018-07-13 2020-01-16 三菱重工サーマルシステムズ株式会社 Dispositif convertisseur, procédé de génération de signal de commande, et programme
JP2020096398A (ja) * 2018-12-10 2020-06-18 三菱電機株式会社 力率補償電源装置およびｌｅｄ照明装置
US11632039B2 (en) 2020-11-06 2023-04-18 Fuji Electric Co., Ltd. Integrated circuit and power supply circuit
WO2023233636A1 (fr) * 2022-06-02 2023-12-07 三菱電機株式会社 Dispositif de conversion de puissance , dispositif d'entraînement de moteur et appareil appliqué à cycle de réfrigération
JP7498060B2 (ja) 2020-08-06 2024-06-11 サンケン電気株式会社 スイッチング電源装置

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JP2005253284A (ja) * 2004-01-08 2005-09-15 Fujitsu General Ltd 電源装置
WO2010109694A1 (fr) * 2009-03-24 2010-09-30 株式会社村田製作所 Dispositif d'alimentation à découpage
JP2014110711A (ja) * 2012-12-04 2014-06-12 Renesas Electronics Corp スイッチング電源装置及び半導体装置
WO2014119040A1 (fr) * 2013-01-29 2014-08-07 三菱電機株式会社 Convertisseur de puissance
JP2015008583A (ja) * 2013-06-25 2015-01-15 シャープ株式会社 力率改善回路
JP2016093001A (ja) * 2014-11-06 2016-05-23 富士電機株式会社 交流−直流変換器の制御装置

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WO2016047207A1 (fr) * 2014-09-24 2016-03-31 三菱電機株式会社 Circuit de compensation de facteur de puissance pour dispositif de commande de bloc d'alimentation, procédé de commande pour un tel circuit, et dispositif d'éclairage à del

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JPH11299244A (ja) * 1998-04-13 1999-10-29 Toshiba Corp 電力変換装置
JP2005253284A (ja) * 2004-01-08 2005-09-15 Fujitsu General Ltd 電源装置
WO2010109694A1 (fr) * 2009-03-24 2010-09-30 株式会社村田製作所 Dispositif d'alimentation à découpage
JP2014110711A (ja) * 2012-12-04 2014-06-12 Renesas Electronics Corp スイッチング電源装置及び半導体装置
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JP2015008583A (ja) * 2013-06-25 2015-01-15 シャープ株式会社 力率改善回路
JP2016093001A (ja) * 2014-11-06 2016-05-23 富士電機株式会社 交流−直流変換器の制御装置

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Publication number Priority date Publication date Assignee Title
WO2020012803A1 (fr) * 2018-07-13 2020-01-16 三菱重工サーマルシステムズ株式会社 Dispositif convertisseur, procédé de génération de signal de commande, et programme
JP2020014273A (ja) * 2018-07-13 2020-01-23 三菱重工サーマルシステムズ株式会社 コンバータ装置、制御信号生成方法及びプログラム
JP7080120B2 (ja) 2018-07-13 2022-06-03 三菱重工サーマルシステムズ株式会社 コンバータ装置、制御信号生成方法及びプログラム
JP2020096398A (ja) * 2018-12-10 2020-06-18 三菱電機株式会社 力率補償電源装置およびｌｅｄ照明装置
JP7498060B2 (ja) 2020-08-06 2024-06-11 サンケン電気株式会社 スイッチング電源装置
US11632039B2 (en) 2020-11-06 2023-04-18 Fuji Electric Co., Ltd. Integrated circuit and power supply circuit
WO2023233636A1 (fr) * 2022-06-02 2023-12-07 三菱電機株式会社 Dispositif de conversion de puissance , dispositif d'entraînement de moteur et appareil appliqué à cycle de réfrigération

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