WO2010059927A1 - Correction de facteur de puissance (pfc) variable et commande de tension de bus connecté à un réseau - Google Patents

Correction de facteur de puissance (pfc) variable et commande de tension de bus connecté à un réseau Download PDF

Info

Publication number
WO2010059927A1
WO2010059927A1 PCT/US2009/065300 US2009065300W WO2010059927A1 WO 2010059927 A1 WO2010059927 A1 WO 2010059927A1 US 2009065300 W US2009065300 W US 2009065300W WO 2010059927 A1 WO2010059927 A1 WO 2010059927A1
Authority
WO
WIPO (PCT)
Prior art keywords
bus
voltage
frequency
scaled
compensation signal
Prior art date
Application number
PCT/US2009/065300
Other languages
English (en)
Inventor
Aaron Jungreis
Original Assignee
Flextronics Ap, Llc
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Flextronics Ap, Llc filed Critical Flextronics Ap, Llc
Priority to KR1020117011450A priority Critical patent/KR101757252B1/ko
Priority to EP09828283.3A priority patent/EP2359461B1/fr
Priority to JP2011537648A priority patent/JP5502887B2/ja
Priority to CN200980152586.0A priority patent/CN102265497B/zh
Priority to CA2743423A priority patent/CA2743423C/fr
Publication of WO2010059927A1 publication Critical patent/WO2010059927A1/fr

Links

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/425—Arrangements for improving power factor of AC input using a single converter stage both for correction of AC input power factor and generation of a high frequency AC output voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H02M1/15—Arrangements for reducing ripples from dc input or output using active elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/40—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J2300/00—Systems for supplying or distributing electric power characterised by decentralized, dispersed, or local generation
    • H02J2300/20—The dispersed energy generation being of renewable origin
    • H02J2300/22—The renewable source being solar energy
    • H02J2300/24—The renewable source being solar energy of photovoltaic origin
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H02J3/381—Dispersed generators
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E10/00—Energy generation through renewable energy sources
    • Y02E10/50—Photovoltaic [PV] energy
    • Y02E10/56—Power conversion systems, e.g. maximum power point trackers

Definitions

  • This invention relates to the generation of compensation signals for power converters including but not limited to power factor correction (PFC) converters and grid-tied inverters. More specifically, the present invention relates to methods of and apparatuses for generating a fast-loop response compensation signal that is substantially free of the second harmonic AC line-frequency ripple, typically 100 Hertz or 120Hz.
  • PFC power factor correction
  • FIG. 5 is an exemplary prior art device for generating a compensation signal for a power converter.
  • the compensation signal is for a power factor correction (PFC) converter.
  • PFC power factor correction
  • a PFC converter causes a power converter to appear as a resistive load to the network supplying power to the power converter.
  • the conversion device By having the conversion device appear as a resistive load, the power required from the network versus the power generated is optimized and thus the system is more efficient and draws less current than if the power converter presented a low-power-factor load to the network.
  • Prior art PFC regulation loops generate a slow (typically 1 OHz - 20Hz) compensation signal for the PFC converter to prevent compensation of second harmonic ripple (e.g. 100Hz or 120Hz) on the output bus.
  • second harmonic ripple e.g. 100Hz or 120Hz
  • FIG. 5 illustrates a prior art circuit 500 for generating a compensation signal 536 for a PFC converter 540.
  • a bus voltage 542 is provided to a voltage divider 520, 520' generating a scaled-bus voltage 521.
  • the scaled-bus voltage 521 is input into an inverting input 531 of an error amplifier 533.
  • a reference voltage source 534 provides a reference voltage to the non- inverting input 532 of the error amplifier 533.
  • the reference voltage source 534 is typically a constant value scaled for a desired bus voltage 542.
  • a filter compensation network 535 response time of 100ms is typical to sufficiently remove the second harmonic ripple from the scaled-bus voltage 521 and produces an error reference signal 536 for the PFC converter controller 540.
  • FIG. 6 illustrates various waveforms for a conventional prior art PFC converter system subject to a large load transient.
  • Waveform 6 A illustrates a curve of a load current as a function of time that incorporates a step in output load.
  • the load current waveform 6A corresponds to a constant power load that is stepped up at time 400ms and stepped down at time 600ms.
  • the ripple in the load current is a direct result of the variations in bus voltage and the fact that the load operates at constant power. Since PFC converters are usually connected to a second-stage converter which is a DC-to-DC converter, and the DC-to-DC converter acts as a constant power load, the curves shown in FIG. 6 correspond to a typical design situation.
  • the waveform 6B is a curve of AC input current into the conventional prior art PFC converter.
  • the envelope of the AC input current rises between 400ms and around 450ms. After removal of the AC load at 600ms, the AC input current drops back to pre-load levels after approximately 50ms.
  • the waveform 6C is a curve of the PFC output bus voltage coupled to a conventional prior art PFC converter.
  • the bus voltage drops by about 50 volts between time 400ms and 430ms. This voltage drop can result in the undesirable side effect of causing the conventional prior art PFC converter to drop out.
  • the bus voltage recovers with some overshoot as shown around time 500ms.
  • the bus-voltage overshoots by around 30 volts between the time of 600ms and 630ms.
  • the bus voltage is shown recovering back to the previous value at about 700ms.
  • the negative-going voltage excursions causes the second-stage converter to operate over a larger voltage range, thus reducing the holdup time of the power supply immediately following a load transient as well as increasing the converter cost and lowering the overall converter efficiency.
  • large electrolytic capacitors are placed on the output of the PFC stage resulting in an increase in the cost and size for the PFC converter.
  • Waveform 6D illustrates a compensation signal (536 in Fig. 5) of a prior art system configured to generate a compensation signal for a PFC converter. Because of the slow response time of the compensated error signal 536, the PFC converter 540 cannot quickly respond to the load transients ( Figure 6, curve 6A), due to the need to not respond to the line- voltage second harmonic. The compensation signal is clearly shown in FIG. 6, curve 6D. The response time of about 60 ms is required to respond to the load transient.
  • prior-art PFC converter designs have a number of design drawbacks.
  • higher voltage rated semiconductors are required because of the voltage transients and thus the PFC converter incurs a higher manufacturing cost.
  • the second-stage converter requires larger transformers to handle the higher voltage range and thus impacts the PFC converter with lower efficiency and higher cost. Additionally, hold-up times following load transients are reduced thus causing performance issues.
  • System design options are also limited because unregulated converter stages cannot be used in tandem with a PFC stage and thus limiting potential increased performance and cost savings.
  • Grid-tied inverters are another example of power converters that must regulate a DC bus while ignoring voltage variations on the bus that contain the second harmonic of the grid frequency.
  • Grid-tied inverters such as some photovoltaic inverters, have the same control and compensation issue as PFC converters.
  • FIG. 9 shows a schematic representation of a typical grid-tied inverter 900.
  • the photovoltaic array 920 or other source of power charges a high-voltage bulk capacitor 940 through a DC-to-DC converter 910 at a constant rate which is dependent on the currently available power (e.g. as a function of the level of light in the case of a photovoltaic inverter).
  • the full-bridge inverter formed by switches (Q9 - Q 12) 951-954 and inductor (L6) 955 then produces a nearly sinusoidal current in phase with the grid voltage.
  • the voltage across bulk capacitor C3 contains a significant amount of second harmonic (e.g. 100/120 Hz) ripple.
  • the inverter must adjust the amplitude of sinusoidal current that it pushes into the grid in order to regulate the voltage across bulk capacitor C3 940 to a desired level.
  • the control loop to accomplish this regulation has exactly the same issues that have already been described for generating a PFC compensation signal.
  • What is needed is a circuit that produces a compensation signal representative of the bus voltage or that is substantially free of 100/120 Hz ripple but has a fast response to transients on the PFC converter (or grid-tied inverter) bus voltage.
  • the present invention provides an apparatus for and method of generating a compensation signal that is substantially free of power source AC line-frequency harmonic ripple.
  • the compensation signal represents the deviation of the bus voltage from a target value that is substantially free of power source AC line-frequency harmonic ripple, and can be either a digital or analog signal.
  • the error-signal provides a fast loop response to the bus- voltage transients.
  • example embodiments are described herein primarily with digital processing of error- information, other implementations including analog filters are contemplated.
  • Another implementation contemplated is a digital representation of the filtered bus information. This information can be scaled bus-voltage or error-voltage information.
  • a PLL can be implemented with analog or digital circuitry
  • the stack containing bus voltage data could be implemented with an analog charge coupled device
  • the compensation module filter can be implemented with analog circuitry.
  • One aspect of the invention is a device for generating a compensation signal for a power converter.
  • the device is comprised of a frequency-locked system clock, a bus-voltage data generator, a stack of bus-voltage data, and a compensation module.
  • the frequency- locked system clock has a frequency that is a multiple of an AC line-voltage frequency.
  • the bus-voltage sampler provides bus-voltage data at the system clock frequency to the stack input.
  • the bus-voltage data can be analog or digital and can represent bus-voltage data or error data representing the difference between the bus-voltage data and a target bus voltage.
  • the stack has one or more outputs where voltage-bus data is output to the compensation module.
  • the stack has a stack length which represents one or more complete cycles of the second harmonic of the AC line-frequency.
  • the compensation module processes data from the stack forming a compensation signal output.
  • the compensation signal is formed to minimize the frequencies at twice the power source AC line-frequency through processing stack data taken at a frequency, locked to the AC line-frequency, while providing a fast response to transients on the bus-voltage.
  • the frequency-locked clock is a phase-locked-loop configured to selectively lock to twice the AC-line voltage frequency.
  • the frequency-locked clock can be configured to only lock to frequencies between 64 Hz and 140 Hz.
  • the stack can have a digital or analog implementation.
  • the analog stack comprises a charged coupled array that samples the bus-voltage data.
  • the frequency-locked system clock is an analog device and the bus error data is analog. Comparing a scaled bus-voltage against a reference signal forms the analog bus-error data.
  • the analog bus-error data is output to the stack.
  • the compensation module comprises a means for summing the bus-voltage data over one-half cycle of the AC-line voltage coupled to the stack producing a summer output, a scaled proportional component and a scaled integral component coupled to the summer output having a scaled proportional output and a scaled integral output, a scaled derivative component coupled to the stack and configured to process two bus-voltage data samples separated by one-half cycle of the AC-line voltage having a scaled proportional output, and a PID summer configured to sum the scaled proportional output, the scaled integral output, and the scaled derivative output, forming a compensation signal.
  • a PID controller not all of the P, the I, and the D signals are used in every case.
  • the compensation signal can further comprise a scaled feed-forward component having an input coupled to the power converter bus, having a feed-forward output coupled to the PID summer, wherein the compensation signal further comprises a scaled feed-forward signal component.
  • the compensation module can be embodied with digital components, analog components, or a combination thereof.
  • the device further comprises a converter module having a converter input configured to receive the compensation signal.
  • the converter module is configured to convert the digital compensation signal to an analog compensation signal on a converter output.
  • the converter output is configured to interface with a power converter.
  • the bus-voltage sampler can generate bus-voltage data or eiTor data representing the differences between the bus-voltage value and a target voltage value.
  • the data can be analog or digital.
  • the device can be implemented in an ASIC (application specific integrated circuit). Some of the larger filter components of the PLL can be implemented outside of the ASIC. Further, the ASIC can include the power converter controller as part of the device.
  • a method comprises the steps: generating a frequency- locked system clock having a frequency that is locked to an integral multiple of an AC-line voltage frequency, generating bus-voltage data at the frequency, queuing the bus-voltage data, and compensating the bus-voltage data thus forming a compensating signal wherein the compensation minimizes the compensation signal frequency components that are at even multiples of the AC-line voltage frequency.
  • the frequency-locked system clock is generated at a multiple of the bus-voltage line frequency.
  • the bus-voltage data can be generated at a scaled multiple of the system clock rate.
  • the bus-voltage data is queued in a stack at the system clock rate.
  • the stack output can be configured to output data in any manner but preferably is configured to output the most recent and the oldest bus-voltage data.
  • the stack length is selected to hold bus-voltage data representing one-half an AC line cycle, but the stack length can instead be selected to hold bus- voltage data representing an integral number of half-cycles.
  • the method includes compensating the bus-voltage data and forming a compensation signal.
  • the method of generating a frequency-locked clock is a phase- locked-loop configured to selectively lock to twice the AC-line voltage frequency.
  • the method of generating a frequency-locked clock can be configured to only lock to frequencies within 64 Hz to 140 Hz.
  • the queuing can be digital or analog.
  • the queuing can be implemented with a stack comprised of a charged coupled array that samples the bus-voltage data. Further, generating the voltage-bus data can be analog or digital and either provides sample data or error data.
  • the compensating includes a compensation module comprising a means for summing the bus-voltage data over one-half cycle of the AC-line voltage frequency coupled to the stack producing a summer output, a scaled proportional component and a scaled integral component coupled to the summer output having a scaled proportional output and a scaled integral output, a scaled derivative component coupled to the stack and configured to process two bus-voltage data samples separated by one-half cycle of the AC- line voltage having a scaled proportional output, and a PID summer configured to sum the scaled proportional output, the scaled integral output, and the scaled derivative output, forming a compensation signal.
  • a compensation module comprising a means for summing the bus-voltage data over one-half cycle of the AC-line voltage frequency coupled to the stack producing a summer output, a scaled proportional component and a scaled integral component coupled to the summer output having a scaled proportional output and a scaled integral output, a scaled derivative component coupled to the stack and configured to process two bus-volt
  • the compensation signal can further comprise a scaled feedforward component having an input coupled to the power converter bus, having a feedforward output coupled to the PID summer, wherein the compensation signal further comprises a scaled feed-forward signal component.
  • the compensation module can be embodied with digital components or with analog components.
  • the device further comprises a converter module having a converter input configured to receive the compensation signal.
  • the converter module is configured to convert the digital compensation signal to an analog compensation signal on a converter output.
  • the converter output is configured to interface with a power converter controller.
  • the generating bus-voltage data can generate bus-voltage data or error data representing the differences between the bus-voltage value and a target voltage value.
  • the data can be analog or digital.
  • the frequency-locked system clock can be analog and the bus-error data can be analog.
  • the stack of bus-error data can be an analog charge coupled device where the delay line length is one or more cycles of the bus-voltage ripple.
  • the analog filtering can form a scaled derivative component and a scaled proportional component of the bus-error data as components of the target- voltage compensation signal.
  • the filter includes a scaled integral component of the bus-error data as part of the target-voltage compensation signal.
  • a scaled feed-forward component is added to the target-voltage compensation signal.
  • a third aspect of an invention discloses method of manufacturing a device for generating a compensation signal for a power converter.
  • the method comprises the steps of providing a frequency-locked clock coupled to an AC-line having an AC-line voltage frequency, wherein the clock is configured to generate a frequency that is frequency locked to an integral multiple of the AC-line voltage frequency, providing a bus-voltage sampler coupled to the frequency-locked clock and coupled to a power converter bus having a bus- voltage and generating bus-voltage data at the frequency, providing a stack coupled to the bus-voltage sampler, wherein the stack is sized to contain bus data-samples spanning a time interval of one-half cycle of the AC-line voltage, and providing a compensation module, operatively coupled to the stack, and configured to generate from the bus-voltage data a compensation signal, and wherein the compensation signal module is configured to produce a compensation signal in which the frequency components at even multiples of AC-line voltage frequency are minimized.
  • the provided frequency-locked clock is a phase-locked-loop configured to selectively lock to twice the AC-line voltage frequency.
  • the provided frequency-locked clock can be configured to only lock to frequencies within 64 Hz to 140 Hz.
  • providing the stack can be with digital or analog components.
  • the stack can be an implemented with a charged coupled array that samples the bus- voltage data.
  • the generating the voltage-bus data can be analog or digital and either provide sample data or error data.
  • the providing the compensating module comprises a means for summing the bus-voltage data over one-half cycle of the AC-line voltage coupled to the stack producing a summer output, a scaled proportional component and a scaled integral component coupled to the summer output having a scaled proportional output and a scaled integral output, a scaled derivative component coupled to the stack and configured to process two bus-voltage data samples separated by one-half cycle of the AC-line voltage having a scaled proportional output, and a PID summer configured to sum the scaled proportional output, the scaled integral output, and the scaled derivative output, forming a compensation signal.
  • a PID controller not all of the P, the I, and the D signals are used in every case.
  • the compensation signal can further comprise a scaled feed-forward component having an input coupled to the power converter bus, having a feed-forward output coupled to the PID summer, wherein the compensation signal further comprises a scaled feed-forward signal component.
  • the compensation module can be embodied with digital components or with analog components.
  • the method further comprises providing a converter module having a converter input configured to receive the compensation signal.
  • the converter module is configured to convert the digital compensation signal to an analog compensation signal on a converter output.
  • the converter output is configured to interface with a power converter.
  • the providing a bus-voltage sampler data can generate bus- voltage data or error data representing the differences between the bus-voltage value and a target voltage value.
  • the data can be analog or digital.
  • the frequency-locked system clock can be analog and the bus-error data can be analog.
  • the stack of bus-error data can be an analog charge coupled array where the delay line length is one or more cycles of the bus-voltage ripple.
  • the analog filtering can form a scaled derivative component and a scaled proportional component of the bus-error data as components of the tar get- voltage compensation signal.
  • the filter includes a scaled integral component of the bus-error data as part of the target-voltage compensations signal.
  • a scaled feed-forward component is added to the target-voltage compensation signal.
  • FIG. 1 illustrates a block diagram of a circuit for generating a fast loop response compensation signal for a power converter that is substantially free of second harmonic AC line-frequency ripple, according to one embodiment of the invention.
  • FIG. 2 illustrates one embodiment of the block diagram of the PLL of FIG. 1.
  • FIG. 3 illustrates the method to generate a compensation signal substantially free of second harmonic AC-line frequency ripple from a power converter bus voltage having a second harmonic AC line-frequency ripple.
  • FIG. 4 illustrates the method to manufacture an apparatus to generate a fast loop response compensation signal for a power converter bus that is substantially free of second harmonic line-frequency ripple.
  • FIG. 5 illustrates the prior art for generating a slow response compensation signal for a power converter.
  • FIG. 6 shows graphs of the bus-voltage, input current, and compensation signal in response to transient load changes for a prior art system.
  • FIG. 7 shows graphs of the bus-voltage, input current, and compensation signal in response to transient load changes for a system with a fast-loop response compensation signal.
  • FIG. 8 shows graphs illustrating the differences in signals between a standard PID fiter and a modified PID' filter in response to an input change.
  • FIG. 9 illustrates a block diagram of a prior-art photovoltaic grid-tied inverter. Detailed Description of the Present Invention
  • Certain embodiments of the present invention are directed to an apparatus and method for generating a fast-loop compensation signal for a power converter, including but not limited to a PFC converter or grid-tied inverter.
  • the compensation signal is designed to have a fast response to changes in the power converter's load but not respond to the second harmonic of the AC power source frequency, typically 100/120 Hz.
  • the following description of the invention is provided as an enabling teaching of the invention that includes its best, currently known embodiment.
  • One skilled in the relevant art will recognize that many changes can be made to the embodiment described, while still obtaining the beneficial results of the present invention. It will also be apparent that some of the desired benefits of the present invention can be obtained by selecting some of the features of the present invention without utilizing other features.
  • a PLL provides a frequency- locked clock to the second harmonic of the power converter's AC power source, typically 100 Hz or 120 Hz.
  • the PLL provides a clock that is frequency-locked to the rectified AC power source line-frequency or at an integral multiple of the AC power source line-frequency.
  • the frequency-locked clock, or an integral multiple or integral fraction, or combination thereof of the frequency-locked system clock is distributed to other circuit modules.
  • a bus-voltage sampler provides information about the bus-voltage.
  • the bus-voltage sampler provides bus- voltage data representing error values proportional to the difference between the bus voltage value and a target value; however, bus-voltage data can instead represent samples of the bus-voltage values rather than error values.
  • the stack contains bus-voltage data which has been sampled over a period equal to an integral multiple of one half of the AC-line cycle.
  • the PID' signal is generated by summing three values either through analog or digital methods.
  • the modified proportional (P') signal is a scaled value of the stack sum.
  • the modified integral (F) signal is an integrated value of the stack sum scaled by a second value.
  • the modified derivative (D') signal is the difference between the oldest stack bus-voltage error data and the current bus-voltage data scaled by a third value.
  • the three scaling values are chosen to produce a fast but stable compensation loop based on techniques commonly practiced by those skilled in the art of power supply design.
  • the P', the I', and the D' signals are substantially free of second-harmonic line frequency content.
  • the P' signal was already shown to contain no second-harmonic line frequency content because the stack contains equally spaced bus-voltage data representing exactly one line-frequency half-cycle due to the frequency locked clock.
  • the F signal is an integral of the P' signal, so it too is substantially free any second-harmonic line frequency content.
  • the D' signal is the difference between two samples (oldest stack sample and current error) which are spaced apart in time by exactly one-half line cycle. In the case of steady-state converter operation, the oldest and newest bus-voltage error data are identical so the D' signal is zero, thus substantially free of second- harmonic line frequency content.
  • the resulting PID' controller therefore ignores 100/120 Hz harmonics but responds in a controlled manner to transients on the bus voltage.
  • the digital PID' target-voltage compensation signal can be directly coupled to a digital input on a digitally implemented PFC converter control, or can be converted into an analog signal and used with an analog PFC converter control.
  • FIG. 1 illustrates a block diagram 100 of one embodiment of a system for generating a fast loop response compensation signal 197 for a power converter controller 200.
  • the system comprises a second-harmonic line frequency detector 113, a phase-locked loop (PLL) 110, a bus-voltage sampler or data generator 120, a stack 140, a modified derivative data D' generator 130, a voltage sealer 170, a summer 180, a proportional data sealer 192, an integrator 191, and integrator data sealer 193, a derivative sealer 194, a feed-forward sealer 195, a compensation signal component summer 196, and an optional signal converter 160.
  • PLL phase-locked loop
  • the second-harmonic line frequency detector 113 can be any means known in the art to produce a strong signal component at twice the frequency of the AC line input. In practice, the second-harmonic line frequency detector 113 will typically be created by comparing a full-wave rectified AC line voltage to a fixed reference level; however, there are many other methods to produce pulses at a frequency equal to the second-harmonic of the input line frequency.
  • the PLL 110 is one means for producing a frequency-locked clock but other means are contemplated. The PLL 110 locks to the second harmonic of the AC-line frequency, typically at 100 Hz or 120 Hz, and generates a phase-locked system clock 112 which is a multiple of the second-harmonic line frequency.
  • the frequency multiplication is accomplished by placing a frequency divider in the feedback path of the PLL 1 10. So, for example, if the frequency divider divides the frequency by a factor of 64 and the AC line frequency is 50Hz, then the second-harmonic line frequency detector will produces pulses at a rate of 100Hz and the PLL 110 will produce a frequency-locked system clock at a frequency of 6.4kHz. One could obtain the same result by having the PLL 110 lock to the AC line frequency (rather than the second harmonic of the line frequency) and then multiply the input frequency by an additional factor of two.
  • the PLL 110 can be analog, digital, or a combination of both. Integrated ASIC version of a PLL 110 can be used. Some of the analog filter components can be too large to economically implement in an ASIC and thus kept external from a PLL ASIC design.
  • the frequency-locked system clock output 112 is coupled to the system components that require synchronization with the frequency-locked system clock which includes the stack 140, and in practice will also include all elements of the system which are implemented digitally.
  • the voltage sealer 170 typically a voltage divider composed of resistors, scales the bus voltage 171 to produce scaled bus input 124 which is input into a bus- voltage data generator 120.
  • the purpose of the sealer 170 is to lower the bus-voltage data samples to a level compatible with the other electronics.
  • the bus voltage 171 can be hundreds of volts.
  • the bus-voltage sampler 120 can generate data in a number of different forms.
  • the bus-voltage data output 121 can be digital or analog.
  • the bus- voltage data can represent a scaled bus-voltage or a scaled error value from a target voltage.
  • an error signal is generated representing the difference between a scaled bus voltage 124 and a reference signal generator 122.
  • a reference voltage 122 is coupled to the alternative input 123 into bus-voltage sampler 120.
  • the reference voltage 123 and the scaled bus voltage 124 are selected such that the difference, i.e. the error, is zero when the bus voltage 171 is at a target level.
  • the bus voltage and thus the scaled-bus voltage input 124 has 100/120Hz ripple on the signal and thus the difference between the reference voltage 122 and the scaled bus voltage 124 will reflect this ripple.
  • other electronics in the summer 180, stack 140, or in other electronics can subtract out the DC bias in the data; however, it is advantageous to use error samples rather than bus- voltage samples in order to reduce the required dynamic range of the computational hardware.
  • a typical value for voltage reference 122 is approximately 2.5 volts or 1.25 volts.
  • the bus-voltage sampler 120 provides bus-voltage data 121 to the stack 140.
  • the bus-voltage data 121 is input into the stack 140 on the input 142.
  • a frequency locked system-clock 1 12 is provided to the stack 140 from the PLL 110 to control the rate and location at which bus-voltage data is input, output, and stored in the stack 140.
  • the stack 140 is sized to hold samples representing a time interval of one-half cycle of an AC-line voltage. Multiples of this stack size are also contemplated by this disclosure.
  • the stack is digital and receives digital bus-voltage data but an analog implementation is also contemplated.
  • An analog version of the stack 140 can be implemented with chain of charged coupled devices called a charge coupled array. Samples of bus-voltage data are stored as voltage charges within the charged coupled array.
  • the summer 180 averages the bus- voltage data 121 from the stack 140 over one-half cycle of the AC-line equivalent to one cycle of the second harmonic of the AC-line. Because the bus-voltage data 121 is frequency locked to the second harmonic of the AC-line frequency, the sum of the second harmonic components will average substantially to zero when the converter has a constant load. Noise and resolution limitations will typically provide some residual data at the second harmonic.
  • the sum 180 can be calculated by either an analog means or a digital means. If the bus-voltage data 121 does not represent error values, then preferably DC offset is subtracted from each bus-voltage data 121 sample. Preferable, the sum is calculated by keeping a running sum of the samples by subtracting the oldest sample and adding the newest sample.
  • the stack 140 provides an output for the latest value, Error(n), and the error value Error(n-64) delayed by one-half cycle of the AC line or a full cycle of the AC-line second harmonic.
  • the derivative generator 130 generates a difference value D' which can be analog or digital.
  • the bus-voltage data samples on the oldest bus- voltage data output 145 are delayed by the stack length that is shown as 64 bus-voltage data samples.
  • the stack length is sized to cover a half-cycle of the AC line or a full cycle of the AC-line second-harmonic ripple.
  • the differential signal component D' is free of the AC second harmonic. This illustrates a relationship between the system clock 112 and the stack depth. A doubling of the system clock 112 will require a doubling of the stack depth.
  • FIG. 1 An implementation of a compensation signal generator utilizing a modified PID' filter is illustrated in FIG. 1.
  • the embodiment shown comprises a modified PID ! (proportional, integral, derivative) filter and can optionally include a feed-forward signal.
  • the feed-forward signal is coupled to the bus voltage 171.
  • the system disclosed generates a compensation signal that differs from prior art compensation signals.
  • the modified PID' filter utilizes AC- line frequency-locked bus-voltage data which provides the advantage of being able to remove the AC-line second harmonic from the generated compensation signal 197.
  • a half-cycle sum 180 of the bus-voltage data 121 is generated and used to generate a modified proportional signal P r , from which a modified integral signal F is generated.
  • the summer 180 can be generated by any standard means including but not limited to inputting from the stack 140 the entire AC-line half-cycle of stack bus-voltage data 121 or by computing a running sum by subtracting the oldest stack value (Data(n-64)) which is being "pushed" out of the stack and adding the newest stack value (Data(n)).
  • the modified P' value is used by the modified integrator 191 to generate the modified I' compensation signal component.
  • a modified derivative value D' is generated by inputting the current bus-voltage data (Data(n)) and the half-cycle delayed bus-voltage data (Data(n-64)) into a differential generator 130 which outputs D'.
  • the differential generator 130 can be analog or digital.
  • the P' data is scaled by Kp 192
  • the F data is scaled by K 1 193
  • the D' data is scaled by K D 194.
  • the scaled P', F, and D r values are input into a summing device 196 producing a composite compensation signal 197.
  • the compensation signal can include a feed-forward signal V feedfwd scaled by Kp 195.
  • the scaled feed-forward signal can also be input into the summer 196 forming a component of the compensation signal 197.
  • the compensation signal 197 can be input into an optional converter 160 that converts the compensation signal into a signal compatible with a power converter controller 200.
  • the converter 160 can convert a digital target bus- voltage compensation signal into an analog signal.
  • the compensation signal 197 provides a scaled fast response control signal for the power converter controller 200 that responds to a change in the bus voltage 171.
  • the compensation signal 197 is substantially free of AC 100/120 Hz line-ripple and accordingly, the power converter 201 bus voltage has better transient response characteristics by not responding to this ripple.
  • the dashed line within the block diagram represents the components that are preferably implemented in a single integrated circuit such as an ASIC (application specific integrated circuit).
  • the filter components of the PLL typically are not included in an ASIC because of the components size.
  • FIG. 2 illustrates a block diagram of the PLL 1 10 of FIG. 1 including the filter components.
  • the PLL clock generation circuit comprises a phase detector 114, notch filter components 115, a notch filter 116, a VCO (voltage controlled oscillator) 117, and clock divider 118.
  • the input to the PLL circuit 1 10 is a second harmonic input at twice the AC-line frequency.
  • the phase detector 114 generates an output signal based on the phase differences between the second harmonic pulses 113 and the output of the frequency divider 118.
  • the output of the phase detector 114 is input into a notch filter 116 with a pass-band between 64 Hz and 140 Hz.
  • the filter components 115 are coupled to the notch filter 1 16.
  • the output of the notch filter 116 is coupled to a VCO 1 17.
  • the output of the VCO 1 17 is a frequency- locked system clock 1 12.
  • the frequency-locked system clock output 1 12 is coupled to the frequency divider 118.
  • the dashed line represents the components that can be formed within the single integrated circuit.
  • the VCO has a frequency that is 64 times the output from the second-harmonic generator, or 128 times the AC-line frequency.
  • the frequency divider 118 divides the VCO 117 clock frequency down to a frequency equal to the output from the second-harmonic generator.
  • the notch filter 116 constrains the PLL 110 to only respond over a narrow frequency band corresponding to the expected range of AC line frequency.
  • the PLL 110 shifts the VCO clock frequency so that the system clock 112 is frequency locked with the AC line-frequency and thus is frequency locked with the ripple components on the bus voltage 171.
  • the PLL 110 provides a system clock output 1 12.
  • Fig. 8 illustrates the difference in signals between a true PID (proportional, integral, derivative) controller and the signals generated by the present invention PID' controller for an input ac line at 50Hz.
  • the waveform 8A shows at time 400ms the derivative response "D" to a step wise increase in the bus voltage and at about time 500ms the step decrease in bus voltage.
  • Waveform 8B shows the effect on D', the modified derivative response signal, to a step increase in bus voltage of the present invention.
  • the derivative signals D and D' have similar fast responses to the onset of the transients; however, whereas D decreases to 0 volts immediately corresponding to the end of the Vbus transient, D' does not decrease to zero until 10ms (a half-cycle for a converter operating at 50Hz) following the onset of the transient.
  • the waveform 8C shows the modified integral signal I'
  • the modified integral signal I' is similar to the true integral signal I with a slight delay of about 4ms. The delay is caused by a difference between the area under P' and the area under P at the transients.
  • modified PID' signals depart from the true PID signals; however, one can also see that the transient information shows up in P', I', and D' immediately following the onset of the transient.
  • PID' controller using the modified signals P', I', and D'.
  • the response using the modified control signals is different from that of a control implemented with true PID signals, but the results are adequate to implement a fast responding and stable control with the modified signals.
  • Figure 7 illustrates the benefits of a power converter utilizing a compensation signal generated from the voltage bus data that has had the second harmonic ripple removed in the generation of a compensation signal.
  • the waveform 7A shows a step increase in the load. The load increase occurs at time 400ms and decreases at 600ms.
  • Waveform 7B shows the corresponding AC current draw. As expected, the AC current draw increases between 400ms and 600ms to meet the increased load. There is a slight overshoot in the current draw as the converter quickly responds to the transient.
  • Waveform 7C shows the bus voltage out of the power converter with a target bus-voltage compensation signal of the present invention. Note the small change in the bus voltage as compared to the prior-art of FIG. 6.
  • the bus voltage deviates from its steady-state value by only about 2V, compared to a 50V deviation seen in the prior-art waveform 6C of FIG. 6.
  • the waveform 7D represents the compensation signal waveform generated by the present invention. One can see that the compensation waveform shows a very fast response to the transient, as compared with the prior-art waveform 6D of FIG. 6.
  • FIG. 3 illustrates a flow chart 300 of the method of generating a compensation signal, either digital or analog, for a power converter where the compensation signal is substantially free of the 100/120 Hz line-frequency ripple.
  • a frequency-locked system clock is generated that is phased locked to an AC line-frequency.
  • the system clock can be generated by a number of means including a PLL comprising analog components, digital components, or a combination thereof.
  • the AC line frequency, to which the PLL locks can be designed to only lock to frequencies of between 64 Hz to 140 Hz. Limiting the frequency range to which the PLL will lock reduces the possibility of accidental locking to a frequency other than the 100/120 Hz second harmonic AC line-frequency ripple.
  • the frequency-locked system clock is provided to the rest of the system for synchronous bus-voltage data generation, queuing the data, and forming the compensation signal.
  • the generation of the frequency-locked system clock can be implemented with discrete components or can be an integrated component. Further, the frequency-locked system clock can be part of a larger system integrated circuit.
  • bus-voltage data is generated at a frequency-locked system clock rate.
  • Digital samples can be generated by an A/D (analog-to-digital) converter and can be buffered. Any A/D converter can be used for generating bus-voltage data so long as the converter can take samples at the desired system clock rate.
  • the resolution (number of bits) of the samples can be selected to provide sufficient sensitivity to process the AC line ripple that is riding on top of the bus voltage. The fewer bits of resolution the A/D has, the more quantization noise in the bus samples and the less sensitivity in generating a control signal that does not contain AC ripple components. Further, the larger the bus voltage, the greater the range needed by the A/D.
  • An A/D with 12-bits of resolution provides a 400V bus with approximately 100 mV resolution, but provides a bus- voltage of 80V with approximately 2OmV resolution.
  • a 10-bit A/D provides substantially the same resolution for a system with a 100V bus voltage or 20 V error range. Thus, the number of bits of resolution on the A/D can be selected to match the minimum desired level AC-ripple on the control signal.
  • the buffered samples are output for queuing. If bus-voltage data is queued in an analog form, then the bus-voltage data can be level shifted, scaled, and buffered before being input into the stack.
  • the bus-voltage data is stored in a stack.
  • the stack provides a way to output data delayed by the stack size and also provide an output for the latest sample value.
  • Bus-voltage data is moved into the stack and output from the stack at a frequency-locked system clock rate.
  • the stack is sized to provide a bus-voltage data delay of one-half cycle of the AC line between the last entered bus-voltage data value and the oldest bus-voltage data value.
  • the stack size is equivalent to one cycle of the rectified bus input signal.
  • a convenient stack depth is sixty- four samples but other stack sizes are within the scope of the invention.
  • the stack is implemented with digital technology known by one skilled in the art of digital components. Also, an analog version of the stack is contemplated. Charge coupled devices can be used to store analog bus-voltage data samples.
  • a compensation signal is generated from bus-voltage data that is substantially free from second harmonic ripple from the AC-input voltage.
  • the ripple free data for a proportional compensation signal component and an integral signal component is generated by a summer that keeps a running sum of the stack values. Because the data in the stack is frequency locked with the AC-line voltage ripple and represents exactly one cycle of the second harmonic AC line ripple, the sum of the stack values will sum to zero. Accordingly the proportional and integral compensation signal generation components are free from second AC-line voltage harmonics.
  • the differential compensation signal component is generated from bus-voltage data that is separated in time by one cycle of the second harmonic of the AC-line voltage. Accordingly, this differential component is free of second harmonic ripple.
  • the compensation signal is converted to a format compatible with the power converter controller (200-FIG. 1). If the compensation signal is digital and the power converter controller takes an analog compensation signal, the compensation signal is converted from digital to analog.
  • FIG. 4 illustrates a flow chart 400 of the method of manufacturing an apparatus for generating a compensation signal that is free of the line-frequency ripple.
  • a frequency-locked system clock generator is provided that can frequency lock to an AC line frequency.
  • the provided system clock generator can be an analog or a digital PLL.
  • the AC line frequency, to which the provided system clock locks, can be limited to between 64 Hz to 140 Hz.
  • the provided frequency-locked system clock generator is configured to provide a system clock for synchronous sampling, stacking, and generating compensation signals.
  • the provided system clock can be implemented with discrete components or can be an integrated circuit. Further, the provided frequency-locked system clock can be part of a larger system ASIC which can include all or part of the power converter controller.
  • a bus-voltage data generator is provided for generation of bus-voltage samples at a synchronous system clock rate.
  • the samples can be analog or digital.
  • the bus- voltage input can be processed by a differential amplifier comparing it to a constant value to generate a bus- voltage error signal.
  • the bus-voltage data can be generated by an A/D (analog-to-digital) converter and can be buffered. Any A/D converter can be used for generating samples so long as the converter can take samples at the desired system clock rate and desired resolution.
  • a stack is provided.
  • the stack inputs and outputs bus- voltage data.
  • the stack output can be configured in multiple ways including but not limited to a latest bus- voltage data output and a delayed bus-voltage output.
  • the bus-voltage data in the stack can be in an analog or a digital form.
  • Bus-voltage error data is moved into the stack and output from the stack at a frequency-locked system clock rate.
  • the stack output can be configured to output the latest data sample and a data sample delayed by one or more complete cycles of the AC-line second harmonic.
  • the stack is preferably sized to provide a bus sample delay of one-half cycle of the AC line, but integral multiples of the one-half cycle are also contemplated.
  • a convenient stack depth is sixty-four samples but other stack sizes are within the scope of the invention.
  • a compensation signal generator is provided.
  • the compensation signal generator provides a compensation signal by the methods described above for the method step 340.
  • a compensation signal converter is provided.
  • the compensation signal can be digital where the power converter expects an analog signal.
  • the signal converter provides any required conversion including but not limited to conversion from digital to analog, analog to digital, level matching, and impedance matching.

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Rectifiers (AREA)
  • Dc-Dc Converters (AREA)
  • Supply And Distribution Of Alternating Current (AREA)
  • Inverter Devices (AREA)

Abstract

L'invention porte sur un appareil pour générer un signal de compensation pour un convertisseur de puissance. L'appareil comprend un générateur d'horloge verrouillée en fréquence, un générateur de données de tension de bus, une pile et un générateur de signal de compensation. L'horloge verrouillée en fréquence est couplée au bus de tension de convertisseur de puissance qui contient des harmoniques de la fréquence de ligne en courant alternatif. Le générateur d'horloge se verrouille en fréquence sur la seconde harmonique de la fréquence de ligne en CA et crée une horloge de système utilisée pour les opérations synchrones dans tout l'appareil. Le générateur de données de tension de bus entre une tension de bus mise à l'échelle par convertisseur de puissance, génère des données de tension de bus à une fréquence d'échantillonnage déterminé par l'horloge de système couplé. La sortie du générateur de bus est entrée dans une pile. La sortie de la pile est couplée à un sommateur pour retirer l'ondulation de seconde harmonique, et est utilisée par un filtre PID modifié pour générer un signal de compensation.
PCT/US2009/065300 2008-11-21 2009-11-20 Correction de facteur de puissance (pfc) variable et commande de tension de bus connecté à un réseau WO2010059927A1 (fr)

Priority Applications (5)

Application Number Priority Date Filing Date Title
KR1020117011450A KR101757252B1 (ko) 2008-11-21 2009-11-20 가변 ｐｆｃ 및 계통 연계형 버스 전압 제어
EP09828283.3A EP2359461B1 (fr) 2008-11-21 2009-11-20 Correction de facteur de puissance (pfc) variable et commande de tension de bus connecté à un réseau
JP2011537648A JP5502887B2 (ja) 2008-11-21 2009-11-20 電力変換装置の補償信号生成装置及び補償信号生成方法並びに補償信号生成装置の製造方法
CN200980152586.0A CN102265497B (zh) 2008-11-21 2009-11-20 可变pfc和并网总线电压控制
CA2743423A CA2743423C (fr) 2008-11-21 2009-11-20 Correction de facteur de puissance (pfc) variable et commande de tension de bus connecte a un reseau

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
US12/292,721 US8081019B2 (en) 2008-11-21 2008-11-21 Variable PFC and grid-tied bus voltage control
US12/292,721 2008-11-21

Publications (1)

Publication Number Publication Date
WO2010059927A1 true WO2010059927A1 (fr) 2010-05-27

Family

ID=42195644

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/US2009/065300 WO2010059927A1 (fr) 2008-11-21 2009-11-20 Correction de facteur de puissance (pfc) variable et commande de tension de bus connecté à un réseau

Country Status (7)

Country Link
US (1) US8081019B2 (fr)
EP (1) EP2359461B1 (fr)
JP (1) JP5502887B2 (fr)
KR (1) KR101757252B1 (fr)
CN (1) CN102265497B (fr)
CA (1) CA2743423C (fr)
WO (1) WO2010059927A1 (fr)

Families Citing this family (9)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US9611835B1 (en) 2013-01-11 2017-04-04 Google Inc. Motor control topology for airborne power generation and systems using same
CN104113261B (zh) * 2013-07-29 2017-03-22 广东美的制冷设备有限公司 一种空调压缩机控制电路及变频空调
US9728974B2 (en) * 2013-10-10 2017-08-08 Tmeic Corporation Renewable energy site reactive power control
US9294016B2 (en) * 2013-12-19 2016-03-22 Google Inc. Control methods and systems for motors and generators operating in a stacked configuration
WO2016080870A1 (fr) * 2014-11-17 2016-05-26 Telefonaktiebolaget L M Ericsson (Publ) Alimentation en mode commuté, station de base, et procédé de fonctionnement d'une alimentation en mode commuté
US9954439B1 (en) * 2016-10-23 2018-04-24 Sanken Electric Co., Ltd. Control circuit of switching power-supply device and switching power-supply device
US11428719B2 (en) 2020-01-10 2022-08-30 Analog Devices International Unlimited Company Electrical signal measurement using subdivision
US11099589B1 (en) * 2020-03-16 2021-08-24 Alpha And Omega Semiconductor (Cayman) Ltd. Digitally programmable, fully differential error amplifier
KR20210123683A (ko) * 2020-04-03 2021-10-14 삼성전자주식회사 신호 수신기 및 그것의 동작 방법

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO1993011600A1 (fr) 1991-11-27 1993-06-10 Sundstrand Corporation Systeme a frequence constante et vitesse variable presentant une ondulation reduite de la liaison cc
US6366483B1 (en) * 2000-07-24 2002-04-02 Rockwell Automation Technologies, Inc. PWM rectifier having de-coupled power factor and output current control loops
US20060057996A1 (en) * 2001-06-01 2006-03-16 Branislav Petrovic High frequency low noise phase-frequency detector and phase noise reduction method and apparatus
US20070139164A1 (en) * 1996-05-13 2007-06-21 O'toole James E Radio frequency data communications device

Family Cites Families (86)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4051425A (en) 1975-02-03 1977-09-27 Telephone Utilities And Communications Industries, Inc. Ac to dc power supply circuit
JPS5579676A (en) * 1978-12-13 1980-06-16 Toshiba Corp Harmonic filter for electric power
US4273406A (en) 1978-12-28 1981-06-16 Mitsuoka Electric Mfg. Co., Ltd. Electrical cord adapter
US4467268A (en) * 1980-04-30 1984-08-21 Raytheon Company Digitally controlled power supply
US4695933A (en) 1985-02-11 1987-09-22 Sundstrand Corporation Multiphase DC-DC series-resonant converter
US4712160A (en) 1985-07-02 1987-12-08 Matsushita Electric Industrial Co., Ltd. Power supply module
DE3604882A1 (de) 1986-02-15 1987-08-20 Bbc Brown Boveri & Cie Leistungshalbleitermodul und verfahren zur herstellung des moduls
US4841220A (en) 1987-09-23 1989-06-20 Tabisz Wojciech A Dc-to-Dc converters using multi-resonant switches
US4857822A (en) 1987-09-23 1989-08-15 Virginia Tech Intellectual Properties, Inc. Zero-voltage-switched multi-resonant converters including the buck and forward type
US4866637A (en) * 1987-10-30 1989-09-12 International Business Machines Corporation Pipelined lighting model processing system for a graphics workstation's shading function
US4866367A (en) 1988-04-11 1989-09-12 Virginia Tech Intellectual Properties, Inc. Multi-loop control for quasi-resonant converters
US4899256A (en) 1988-06-01 1990-02-06 Chrysler Motors Corporation Power module
US4893227A (en) 1988-07-08 1990-01-09 Venus Scientific, Inc. Push pull resonant flyback switchmode power supply converter
US4890217A (en) 1988-07-26 1989-12-26 Norand Corporation Universal power supply, independent converter stages for respective hardware components of a computerized system
US5164657A (en) 1988-08-08 1992-11-17 Zdzislaw Gulczynski Synchronous switching power supply comprising buck converter
US4975821A (en) 1989-10-10 1990-12-04 Lethellier Patrice R High frequency switched mode resonant commutation power supply
US5101322A (en) 1990-03-07 1992-03-31 Motorola, Inc. Arrangement for electronic circuit module
DE4015030C1 (fr) 1990-05-10 1991-11-21 Bicc-Vero Elektronics Gmbh, 2800 Bremen, De
GB9104482D0 (en) 1991-03-04 1991-04-17 Cooperheat Int Ltd Solid state dc power supply
FR2679075B1 (fr) 1991-07-09 1993-10-22 Moulinex Sa Dispositif de detection du dysfonctionnement d'une charge telle qu'un magnetron.
JP2642548B2 (ja) 1991-09-26 1997-08-20 株式会社東芝 半導体装置およびその製造方法
GB9206012D0 (en) 1992-03-19 1992-04-29 Astec Int Ltd Mosfet gate drive circuit
DE4313359A1 (de) 1992-04-24 1993-10-28 Matsushita Electric Ind Co Ltd Schaltnetzteil
US6091611A (en) 1994-04-26 2000-07-18 Comarco Wireless Technologies, Inc. Connectors adapted for controlling a small form factor power supply
JP2776493B2 (ja) 1994-08-12 1998-07-16 インターナショナル・ビジネス・マシーンズ・コーポレイション 電子機器用電源装置及びその制御方法
US5565761A (en) 1994-09-02 1996-10-15 Micro Linear Corp Synchronous switching cascade connected offline PFC-PWM combination power converter controller
US5712772A (en) 1995-02-03 1998-01-27 Ericsson Raynet Controller for high efficiency resonant switching converters
US5592128A (en) 1995-03-30 1997-01-07 Micro Linear Corporation Oscillator for generating a varying amplitude feed forward PFC modulation ramp
US5747977A (en) 1995-03-30 1998-05-05 Micro Linear Corporation Switching regulator having low power mode responsive to load power consumption
US5903138A (en) 1995-03-30 1999-05-11 Micro Linear Corporation Two-stage switching regulator having low power modes responsive to load power consumption
DE19513065A1 (de) 1995-04-07 1996-10-10 Philips Patentverwaltung Schaltungsanordnung zum Erzeugen einer galvanisch getrennten Ausgangsgleichspannung
EP1022727B1 (fr) 1996-04-15 2002-12-18 Matsushita Electric Industrial Co., Ltd. Disque optique et appareil d'enregistrement et de reproduction
US5798635A (en) 1996-06-20 1998-08-25 Micro Linear Corporation One pin error amplifier and switched soft-start for an eight pin PFC-PWM combination integrated circuit converter controller
US5804950A (en) 1996-06-20 1998-09-08 Micro Linear Corporation Input current modulation for power factor correction
US5742151A (en) 1996-06-20 1998-04-21 Micro Linear Corporation Input current shaping technique and low pin count for pfc-pwm boost converter
US7047059B2 (en) 1998-08-18 2006-05-16 Quantum Magnetics, Inc Simplified water-bag technique for magnetic susceptibility measurements on the human body and other specimens
US5905369A (en) 1996-10-17 1999-05-18 Matsushita Electric Industrial Co., Ltd. Variable frequency switching of synchronized interleaved switching converters
US5786687A (en) 1996-12-03 1998-07-28 Compaq Computer Corporation Transformer-isolated pulse drive circuit
US5818207A (en) 1996-12-11 1998-10-06 Micro Linear Corporation Three-pin buck converter and four-pin power amplifier having closed loop output voltage control
US5894243A (en) 1996-12-11 1999-04-13 Micro Linear Corporation Three-pin buck and four-pin boost converter having open loop output voltage control
KR100224103B1 (ko) 1996-12-14 1999-10-15 윤종용 공진형 전원 스위치장치
US5870294A (en) 1997-09-26 1999-02-09 Northern Telecom Limited Soft switched PWM AC to DC converter with gate array logic control
US6147869A (en) 1997-11-24 2000-11-14 International Rectifier Corp. Adaptable planar module
DE19808637A1 (de) 1998-02-28 1999-09-09 Bosch Gmbh Robert Gleichspannungs-/Gleichspannungswandler mit einem Transformator und einer Drossel
JP4121185B2 (ja) 1998-06-12 2008-07-23 新電元工業株式会社 電子回路装置
US6326740B1 (en) 1998-12-22 2001-12-04 Philips Electronics North America Corporation High frequency electronic ballast for multiple lamp independent operation
US6344980B1 (en) 1999-01-14 2002-02-05 Fairchild Semiconductor Corporation Universal pulse width modulating power converter
US6091233A (en) 1999-01-14 2000-07-18 Micro Linear Corporation Interleaved zero current switching in a power factor correction boost converter
US6069803A (en) 1999-02-12 2000-05-30 Astec International Limited Offset resonance zero volt switching flyback converter
US6160725A (en) 1999-03-12 2000-12-12 Nmb Usa Inc. System and method using phase detection to equalize power from multiple power sources
US6058026A (en) 1999-07-26 2000-05-02 Lucent Technologies, Inc. Multiple output converter having a single transformer winding and independent output regulation
US6396277B1 (en) 1999-10-01 2002-05-28 Snap-On Technologies, Inc. Coil on plug signal detection
US6452366B1 (en) 2000-02-11 2002-09-17 Champion Microelectronic Corp. Low power mode and feedback arrangement for a switching power converter
US6480399B2 (en) 2000-03-02 2002-11-12 Power Integrations, Inc. Switched mode power supply responsive to current derived from voltage across energy transfer element input
JP2001356650A (ja) * 2000-06-12 2001-12-26 Minolta Co Ltd 画像形成装置に関する情報の表示方法及び画像形成装置。
KR100595718B1 (ko) 2000-07-28 2006-07-03 엘지전자 주식회사 휴대용 컴퓨터 시스템의 2차 배터리 연결장치 및 방법
US6407514B1 (en) 2001-03-29 2002-06-18 General Electric Company Non-synchronous control of self-oscillating resonant converters
US6531854B2 (en) 2001-03-30 2003-03-11 Champion Microelectronic Corp. Power factor correction circuit arrangement
US20020196006A1 (en) 2001-06-21 2002-12-26 Champion Microelectronic Corp. Volt-second balanced PFCPWM power converter
US6487095B1 (en) 2001-10-31 2002-11-26 International Business Machines Corporation Multiphase zero-volt-switching resonant DC-DC regulator
US6671189B2 (en) 2001-11-09 2003-12-30 Minebea Co., Ltd. Power converter having primary and secondary side switches
US7061775B2 (en) 2002-01-16 2006-06-13 Rockwell Automation Technologies, Inc. Power converter having improved EMI shielding
US6583999B1 (en) 2002-01-25 2003-06-24 Appletec Ltd. Low output voltage, high current, half-bridge, series-resonant, multiphase, DC-DC power supply
US7212420B2 (en) 2002-02-12 2007-05-01 Sheng Hsin Liao Universal serial bus voltage transformer
US6657417B1 (en) 2002-05-31 2003-12-02 Champion Microelectronic Corp. Power factor correction with carrier control and input voltage sensing
US7035126B1 (en) 2002-06-10 2006-04-25 Comarco Wireless Technologies, Inc. Programmable power supply capable of receiving AC and DC power input
US7038406B2 (en) 2003-02-07 2006-05-02 Visteon Global Technologies, Inc. Bi-directional field control for proportional control based generator/alternator voltage regulator
US6970366B2 (en) 2003-04-03 2005-11-29 Power-One As Phase-shifted resonant converter having reduced output ripple
US20040228153A1 (en) 2003-05-14 2004-11-18 Cao Xiao Hong Soft-switching techniques for power inverter legs
US6958920B2 (en) 2003-10-02 2005-10-25 Supertex, Inc. Switching power converter and method of controlling output voltage thereof using predictive sensing of magnetic flux
JP2005151662A (ja) 2003-11-13 2005-06-09 Sharp Corp インバータ装置および分散電源システム
US7418106B2 (en) 2004-06-21 2008-08-26 Nokia Corporation Apparatus and methods for increasing magnetic field in an audio device
US7355865B2 (en) * 2004-08-13 2008-04-08 Rockwell Automation Technologies, Inc. Method and apparatus for rejecting the second harmonic current in an active converter with an unbalanced AC line voltage source
JP2006223008A (ja) 2005-02-08 2006-08-24 Hitachi Ltd Ｄｃ−ｄｃコンバータ
US7064497B1 (en) 2005-02-09 2006-06-20 National Taiwan University Of Science And Technology Dead-time-modulated synchronous PWM controller for dimmable CCFL royer inverter
US7099165B1 (en) * 2005-04-12 2006-08-29 Hamilton Sundstrand Corporation Network harmonic scrubber
TWI297977B (en) 2005-07-05 2008-06-11 Delta Electronics Inc Soft switching dc-dc converter
US7324354B2 (en) 2005-07-08 2008-01-29 Bio-Rad Laboratories, Inc. Power supply with a digital feedback loop
US7286376B2 (en) 2005-11-23 2007-10-23 System General Corp. Soft-switching power converter having power saving circuit for light load operations
JP5420910B2 (ja) 2006-02-14 2014-02-19 フレクストロニクス エーピー，リミテッド ライアビリティ カンパニー 電力変換装置
GB0610422D0 (en) 2006-05-26 2006-07-05 Cambridge Semiconductor Ltd Forward power converters
US7499301B2 (en) 2006-07-07 2009-03-03 Tinyplug Technology (Shenzhen) Limited Plugtype power supply unit
US7639520B1 (en) * 2007-02-26 2009-12-29 Network Appliance, Inc. Efficient power supply
TWI367623B (en) 2008-03-14 2012-07-01 Delta Electronics Inc Parallel-connected resonant converter circuit and controlling method thereof
US8102678B2 (en) 2008-05-21 2012-01-24 Flextronics Ap, Llc High power factor isolated buck-type power factor correction converter
US8693213B2 (en) 2008-05-21 2014-04-08 Flextronics Ap, Llc Resonant power factor correction converter

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO1993011600A1 (fr) 1991-11-27 1993-06-10 Sundstrand Corporation Systeme a frequence constante et vitesse variable presentant une ondulation reduite de la liaison cc
US20070139164A1 (en) * 1996-05-13 2007-06-21 O'toole James E Radio frequency data communications device
US6366483B1 (en) * 2000-07-24 2002-04-02 Rockwell Automation Technologies, Inc. PWM rectifier having de-coupled power factor and output current control loops
US20060057996A1 (en) * 2001-06-01 2006-03-16 Branislav Petrovic High frequency low noise phase-frequency detector and phase noise reduction method and apparatus

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
See also references of EP2359461A4

Also Published As

Publication number Publication date
US20100127737A1 (en) 2010-05-27
KR101757252B1 (ko) 2017-07-12
US8081019B2 (en) 2011-12-20
CN102265497B (zh) 2014-06-25
CN102265497A (zh) 2011-11-30
EP2359461A1 (fr) 2011-08-24
JP5502887B2 (ja) 2014-05-28
KR20110086094A (ko) 2011-07-27
JP2012510252A (ja) 2012-04-26
EP2359461B1 (fr) 2021-09-15
EP2359461A4 (fr) 2017-11-08
CA2743423C (fr) 2017-04-04
CA2743423A1 (fr) 2010-05-27

Similar Documents

Publication Publication Date Title
CA2743423C (fr) Correction de facteur de puissance (pfc) variable et commande de tension de bus connecte a un reseau
US6977827B2 (en) Power system having a phase locked loop with a notch filter
US7986539B2 (en) Method and apparatus for maximum power point tracking in power conversion based on dual feedback loops and power ripples
US7239043B2 (en) Power conversion apparatus and methods using an adaptive waveform reference
US10848057B2 (en) Synchronous sampling DC link voltage control for microinverters
JP3914001B2 (ja) マルチエネルギー源からの電力充電装置
EP2856598A1 (fr) Convertisseurs de puissance bidirectionnels avec commande de tension alternative d'entrée
US7778053B2 (en) Power system having a voltage regulator with a notch filter
EP1741178A2 (fr) Appareil de conversion de courant et procedes utilisant une reference de phase derivee d'une tension de bus de courant continu
Lim et al. PR based indirect current control for seamless transfer of grid-connected inverter
Saxena et al. PV integrated grid synchronization technique using modified SOGI-FLL and zero-crossing detector
Silva et al. Nineteen-level active filter system using asymmetrical cascaded converter with DC voltages control
US8559198B2 (en) High yield AC-AC power converter and method therefor
Brambhatt et al. Experimental assessment of Single Phase Grid assisted system using Second Order Generalized Integrator–Frequency Locked Loop
Meena et al. Comparison of modern voltage parameter estimation techniques under various voltage conditions
JP2006511188A (ja) 交流電気幹線に電気エネルギーを供給する方法及びシステム
Chatterjee Solar Photovoltaic Grid Interactive Inverter Control with Reduced Sensors Count
Patel et al. Current Control Strategy using Sinusoidal Signal Integrators for Harmonic Elimination in a Standalone Microgird
Roy et al. DTLMMS Based Current Control for Three-Phase Single-Stage SPV-Battery Sourced Microgrid with CCROGI-FLL Enabled Synchronization
Nasiri et al. Full digital control of a single-phase series-parallel uninterruptible power supply
JPH02280699A (ja) 発電装置の電圧制御装置
Lee et al. Predictive Current Control of a Grid-Connected Inverter with Grid Voltage Observer
JP2000060150A (ja) 電力変換器の制御装置

Legal Events

Date Code Title Description
WWE Wipo information: entry into national phase

Ref document number: 200980152586.0

Country of ref document: CN

121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 09828283

Country of ref document: EP

Kind code of ref document: A1

WWE Wipo information: entry into national phase

Ref document number: 2743423

Country of ref document: CA

WWE Wipo information: entry into national phase

Ref document number: 2009828283

Country of ref document: EP

ENP Entry into the national phase

Ref document number: 20117011450

Country of ref document: KR

Kind code of ref document: A

WWE Wipo information: entry into national phase

Ref document number: 2011537648

Country of ref document: JP

NENP Non-entry into the national phase

Ref country code: DE