WO2009124469A1 - 双三极管电流控制型自振荡反激变换器 - Google Patents

双三极管电流控制型自振荡反激变换器 Download PDF

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Publication number
WO2009124469A1
WO2009124469A1 PCT/CN2009/070007 CN2009070007W WO2009124469A1 WO 2009124469 A1 WO2009124469 A1 WO 2009124469A1 CN 2009070007 W CN2009070007 W CN 2009070007W WO 2009124469 A1 WO2009124469 A1 WO 2009124469A1
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Prior art keywords
mos transistor
resistor
transistor
gate
current control
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PCT/CN2009/070007
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English (en)
French (fr)
Inventor
尹向阳
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广州金升阳科技有限公司
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Application filed by 广州金升阳科技有限公司 filed Critical 广州金升阳科技有限公司
Priority to DE112009000740T priority Critical patent/DE112009000740T5/de
Priority to US12/935,229 priority patent/US20110026278A1/en
Publication of WO2009124469A1 publication Critical patent/WO2009124469A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/338—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement
    • H02M3/3385—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement with automatic control of output voltage or current

Definitions

  • the present invention relates to a self-oscillating flyback converter for use in a low power DC-DC converter power supply, and more particularly to an input terminal dual triode current controlled self-oscillating flyback converter.
  • the circuit block diagram of the prior art self-oscillating flyback converter (RCC, Ring Choke Converter) is shown in Figure 1.
  • the RCC mainly includes a filtering part, a soft start part, a MOS tube, a transformer, a pulse frequency modulation part (PFM), a reference amplifying part, an isolated optocoupler, and a regulated output loop part.
  • the input power is connected to the output circuit via the transformer; the soft start portion is connected to the gate of the MOS transistor; and the gate of the MOS transistor is also connected to the PFM.
  • a reference amplification portion and an isolation optocoupler are connected between the PFM and the regulated output loop portion to form a voltage negative feedback loop.
  • FIG. 2 A self-oscillating flyback converter of a low-power DC-DC converter power supply commonly used in the industry is shown in FIG. 2 .
  • the soft start part is mainly composed of resistors Rl, R7, R8 and capacitor C9.
  • the resistors R1, R7 and R8 are connected in series; the capacitor C9 is connected in parallel across the resistors R7 and R8.
  • the PFM includes an NPN type transistor TR2, capacitors C1, C2, resistors R2, R3, R4, a freewheeling diode D3, and a feedback winding P2.
  • the input voltage is connected from the same name end of the primary winding PI.
  • the different name of the primary winding P1 is terminated with the drain of the MOS transistor TR1.
  • the source of the MOS transistor TR1 is grounded via a resistor R4 and the bias resistor R3 is connected to the base of the transistor TR2.
  • a capacitor C2 is connected in parallel across the bias resistor R3.
  • the collector of the transistor TR2 is connected to the gate of the MOS transistor TR1.
  • the emitter of the transistor TR2 is grounded.
  • the same-name end of the feedback winding P2 is connected to the gate of the MOS transistor TR1 via a capacitor C1 and a resistor R2.
  • the cathode of the freewheeling diode D3 is connected to the same end of the feedback winding P2.
  • the anode of the freewheeling diode D3 is grounded one way, and the other is connected to the optocoupler OCl via a capacitor C51.
  • the input voltage is additionally connected to the gate of the MOS transistor TR1 via a soft start portion.
  • the reference amplification section is composed of a voltage regulator Adj.
  • the function is that the sampling voltage of the output loop portion is a negative feedback signal, and is input to the base of the transistor TR2 of the PFM via the optocoupler OC1 to form a voltage negative feedback loop.
  • the regulated output loop part is mainly formed by connecting the secondary winding P3 of the transformer T1, the rectifier diode D1 and the filter capacitor C3. Since the MOS transistor TR1 is in the process of being turned off, the internal junction capacitance Ciss can only be charged through the capacitor C1, the resistor R2, and the transformer T1 feedback winding P2 to form a discharge loop. Due to the long discharge time constant, its turn-off waveform is distorted. The power loss of the MOS transistor TR1 is large, which makes the overall efficiency of the product low.
  • the Vgl point voltage is high due to the very large current at the short circuit.
  • the conduction strength of the MOS transistor TR1 is increased, the drain current Id is increased, and the voltage drop across R4 is also increased.
  • the conduction strength of the transistor TR2 is increased, and the potential of the Vgl point is pulled low, and the TR1 gradually withdraws from the saturation state.
  • the conduction internal resistance of the MOS transistor TR1 increases the drain current Id, but since the transistor TR2 operates in an amplified state, the gate voltage Vgl of the MOS transistor TR1 is not pulled low, and the MOS transistor TR1 does not enter a reliable cutoff. In the state, a large drain current Id still appears, and the short-circuit power consumption is large.
  • Short circuit power? 8 input voltage Vin* input short-circuit current Ii (Ii is approximately MOS transistor TR1 drain current Id). It is known that the short-circuit power Ps is proportional to the input voltage Vin and will increase as the input voltage Vin increases.
  • a circuit having a change in input voltage ratio of 2:1 to 4:1 or more has some troublesome problems in practical applications.
  • the main shortcomings are: waveform distortion occurs when MOS transistor TR1 is cut off, increasing the switching loss of MOS transistor TR1, making the product overall efficiency less, increasing product noise; short-circuit power is large, and increases with input voltage; MOS tube The drain-source voltage Vds peak voltage difference of TR1 is high; the operating frequency varies with input voltage and output load, and electromagnetic compatibility (EMI, Electromagnetic Interference is difficult to design, and it is easy to generate oscillation when no-load operation, resulting in unstable output voltage.
  • EMI Electromagnetic Interference
  • the object of the present invention is to provide a dual triode current control type self-oscillating flyback converter, which can reduce switching loss and short circuit power consumption, and improve the overall performance of air and full load of the whole machine.
  • the present invention provides a dual triode current control type self-oscillating flyback converter including a soft start portion, a MOS transistor TR1, a transformer T1, a pulse frequency modulation portion PFM, a reference amplification portion, an isolated optocoupler OC1, and a regulated output loop portion;
  • the input power is connected to the output circuit via the transformer T1;
  • the soft start portion is connected to the gate of the MOS transistor TR1;
  • the gate of the MOS transistor TR1 is also connected to the pulse frequency modulation portion PFM;
  • the pulse frequency modulation portion PFM and the regulated output loop portion are connected between the reference amplifying portion and the isolation optocoupler OC1 to form a voltage negative feedback loop;
  • the pulse frequency modulation portion PFM mainly includes a transistor TR2, a resistor R3, a capacitor C2, and a resistor R4; a base of the transistor TR2 is connected to a source of the MOS transistor TR1 through a parallel bias resistor R3 and a capacitor C2; The source of the MOS transistor TR1 is grounded through a resistor R4;
  • a triode current control circuit is added to the pulse frequency modulation portion PFM, and the triode current control circuit is connected between the MOS transistor TR1 and the triode TR2 to realize a self-oscillation output of the input double-triode current control.
  • the triode current control circuit comprises a triode TR3 and a resistor R36;
  • the emitter of the transistor TR3 is connected to the gate of the MOS transistor TR1, the base is connected to the gate of the MOS transistor TR1 via a bias resistor R36, and the collector of the transistor TR2 is further connected; the collector is connected to the transistor TR2 base;
  • the base of the transistor TR2 is connected to the source of the MOS transistor TR1 via a bias resistor R3; the source of the MOS transistor TR1 is grounded via a resistor R4.
  • the triode current control circuit comprises a triode TR3 and a resistor R36;
  • the emitter of the transistor TR3 is connected to the gate of the MOS transistor TR1; the base is connected to the gate of the MOS transistor TR1 via a bias resistor R36, and the other is connected to the collector of the transistor TR2; the collector is connected to the MOS transistor TR1 Source
  • the source of the MOS transistor TR1 is connected to the base of the transistor TR2 via a bias resistor R3; the source of the MOS transistor TR1 is grounded via a resistor R4.
  • the triode current control circuit comprises a triode TR3, a resistor R36 and a resistor R27; the emitter of the triode TR3 is connected to the gate of the MOS transistor TR1; the base is connected to the MOS transistor TR1 via a bias resistor R36. The other pole is connected to the collector of the transistor TR2 via the bias resistor R27; the collector is connected to the source of the MOS transistor TR1;
  • the source of the MOS transistor TR1 is connected to the base of the transistor TR2 via a bias resistor R3;
  • the source of the MOS transistor TR1 is grounded via a resistor R4.
  • the current transformer S1 and the freewheeling diode D5 are connected between the source of the MOS transistor TR1 and the resistor R4;
  • the primary of the current transformer S1 is terminated with the same name as the source of the MOS transistor TR1;
  • the secondary of the current transformer S1 is terminated by the same name as the anode of the diode D5;
  • the cathode of the diode D5 is connected to the resistor R4; the two different ends of the current transformer S 1 are grounded.
  • a capacitor C34 is connected in parallel across the bias resistor R36; a capacitor C2 is coupled across the bias resistor R3.
  • the MOS transistor TR1 is of an N-channel type
  • the transistor TR2 is of the NPN type
  • the transistor TR3 is of a PNP type.
  • the gate of the MOS transistor TR1 is connected to the Zener diode Z1;
  • the cathode of the Zener diode Z1 is connected to the gate of the MOS transistor TR1, and the anode is grounded.
  • the soft start circuit is composed of a resistor R1, a resistor R8, a capacitor C9 and a diode D2;
  • the input terminal VIN is connected in series with the resistor R1, and the other circuit is connected to the anode of the diode D2;
  • the cathode of the diode D2 is grounded through a resistor R8; the cathode of the diode D2 also has a gate of the MOS transistor TR1.
  • the diode D2 is a fast recovery diode.
  • the invention adopts double triode pulse frequency modulation at the input end, the off time of the MOS transistor TR1 is greatly shortened, and the overall machine efficiency of the product is improved.
  • the triode TR3 forms the discharge loop of the internal junction capacitance Ciss of the MOS transistor TR1
  • the short-circuit power of the product is largely reduced.
  • the advantage of the present invention over the prior art is that the dual triode current control type self-oscillating flyback converter has high working efficiency, can work at no load, and can ensure stable output voltage; and the no-load power consumption is very small, only ⁇ - ⁇ ⁇ level no-load power consumption; short-circuit power is very small, basically does not change with input voltage; achieve continuous short-circuit protection; dynamic response is fast.
  • FIG. 1 is a circuit block diagram of a prior art
  • FIG. 3 is a schematic circuit diagram of a first embodiment of the present invention.
  • FIG. 5 is a waveform diagram of a gate voltage (Vgl) when the MOS transistor of the prior art is at a nominal full load
  • FIG. 6 is a waveform diagram of a gate voltage (Vgl) when the MOS transistor is in a steady state nominal full load according to the first embodiment of the present invention
  • FIG. 8 is a waveform diagram of a drain voltage (Vds) when the MOS transistor of the prior art is at a nominal full load
  • FIG. 8 is a waveform diagram of a drain voltage (Vds) when the MOS transistor is in a steady state nominal full load according to the first embodiment of the present invention
  • 9 is a schematic circuit diagram of a second embodiment of the present invention.
  • Figure 10 is a circuit schematic diagram of Embodiment 3 of the present invention.
  • Figure 11 is a circuit diagram of the fourth embodiment of the present invention.
  • the present invention provides a dual triode current control type self-oscillating flyback converter mainly comprising a soft start part, a MOS transistor TR1, a transformer Tl, a PFM, a reference amplification part, an optocoupler isolation and a regulated output loop. section.
  • PFM mainly includes NPN type transistor TR2, PNP type transistor TR3, capacitors C1, C2, resistors R2, R3, R4, R27, R36, C34, Zener Z1, freewheeling diode D3 and feedback winding P2.
  • the input voltage is connected to the gate of the MOS transistor TR1 via the soft start portion, and the other is connected from the same end of the primary winding P1.
  • the primary winding P1 is terminated by the drain of the MOS transistor TR1.
  • the source of the MOS transistor TR1 is grounded through the resistor R4 and connected to the base of the transistor TR2 through the bias resistor R3, and the capacitor C2 is connected in parallel across the resistor R3.
  • Transistor TR2 collector is connected to resistor R27, and the emitter is grounded.
  • the feedback winding P2 is connected to the gate of the MOS transistor TR1 via the capacitor C1 and the resistor R2.
  • the emitter of the transistor TR3 is connected to the gate of the MOS transistor TR1; the collector is connected to the source of the MOS transistor TR1; the base is connected via the bias resistor R36, the capacitor C34 is connected to the gate of the MOS transistor TR1, and the other is connected in series with a bias resistor R27.
  • Transistor TR2 collector is connected to the gate of the MOS transistor TR1 via the capacitor C1 and the resistor R2.
  • the cathode of the freewheeling diode D3 is connected to the same name end of the feedback winding P2, the anode is grounded one way, and the other is connected to the optocoupler OCl via the capacitor C51.
  • Zener diode Z1 is connected to the gate of the MOS transistor TR1.
  • the cathode of the Zener diode Z1 is connected to the gate of the MOS transistor TR1, and the anode is grounded.
  • Zener diode Z1 is used to limit the gate voltage of MOSFET TR1 when it is used for high voltage input, and it can also improve the phenomenon of no-load oscillation.
  • the regulated output loop part is mainly formed by connecting the secondary winding P3 of the transformer T1, the rectifier diode D1 and the filter capacitor C3.
  • the reference amplification part is composed of a voltage regulator Adj, which functions as a negative feedback signal of the output loop portion, and is input to the base of the transistor TR2 of the pulse frequency modulation portion via the optocoupler OC1 to form a negative feedback voltage loop.
  • the specific working principle of the present invention is as follows: After the input terminal VIN is applied with a voltage, the resistors R1 and D2 are applied to the gate of the MOS transistor TR1 to charge the internal junction capacitance Ciss of the MOS transistor TR1.
  • the MOS transistor TR1 When the gate voltage Vgl of the MOS transistor TR1 reaches the on-voltage Vth, the MOS transistor TR1 is turned on. Thus, the primary winding P1 of the transformer T1 generates a self-induced electromotive force that is positively negative.
  • the rectifying and filtering circuit connected to the secondary winding P3 of the transformer T1 is stopped by the inversion of the induced electromotive force, the electric energy is stored in the primary winding P1 of the transformer T1 by magnetic energy. Due to the positive feedback avalanche process time is very short, the capacitor C1 is too late to charge. At the same time, due to the mutual inductance, the feedback winding P2 of the transformer T1 also generates a positive and negative induced electromotive force, and a positive feedback loop composed of a capacitor C1 and a resistor R2 is applied to the gate of the MOS transistor TR1 to further the gate voltage Vgl. Ascending, the MOS transistor TR1 quickly enters a saturated state.
  • the induced voltage on the feedback winding P2 charges the capacitor C1.
  • the potential difference across the capacitor C1 rises.
  • the gate voltage Vgl of the MOS transistor TR1 drops, causing the MOS transistor TR1 to gradually exit the saturation state.
  • the drain current Id flowing through the primary winding PI and the MOS transistor TR1 increases with time, and the voltage drop across the resistor R4 also increases.
  • the voltage reaches (0.7 + VR3) (where VR3 is the resistance R3 voltage)
  • the transistor TR2 is turned on, and the base voltage of the transistor TR3 is lowered, thereby turning on the transistor TR3.
  • the collector current of the transistor TR3 increases, and the transistor TR2 conductance increases, so that the cycle reciprocates, and finally the transistors TR2, TR3 become saturated.
  • the energy stored in the Ciss during the saturation conduction of the MOS transistor TR1 is released to the ground through TR3, thereby causing the MOS transistor TR1 to enter a reliable cut-off state.
  • the primary winding P1 When the primary winding PI energy drops to a certain value, the primary winding P1 generates an inverse electromotive force to prevent the primary current from dropping. This current produces a positive induced negative electromotive force on the primary winding P1.
  • the feedback winding P2 generates a positive pulse voltage through the positive feedback loop, causing the transistor TR1 to turn back on. Therefore, the switching power supply operates in a self-oscillating state.
  • the oscillation frequency is mainly determined by the inductance Lp of the transformer T1.
  • This circuit will be flyback after the self-oscillation operation.
  • the transformer T1 stores energy; when the MOS transistor is turned off, the transformer T1 outputs energy, and the energy is output through the regulated output loop to realize energy transmission.
  • the output energy is supplied to the load all the way, and the other path is sampled and compared by the reference amplification part, and is input to the base of the PFM transistor TR2 via the optocoupler OC1 to control the current on the base of the transistor TR2, thereby adjusting the MOS transistor TR1 and the triode
  • the on-off time of TR2 realizes the flyback process of the circuit.
  • the drain current Id flowing through the primary winding PI and the MOS transistor TR1 increases with time, and the voltage drop across the resistor R4 also increases.
  • the transistor TR2 When the voltage reaches (0.7V + VR3), the transistor TR2 is turned on, and the base voltage of the transistor TR3 is lowered, so that the transistor TR3 is turned on.
  • the collector current of the transistor TR3 increases, and the transistor TR2 conductance is enhanced, so that the cycle is reciprocated, and finally the transistors TR2 and TR3 are saturated.
  • the circuit When the circuit operates in the output short-circuit state, the current is very large due to the short-circuit, resulting in a high Vgl voltage, the conduction strength of the MOS transistor TR1 is increased, the drain current Id of the MOS transistor TR1 is increased, and the voltage drop across the resistor R4 is also increased.
  • the transistor TR2 When the voltage reaches (0.7+VR3), the transistor TR2 is turned on, the base voltage of the transistor TR3 is lowered, and the transistor TR3 is turned on, and the collector current of the transistor TR3 is increased, so that the conduction capability of the transistor TR2 is enhanced, and thus the cycle is repeated.
  • the transistors TR2, TR3 are saturated.
  • the energy stored in the Ciss during the saturation conduction of the MOS transistor TR1 is released to the ground through the transistor TR3, thereby causing the MOS transistor TR1 to enter a reliable off state.
  • the drain current Id of the MOS transistor TR1 is close to zero, so that the short-circuit power consumption is close to zero.
  • the transformer T1 induces an electromotive force to reverse.
  • the transistors TR2 and TR3 are turned off.
  • the gate voltage Vgl of the MOS transistor TR1 quickly returns to the high level, and the MOS transistor TR1 is turned on, and automatically returns to the normal self-excited oscillation mode of the circuit, thereby realizing continuous short circuit protection of the circuit.
  • the soft-start circuit consists of resistor R1, resistor R8, capacitor C9 and diode D2.
  • Input VIN is connected in series with resistor R1 and then grounded through capacitor C9; the other is connected to the anode of diode D2.
  • the cathode of diode D2 is grounded through resistor R8.
  • the diode D2 cathode also has a MOSFET gate.
  • the fast recovery diode D2 replaces the resistor R7 in the existing circuit shown in FIG. Under normal operating conditions, the fast-recovery diode D2 has an internal resistance rd «R7.
  • the MOS tube TR1 has enhanced startup performance and enhanced capacity with capacitive loads.
  • the characteristic of the diode reverse-cutting is skillfully used to effectively avoid the interference of the driving signal generated by the positive feedback winding to the soft-start circuit, which greatly improves the starting performance of the product.
  • the basic parameters used for the power supply are:
  • the input DC voltage range is 9 ⁇ 18V, and the output is 12V/500mA. It can work normally under no load, light load and full load.
  • the same components of the present invention are the same as those of the prior art.
  • the input voltage efficiency of the present invention is significantly higher than that of the circuit of FIG. 2 in the output load range of 0 to 500 mA, and the smaller the load current is, The difference is greater.
  • the MOS transistor TR1 functions as a power switch tube, and the amplitude of the gate voltage Vgl reaches 9.62 V in the circuit of the present invention at a steady state nominal full load, and only in the circuit shown in FIG. It reached 5.52V.
  • the amplitude of the drain voltage Vds is only 27.4V in the circuit of the present invention, but reaches 32.6V in the circuit shown in FIG.
  • the device has a higher withstand voltage.
  • the cathode of diode D5 is connected to resistor R4, and the two different terminals of current transformer S1 are grounded.
  • the embodiment shown in Fig. 10 is basically the same as the embodiment shown in Fig. 3, except for the connection of the triode TR3.
  • the emitter of the transistor TR3 is connected to the gate of the MOS transistor TR1; the base is connected to the gate of the MOS transistor TR1 via a bias resistor R36, and the other is connected to the collector of the transistor TR2; Tube TR2 base.
  • Transistor TR2 base is connected to MOS transistor TR1 source via bias resistor R3.
  • Capacitor C2 is connected in parallel across the resistor R3.
  • the source of the MOS transistor TR1 is grounded via a resistor R4.
  • the current transformer S1 and the rectifier diode D5 may be connected between the source of the MOS transistor TR1 and the resistor R4, which has the same effect.
  • Fig. 11 The embodiment shown in Fig. 11 is basically the same as the embodiment shown in Fig. 3, except for the connection of the triode TR3.
  • the emitter of the transistor TR3 is connected to the gate of the MOS transistor TR1; the base is connected to the gate of the MOS transistor TR1 via a bias resistor R36, and the collector of the transistor TR2 is connected to the collector; the collector is connected to the source of the MOS transistor TR1.
  • MOS transistor TR1 source is connected to the base of TR2 via bias resistor R3.
  • Capacitor C2 is connected in parallel across the resistor R3.
  • the source of the MOS transistor TR1 is grounded via a resistor R4.
  • the current transformer S1 and the rectifier diode D5 can be connected between the source of the MOS transistor TR1 and the resistor R4, and the same effect can be obtained.

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Description

双三极管电流控制型自振荡反激变换器
本申请要求于 2008 年 4 月 8 日提交中国专利局、 申请号为 200810027284.8、 发明名称为"双三极管电流控制型自振荡反激变换器"的中国 专利申请的优先权， 其全部内容通过引用结合在本申请中。
技术领域
本发明涉及一种应用于小功率 DC-DC变换电源的自振荡反激变换器， 特 别涉及一种输入端双三极管电流控制型自振荡反激变换器。
背景技术
现有技术中自振荡反激变换器（RCC， Ring Choke Converter )的电路原理 框图如图 1所示。 RCC主要包括滤波部分、 软启动部分、 MOS管、 变压器、 脉冲频率调制部分（PFM, Pulse Freqency Modulation ), 基准放大部分、 隔离 光耦、稳压输出回路部分。 输入电量经变压器连接输出回路部分； 软启动部分 连接 MOS管的栅极； MOS管的栅极还连接所述 PFM。 所述 PFM和稳压输出 回路部分之间接基准放大部分和隔离光耦 , 形成电压负反馈回路。
目前业界常用的一种小功率 DC-DC变换电源的自振荡反激变换器如图 2 所示。其中软启动部分主要由电阻 Rl、 R7、 R8和电容 C9组成。其中电阻 Rl、 R7和 R8串联； 电容 C9并联在电阻 R7和 R8两端。
所述 PFM包括 NPN型三极管 TR2、 电容 Cl、 C2、 电阻 R2、 R3、 R4、 续流二极管 D3和反馈绕组 P2。 输入电压从初级绕组 PI的同名端接入。 初级 绕组 P1的异名端接 MOS管 TR1的漏极。 MOS管 TR1的源极分别通过电阻 R4接地和偏置电阻 R3接三极管 TR2的基极。 偏置电阻 R3的两端并联电容 C2。 三极管 TR2集电极接 MOS管 TR1的栅极。 三极管 TR2的发射极接地。 反馈绕组 P2的同名端经电容 Cl、 电阻 R2接 MOS管 TR1的栅极。 续流二极 管 D3的阴极与反馈绕组 P2的同名端相连。 续流二极管 D3的阳极一路接地， 另一路经电容 C51接光耦 OCl。 输入电压另外一路经软启动部分接 MOS管 TR1的栅极。基准放大部分由稳压器 Adj组成。其作用以输出回路部分的采样 电压为负反馈信号， 经光耦 OC1输入到所述 PFM的三极管 TR2的基极， 形 成电压负反馈回路。 稳压输出回路部分主要由变压器 T1的次级绕组 P3、整流 二极管 D1和滤波电容 C3连接而成。 由于 MOS管 TR1在关断过程中，其内部结电容 Ciss所充电量只能通过电 容 Cl、 电阻 R2、 与变压器 T1反馈绕组 P2再到地形成放电回路。 由于放电时 间常数较长， 造成其关断波形失真。 MOS管 TR1关断过程的功率损耗较大， 使产品的整机效率偏低。
当电路工作在输出短路状态时， 由于短路瞬间电流非常大， 导致 Vgl点电 压偏高。 MOS管 TR1导通强度加强， 则漏极电流 Id增大， R4上的压降也增 加， 于是三极管 TR2导通强度增强， 则 Vgl点电位被拉低， TR1逐渐退出饱 和状态。 MOS管 TR1的导通内阻增大漏极电流 Id下降， 但由于三极管 TR2 工作在放大状态， MOS管 TR1的栅极电压 Vgl不会被拉的艮低， MOS管 TR1 不会进入可靠的截止状态， 仍然会出现较大的漏极电流 Id, 短路功耗较大。
当三极管 TR2的基极电压小于 (0.7V+VR3)时（其中 VR3为电阻 R3电压）， TR2截止， Vgl点电位又会升高， 于是 MOS管 TR1导通强度又加强， MOS 管 TR1的漏极电流 Id又增大。 如此循环往复， 电路出现高频自激振荡， MOS 管的开关损耗较大。 根据公式： 短路功率？8 =输入电压 Vin*输入短路电流 Ii (此时 Ii近似为 MOS管 TR1漏极电流 Id )知道短路功率 Ps与输入电压 Vin 成一定的比例关系， 且会随输入电压 Vin增大而增大。假设一款标称输入电压 为 5VDC、 输出功率为 3W的产品， 输入电压在 4.5 ~ 9VDC间变化。 当短路 时输入电压为 5VDC, 短路电流为 0.34A, 则短路功率 Ps = 5*0.34 = 1.7W。 如 此时输入电压为 9VDC, 短路电流为 0.27A, 则短路功率 Ps = 9*0.27 = 2.43W, 可见短路功耗增大。 另一方面， 当 Vgl点电位高于 VI点电位时， 电流将反向 向前级流动，产生对前级电路的干扰。 同时由于变压器缠线工艺过程中具有一 定的离散性， 初次级绕线不平整， 导致初次级漏感较大， 短路功耗也会急剧的 增力口。
上述电路在输入电压变化大的场合， 特别是功率在 10W以下的微功率电 路中，输入电压变化比在 2:1 ~ 4:1变化范围或以上的电路， 实际应用中出现了 一些棘手问题。 主要缺点表现在： MOS 管 TR1 截止时出现波形失真， 增大 MOS管 TR1开关损耗， 使产品整机效率不高， 增加产品噪声； 短路功率大， 且随输入电压增大而增大； MOS管 TR1的漏-源电压 Vds峰值压差高； 工作 频率随输入电压和输出 负 载变化而变化， 电磁兼容 （ EMI , Electromagnetic Interference难设计， 空载工作时易产生振荡 , 导致输出电压不 稳定。
发明内容
本发明目的在于提供一种双三极管电流控制型自振荡反激变换器， 能够降 低开关损耗和短路功耗， 改善整机空、 满载综合性能。
本发明提供一种双三极管电流控制型自振荡反激变换器， 包括软启动部 分、 MOS管 TR1、 变压器 Tl、 脉冲频率调制部分 PFM、 基准放大部分、 隔离 光耦 OC1和稳压输出回路部分；
输入电量经变压器 T1连接输出回路部分；
所述软启动部分连接所述 MOS管 TR1的栅极；
所述 MOS管 TR1的栅极还连接所述脉冲频率调制部分 PFM;
所述脉冲频率调制部分 PFM和稳压输出回路部分之间接所述基准放大部 分和隔离光耦 OC1 , 形成电压负反馈回路；
所述脉冲频率调制部分 PFM主要包括三极管 TR2、 电阻 R3、 电容 C2和 电阻 R4; 所述三极管 TR2的基极通过并联的偏置电阻 R3和电容 C2接所述 MOS管 TR1的源极； 所述 MOS管 TR1的源极通过电阻 R4接地；
所述脉冲频率调制部分 PFM 中增设三极管电流控制电路， 该三极管电流 控制电路连接在所述 MOS管 TR1和所述三极管 TR2之间， 实现输入端双三 极管电流控制的自激振荡输出。
优选地， 所述三极管电流控制电路包括三极管 TR3和电阻 R36;
所述三极管 TR3的发射极接所述 MOS管 TR1的栅极， 基极一路经偏置 电阻 R36接所述 MOS管 TR1栅极， 另外一 矣所述三极管 TR2集电极； 集 电极接所述三极管 TR2基极；
所述三极管 TR2基极经偏置电阻 R3接所述 MOS管 TR1的源极； 所述 MOS管 TR1的源极经电阻 R4接地。
优选地， 所述三极管电流控制电路包括三极管 TR3和电阻 R36;
所述三极管 TR3的发射极接所述 MOS管 TR1的栅极；基极一路经偏置电 阻 R36接所述 MOS管 TR1栅极， 另外一路接三极管 TR2集电极； 集电极接 所述 MOS管 TR1的源极； 所述 MOS管 TR1的源极经偏置电阻 R3接所述三极管 TR2基极； 所述 MOS管 TR1的源极经电阻 R4接地。
优选地 ,所述三极管电流控制电路包括三极管 TR3、电阻 R36和电阻 R27; 所述三极管 TR3的发射极接所述 MOS管 TR1栅极； 基极一路经偏置电 阻 R36接所述 MOS管 TR1栅极， 另外一路经偏置电阻 R27接三极管 TR2集 电极； 集电极接所述 MOS管 TR1的源极；
所述 MOS管 TR1的源极经偏置电阻 R3接三极管 TR2基极；
所述 MOS管 TR1的源极经电阻 R4接地。
优选地，所述 MOS管 TR1的源极与电阻 R4之间连接电流互感器 S1和续 流二极管 D5;
所述电流互感器 S1的初级同名端接所述 MOS管 TR1的源极；
所述电流互感器 S1的次级同名端接所述二极管 D5的阳极；
所述二极管 D5的阴极接电阻 R4; 电流互感器 S 1的两个异名端接地。 优选地， 所述偏置电阻 R36的两端并联电容 C34; 偏置电阻 R3的两端并 联电容 C2。
优选地， 所述 MOS管 TR1为 N沟道型 , 所述三极管 TR2为 NPN型 , 所 述三极管 TR3为 PNP型。
优选地， 所述 MOS管 TR1的栅极连接稳压二极管 Z1;
所述稳压二极管 Z1的阴极接所述 MOS管 TR1的栅极， 阳极接地。
优选地， 所述的软启动电路由电阻 Rl、 电阻 R8, 电容 C9和二极管 D2组 成；
输入端 VIN串接电阻 R1后一路经电容 C9接地； 另外一路接所述二极管 D2的阳极；
所述二极管 D2 阴极通过电阻 R8接地； 二极管 D2阴极还有一 矣所述 MOS管 TR1的栅极。
优选地， 所述二极管 D2为快恢复二极管。
由于本发明在输入端采用双三极管脉冲频率调制，大大缩短了 MOS管 TR1 的关断时间，提高了产品的整机效率。 同时由于三极管 TR3形成 MOS管 TR1 的内部结电容 Ciss的放电回路， 使产品的短路功率得到了很大程度上的降低。 本发明相对现有技术优点在于：双三极管电流控制型自振荡反激变换器的 工作效率高， 能空载工作， 且能保证输出电压稳定； 并且空载功耗非常小， 仅 有 ιο-^ν级别的空载功耗； 短路功率非常小， 基本不随输入电压变化； 实现持 续的短路保护； 动态响应快速。
附图说明
图 1为现有技术的电路原理框图；
图 2为现有技术的电路原理图；
图 3为本发明实施一的电路原理图；
图 4为本发明电路标称输入电压效率与输出负载的曲线特性图；
图 5为现有技术的 MOS管稳态标称满载时栅极电压（ Vgl ) 波形图； 图 6为本发明实施一中的 MOS管稳态标称满载时栅极电压（ Vgl )波形图； 图 Ί为现有技术的 MOS管稳态标称满载时漏极电压（ Vds ) 波形图； 图 8为本发明实施一中的 MOS管稳态标称满载时漏极电压（ Vds )波形图； 图 9为本发明实施二的电路原理图；
图 10为本发明实施三的电路原理图；
图 11为本发明实施四的电路原理图。
具体实施方式
如图 3所示，本发明提供的一种双三极管电流控制型自振荡反激变换器主 要包括软启动部分、 MOS管 TR1、 变压器 Tl、 PFM、 基准放大部分、 光耦隔 离和稳压输出回路部分。
其中 PFM主要包括 NPN型三极管 TR2、 PNP型三极管 TR3、 电容 C1、 C2、 电阻 R2、 R3、 R4、 R27、 R36、 C34、 稳压管 Zl、 续流二极管 D3和反馈 绕组 P2。
输入电压一路经软启动部分接 MOS管 TR1 的栅极， 另一路从初级绕组 P1的同名端接入。
初级绕组 P1异名端接 MOS管 TR1的漏极。 MOS管 TR1的源极分别通 过电阻 R4接地和通过偏置电阻 R3接三极管 TR2基极 , 电阻 R3的两端并联 电容 C2。 三极管 TR2集电极接电阻 R27， 发射极接地。
反馈绕组 P2同名端经电容 C1、 电阻 R2接 MOS管 TR1的栅极。 三极管 TR3的发射极接 MOS管 TR1栅极； 集电极接 MOS管 TR1的源 极； 基极一路经偏置电阻 R36电容 C34接 MOS管 TR1栅极， 另外一路串接 一个偏置电阻 R27后接三极管 TR2集电极。
续流二极管 D3的阴极与反馈绕组 P2的同名端相连， 阳极一路接地， 另 一路经电容 C51接光耦 OCl。
另外在 MOS管 TR1栅极连接稳压二极管 Z1。
稳压二极管 Z1的阴极接 MOS管 TR1栅极， 阳极接地。
稳压二极管 Z1用于高电压输入时限制 MOS管 TR1栅极电压， 同时还可 以改善空载振荡的现象。
稳压输出回路部分主要由变压器 T1的次级绕组 P3、 整流二极管 D1和滤 波电容 C3连接而成。
基准放大部分由稳压器 Adj组成，其作用以输出回路部分的采样电压为负 反馈信号， 经光耦 OC1输入到脉冲频率调制部分的晶极管 TR2的基极， 形成 负反馈电压回路。
本发明的具体工作原理如下： 当输入端 VIN加上电压后， 通过电阻 Rl、 D2加到 MOS管 TR1的栅极， 对 MOS管 TR1内部结电容 Ciss进行充电。
当 MOS管 TR1的栅极电压 Vgl达到导通电压 Vth时， MOS管 TR1导通。 于是变压器 T1的初级绕组 P1产生上正下负的自感电动势。
由于变压器 T1次级绕组 P3所接的整流滤波电路因感应电动势反相而截 止， 电能便以磁能的方式存储在变压器 T1的初级绕组 P1 内部。 由于正反馈 雪崩过程时间极短， 电容 C1来不及充电。 与此同时， 由于互感作用变压器 T1 的反馈绕组 P2也产生上正下负的感应电动势， 通过电容 Cl、 电阻 R2组成的 正反馈回路， 加到 MOS管 TR1的栅极， 使栅极电压 Vgl进一步增大， 于是 MOS管 TR1迅速进入饱和状态。
MOS管 TR1饱和后， 反馈绕组 P2上的感应电压对电容 C1充电， 随着电 容 C1充电的不断进行， 电容 C1两端的电位差升高。 于是 MOS管 TR1的栅 极电压 Vgl下降， 使 MOS管 TR1逐渐退出饱和状态。
MOS管 TR1退出饱和后， 其内阻增大， 导致 MOS管 TR1的漏极电流 Id 进一步下降， 由于电感中的电流不能突变， 于是变压器 T1各个绕组的感应电 动势反相。
同时在 MOS管 TR1饱和导通过程中 , 流经初级绕组 PI、 MOS管 TR1的 漏极电流 Id随着时间的增加而增大， 电阻 R4上的压降也增加。 当电压达到 (0.7+VR3)时（其中 VR3为电阻 R3电压）， 三极管 TR2导通， 三极管 TR3的基 极电压下降， 从而使三极管 TR3导通。 三极管 TR3的集电极电流增大， 三极 管 TR2导通能力增强， 如此循环往复， 最终使三极管 TR2、 TR3进入饱和。 同时， 由于三极管 TR3的导通， MOS管 TR1饱和导通过程中储存在 Ciss的 能量通过 TR3释放到地， 从而使 MOS管 TR1进入可靠的截止状态。
当 MOS管 TR1截止时， 由续流二极管 D3、反馈绕组 P2和电容 C51组成 续流回路。 一方面释放反馈绕组 P2的感应电势对 C51进行充电； 另一方面将 反馈绕组 P2的感应电势提供给光耦 OCl。
当初级绕组 PI能量下降到一定值时， ^^据电感中的电流不能突变的原理， 初级绕组 P1产生一个反相电动势， 以阻止初级电流的下降。 该电流在初级绕 组 P1产生上正下负的感应电动势。反馈绕组 P2产生正脉冲电压通过正反馈回 路， 使三极管 TR1重新导通。 因此， 开关电源便工作在自激振荡状态。
振荡频率主要由变压器 T1的电感量 Lp决定。 自激振荡工作后， 此电路 将进行反激。 MOS管 TR1导通时变压器 T1储能； MOS管 TR1关断时变压器 T1 输出能量， 能量再经稳压输出回路输出， 实现能量的传递。 输出的能量一 路提供给负载，而另一路经基准放大部分采样比较后，经光耦 OC1输入到 PFM 的三极管 TR2的基极， 来控制三极管 TR2基极上的电流， 从而调节 MOS管 TR1和三极管 TR2的通断时间 , 实现电路的反激过程。
以上是本发明电路的整个工作过程。
上述 MOS管 TR1在饱和导通过程中 , 流经初级绕组 PI、 MOS管 TR1的 漏极电流 Id随着时间的增加而增大， 电阻 R4上的压降也增加。
当电压达到 (0.7V+VR3)时，三极管 TR2导通，三极管 TR3的基极电压下降， 从而三极管 TR3导通。 三极管 TR3的集电极电流增大， 三极管 TR2导通能力 增强， 如此循环往复， 最终使三极管 TR2、 TR3进入饱和。
同时， 由于三极管 TR3的导通， MOS管 TR1饱和导通过程中储存在 Ciss 的能量通过三极管 TR3释放到地，放电时间常数很短， MOS管 TR1的关断损 耗很低， 从而使产品的整机效率得到了很大程度上的提高。
当电路工作在输出短路状态时， 由于短路瞬间电流非常大， 导致 Vgl点电 压高， MOS管 TR1导通强度加强， MOS管 TR1的漏极电流 Id增大， 电阻 R4上的压降也增加。
当电压达到 (0.7+VR3)时， 三极管 TR2导通， 三极管 TR3的基极电压下降， 从而三极管 TR3导通， 三极管 TR3的集电极电流增大， 于是三极管 TR2导通 能力增强， 如此循环往复， 最终使三极管 TR2、 TR3进入饱和。 同时， 由于三 极管 TR3的导通， MOS管 TR1饱和导通过程中储存在 Ciss的能量通过三极 管 TR3释放到地， 从而致使 MOS管 TR1进入可靠的截止状态。 MOS管 TR1 的漏极电流 Id接近为零， 从而使短路功耗接近为零。
直至短路状态消失后， 变压器 T1感应电动势反相。 当提供到三极管 TR2 基极的电流小于导通电流时， 三极管 TR2、 TR3关断。 MOS管 TR1的栅极电 压 Vgl迅速回复高位， 则 MOS管 TR1导通， 自动恢复到电路正常的自激振 荡工作模式， 实现电路持续的短路保护。
另外， 本发明对软启动部分作了进一步的改进。 如图 3所示， 软启动电路 由电阻 Rl、 电阻 R8, 电容 C9和二极管 D2组成。
输入端 VIN串接电阻 R1后一路通过电容 C9接地； 另外一路接二极管 D2 的阳极。二极管 D2的阴极通过电阻 R8接地。其中二极管 D2阴极还有一 矣 MOS管栅极。
软启动部分中将快恢复二极管 D2替代如图 2所示现有电路中的电阻 R7。 一般工作情况下， 快恢复二极管 D2的导通内阻 rd«R7。
当电路刚上电从 t = 0开始工作时，输入电压通过电阻 R1对电容 C9充电。 当 C9上的电压达到 0.7V时，快恢复二极管 D2导通。从而开始对 MOS管 TR1 的内部结电容 Ciss充电。
当达到 MOS管 TR1的栅极门限电压 Vth时， MOS管 TR1导通。 此时的 充电时间常数 rdCgs«R7Cgs ( rd为二极管 D2的内阻）。 MOS管 TR1的启动 性能增强， 带容性负载的能力加强。
另外当 Vgl点电位高于 VI点电位时， 由于二极管的单向导通性， 电流不 能反向向前级流动，避免了电量对前级电路的干扰，提高了产品工作时的可靠 性。
通过对软启动电路的改装， 巧妙地运用二极管反向截止的特性有效地避免 了正反馈绕组产生的驱动信号对软启动电路的干扰，大大提升了产品的启动性 能。
下面对本发明图 3所示实施例和图 2所示现有技术的具体实施中的各项参 数进行实验对比：
电源采用的基本参数为： 输入直流电压范围为 9 ~ 18V、 输出为 12V/500mA。 在空载、 轻载及满载的情况下都能够正常工作， 本发明与现有技 术相同的部分采用相同元器件。
如图 5所示， 使用图 2电路在标称满载时， 在 0 ~ 500mA的输出负载范围 里，本发明的输入电压效率明显比图 2电路的要高，在负载电流越小的情况下， 差别越大。
如图 5和图 6所示， MOS管 TR1作为功率开关管， 在稳态标称满载时， 其栅极电压 Vgl的波幅在本发明电路达到 9.62V,而在图 2所示电路里只能达 到 5.52V。
如图 7和图 8所示， MOS管 TR1在稳态标称满载时， 其漏极电压 Vds的 波幅在本发明电路只有 27.4V, 而在图 2所示电路里却达到 32.6V, 所需器件 的耐压值更高。
下表是其他测试指标对比：
最小输入电压 标称输入电压 最大输入电压 测试项目及条件 (9VDC) (12VDC) (18VDC) 单管驱动;双管驱动单管驱动双管驱动单管驱动 i双管驱动 满载 i效率 η(%) 76.9！ 83.9 78.3 86.6 74.3！ 86.9 线性调节率（％)
！ (TYP) 0 0
负 载调节率（％)
！ (TYP) -0.75 -0.42
！纹波（mV) 20！ 15 10 10 10！ 10
！燥声 (mV) 21.6！ 17.6 15 13 12！ 11 最大容性负（uF)
(TYP) 220 1680 \
空载功耗 (Mw) 0.504！ 0.27 \ 0.708 0.312 \ 0.684！ 0.45 I 短路功耗 (W) 1.0359！ 0.126 \ 0.852 0.126 1.7478！ 0.198 I
I 25%-50%-2 ！过冲幅度 (％) 3.16！ 1.97 \ 3.6 1.7 \ 3.36！ 1.47
I 5% ！欠冲幅度 (％) 3.3！ 2.13 3.77 1.87 \ 3.07！ 1.6 1 跳变 ！恢复时间 (ms) 3.26！ 3.24 3.27 3.26 \ 3.27！ 3.28
50-75%-50 ！过冲幅度 (％) 2.53： 1.93 \ 2.63 1.67 i 3.5！ 1.47
% ！欠冲幅度 (％) 2.73！ 2.03 \ 2.8 1.83 \ 3.83！ 1.57 I 跳变 ！恢复时间 (ms) 3.27！ 3.26 \ 3.26 3.28 \ 3.27！ 3.26 1
10%-100%- 、过冲幅度 (％) 10.7！ 7.08 \ 10.3 6.17 \ 12.1！ 5.13
10% ！欠冲幅度 (％) 11； 7.25 \ 11 6.33 i 12.1！ 5.13 I 跳变 ！恢复时间 (ms) 3.33 \ 3.3 \ 3.3 3.33 \ 3.34！ 3.33 I 如图 9所示， 为了进一步改善本发明， 在图 3所示的实施例的基础上， 所 述 MOS管 TR1的源极与电阻 R4连接电流互感器 S1和整流二极管 D5， 电流 互感器 S1的初级绕组 N1的同名端接 MOS管 TR1的源极，次级绕组 N2的同 名端接二极管 D5的阳极。
二极管 D5的阴极接电阻 R4, 电流互感器 S1的两个异名端接地。
其工作原理是：根据初、次级线圏匝数比与电流比关系式： Nl/N2=Is2/Isl , 可知 Is2 = Isl*Nl/N2。
I设取 Nl = 1匝， N2 = 50匝， Isl = 5A, R4 = 1Ω,则 Is2 = Is2*R4 = 5* 1/50 = 0.1A, PR4 = Is22*R4 = 0.12*1 = 0.01W。 而原电路功率 PR4 = Is22*R4 = Isl2*R4 = 52*1 = 25W。
可见此电路的优点是： 产品在满载时短路功耗非常小，仅有 io-2w级别的 功耗。
如图 10所示的实施方式与图 3所示的实施例基本相同， 不同之处在于三 极管 TR3的连接。
本实施例中三极管 TR3的发射极接 MOS管 TR1栅极；基极一路经偏置电 阻 R36接 MOS管 TR1栅极， 另外一路接三极管 TR2集电极； 集电极接三极 管 TR2基极。
三极管 TR2基极经偏置电阻 R3接 MOS管 TR1源极。
电容 C2并联于电阻 R3的两端。
MOS管 TR1的源极经电阻 R4接地。
同样在图 10所示的实施方式中， 也可以在 MOS管 TR1的源极与电阻 R4 之间连接电流互感器 S1和整流二极管 D5， 起到同样的效果。
如图 11所示的实施方式与图 3所示的实施例基本相同， 不同之处在于三 极管 TR3的连接。
本实施例中三极管 TR3的发射极接 MOS管 TR1栅极； 基极一路经偏置 电阻 R36接 MOS管 TR1栅极，另外一 矣三极管 TR2集电极；集电极接 MOS 管 TR1的源极。
MOS管 TR1源极经偏置电阻 R3接三极管 TR2基极。
电容 C2并联于电阻 R3的两端。
MOS管 TR1的源极经电阻 R4接地。
同样在图 11所示的实施方式中，也可以在 MOS管 TR1的源极与电阻 R4 之间连接电流互感器 S1和整流二极管 D5， 起到同样的效果。

Claims

权 利 要 求
1、 一种双三极管电流控制型自振荡反激变换器， 包括软启动部分、 MOS 管 TR1、 变压器 Tl、 脉冲频率调制部分 PFM、 基准放大部分、 隔离光耦 OC1 和稳压输出回路部分；
输入电量经变压器 T1连接输出回路部分；
所述软启动部分连接所述 MOS管 TR1的栅极；
所述 MOS管 TR1的栅极还连接所述脉冲频率调制部分 PFM;
所述脉冲频率调制部分 PFM和稳压输出回路部分之间接所述基准放大部 分和隔离光耦 OC1 , 形成电压负反馈回路；
所述脉冲频率调制部分 PFM主要包括三极管 TR2、 电阻 R3、 电容 C2和 电阻 R4; 所述三极管 TR2的基极通过并联的偏置电阻 R3和电容 C2接所述 MOS管 TR1的源极； 所述 MOS管 TR1的源极通过电阻 R4接地； 其特征在 于： 所述脉冲频率调制部分 PFM中增设三极管电流控制电路， 该三极管电流 控制电路连接在所述 MOS管 TR1和所述三极管 TR2之间， 实现输入端双三 极管电流控制的自激振荡输出。
2、 根据权利要求 1所述的双三极管电流控制型自振荡反激变换器， 其特 征在于： 所述三极管电流控制电路包括三极管 TR3和电阻 R36;
所述三极管 TR3的发射极接所述 MOS管 TR1的栅极， 基极一路经偏置 电阻 R36接所述 MOS管 TR1栅极， 另外一 矣所述三极管 TR2集电极； 集 电极接所述三极管 TR2基极；
所述三极管 TR2基极经偏置电阻 R3接所述 MOS管 TR1的源极； 所述 MOS管 TR1的源极经电阻 R4接地。
3、 根据权利要求 1 所述的双三极管电流控制型自振荡反激变换器， 其特 征在于： 所述三极管电流控制电路包括三极管 TR3和电阻 R36;
所述三极管 TR3的发射极接所述 MOS管 TR1的栅极；基极一路经偏置电 阻 R36接所述 MOS管 TR1栅极， 另外一路接三极管 TR2集电极； 集电极接 所述 MOS管 TR1的源极；
所述 MOS管 TR1的源极经偏置电阻 R3接所述三极管 TR2基极； 所述 MOS管 TR1的源极经电阻 R4接地。
4、 根据权利要求 1所述的双三极管电流控制型自振荡反激变换器， 其特 征在于： 所述三极管电流控制电路包括三极管 TR3、 电阻 R36和电阻 R27; 所述三极管 TR3的发射极接所述 MOS管 TR1栅极； 基极一路经偏置电 阻 R36接所述 MOS管 TR1栅极， 另外一路经偏置电阻 R27接三极管 TR2集 电极； 集电极接所述 MOS管 TR1的源极；
所述 MOS管 TR1的源极经偏置电阻 R3接三极管 TR2基极；
所述 MOS管 TR1的源极经电阻 R4接地。
5、 根据权利要求 2至 4任一项所述的双三极管电流控制型自振荡反激变 换器， 其特征在于： 所述 MOS管 TR1的源极与电阻 R4之间连接电流互感器 S1和续流二极管 D5;
所述电流互感器 S1的初级同名端接所述 MOS管 TR1的源极；
所述电流互感器 S1的次级同名端接所述二极管 D5的阳极；
所述二极管 D5的阴极接电阻 R4; 电流互感器 S 1的两个异名端接地。
6、 根据权利要求 5 所述的双三极管电流控制型自振荡反激变换器， 其特 征在于： 所述偏置电阻 R36的两端并联电容 C34; 偏置电阻 R3的两端并联电 容 C2。
7、 根据权利要求 6所述的双三极管电流控制型自振荡反激变换器， 其特 征在于： 所述 MOS管 TR1为 N沟道型 , 所述三极管 TR2为 NPN型 , 所述三 极管 TR3为 PNP型。
8、 根据权利要求 7所述的双三极管电流控制型自振荡反激变换器， 其特 征在于： 所述 MOS管 TR1的栅极连接稳压二极管 Z1 ;
所述稳压二极管 Z1的阴极接所述 MOS管 TR1的栅极， 阳极接地。
9、 根据权利要求 1 所述的双三极管电流控制型自振荡反激变换器， 其特 征在于： 所述的软启动电路由电阻 Rl、 电阻 R8, 电容 C9和二极管 D2组成； 输入端 VIN串接电阻 R1后一路经电容 C9接地； 另外一路接所述二极管
D2的阳极；
所述二极管 D2 阴极通过电阻 R8接地； 二极管 D2阴极还有一 矣所述 MOS管 TR1的栅极。
10、 根据权利要求 9所述的双三极管电流控制型自振荡反激变换器， 其特 征在于： 所述二极管 D2为快恢复二极管。
PCT/CN2009/070007 2008-04-08 2009-01-04 双三极管电流控制型自振荡反激变换器 WO2009124469A1 (zh)

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