Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04W—WIRELESS COMMUNICATION NETWORKS
  • H04W52/00—Power management, e.g. TPC [Transmission Power Control], power saving or power classes
  • H04W52/04—TPC
  • H04W52/30—TPC using constraints in the total amount of available transmission power
  • H04W52/36—TPC using constraints in the total amount of available transmission power with a discrete range or set of values, e.g. step size, ramping or offsets
  • H04W52/362—Aspects of the step size
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/69—Spread spectrum techniques
  • H04B1/7163—Spread spectrum techniques using impulse radio
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04W—WIRELESS COMMUNICATION NETWORKS
  • H04W52/00—Power management, e.g. TPC [Transmission Power Control], power saving or power classes
  • H04W52/04—TPC
  • H04W52/18—TPC being performed according to specific parameters
  • H04W52/20—TPC being performed according to specific parameters using error rate
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04W—WIRELESS COMMUNICATION NETWORKS
  • H04W52/00—Power management, e.g. TPC [Transmission Power Control], power saving or power classes
  • H04W52/04—TPC
  • H04W52/18—TPC being performed according to specific parameters
  • H04W52/24—TPC being performed according to specific parameters using SIR [Signal to Interference Ratio] or other wireless path parameters
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04W—WIRELESS COMMUNICATION NETWORKS
  • H04W52/00—Power management, e.g. TPC [Transmission Power Control], power saving or power classes
  • H04W52/04—TPC
  • H04W52/18—TPC being performed according to specific parameters
  • H04W52/24—TPC being performed according to specific parameters using SIR [Signal to Interference Ratio] or other wireless path parameters
  • H04W52/241—TPC being performed according to specific parameters using SIR [Signal to Interference Ratio] or other wireless path parameters taking into account channel quality metrics, e.g. SIR, SNR, CIR, Eb/lo
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04W—WIRELESS COMMUNICATION NETWORKS
  • H04W52/00—Power management, e.g. TPC [Transmission Power Control], power saving or power classes
  • H04W52/04—TPC
  • H04W52/30—TPC using constraints in the total amount of available transmission power
  • H04W52/36—TPC using constraints in the total amount of available transmission power with a discrete range or set of values, e.g. step size, ramping or offsets
  • H04W52/367—Power values between minimum and maximum limits, e.g. dynamic range
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/69—Spread spectrum techniques
  • H04B1/7163—Spread spectrum techniques using impulse radio
  • H04B1/7183—Synchronisation
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/69—Spread spectrum techniques
  • H04B2001/6908—Spread spectrum techniques using time hopping

Definitions

• the present invention generally relates to wireless communications, and more specifically, to a system and a method for impulse radio power control.
• impulse radio impulse radio communications systems
• Impulse radio systems emit short pulses approaching a Gaussian monocycle with tightly controlled pulse-to-pulse intervals.
• Impulse radio systems typically use pulse position modulation, which is a form of time modulation where the value of each instantaneous sample of a modulating signal is caused to modulate the position of a pulse in time.
• the pulse-to-pulse interval is varied on a pulse-by-pulse basis by two components: an information component and a pseudo-random code component.
• the pseudo-random code for impulse radio communications is not necessary for energy spreading because the monocycle pulses themselves have an inherently wide bandwidth. Instead, the pseudo-random code of an impulse radio system is used for channelization, energy smoothing in the frequency domain and for interference suppression.
• an impulse radio receiver is a direct conversion receiver with a cross correlator front end.
• the front end coherently converts an electromagnetic pulse train of monocycle pulses to a baseband signal in a single stage.
• the data rate of the impulse radio transmission is typically a fraction of the periodic timing signal used as a time base. Because each data bit modulates the time position of many pulses of the periodic timing signal, this yields a modulated, coded timing signal that comprises a train of identically shaped pulses for each single data bit.
• the impulse radio receiver integrates multiple pulses to recover the transmitted information. In a multi-user environment, impulse radio depends, in part, on processing gain to achieve rejection of unwanted signals.
• impulse radio Because of the extremely high processing gain achievable with impulse radio, much higher dynamic ranges are possible than are commonly achieved with other spread spectrum methods, some of which must use power control in order to have a viable system. Further, if power is kept to a minimum in an impulse radio system, this will allow closer operation in co-site or nearly co-site situations where two impulse radios must operate concurrently, or where an impulse radio and a narrow band radio must operate close by one another and share the same band.
• power control may be used to reduce the multi-user background noise to improve the number of channels available and the aggregate traffic density of the area.
• power control generally refers to adjusting the transmitter output power to the minimum necessary power to achieve acceptable signal reception at an impulse radio receiver. If the received signal power drops too low, the transmitter power should be increased. Conversely, if the received signal power rises too high, the transmitter power should be decreased. This potentially reduces interference with other services and increases the channelization (and thus, capacity) available to a multi-user impulse radio system.
• Kolenchery et al. describe the combined use of a variable data rate with power control.
• Kolenchery et al. propose a system that uses closed loop power control with an open loop adjustment of power associated with each change in data rate to maintain constant signal to noise during the transient event of changing the data rate.
• the system proposed by Kolenchery et al. does not make full use of the properties of UWB.
• Kolenchery et al. do not describe a system and method for measuring signal quality and applying such a system and method to power control.
• the present invention is directed to a system and method for impulse radio power control.
• a first transceiver transmits an impulse radio signal to a second transceiver.
• a power control update is calculated according to a performance measurement of the impulse radio signal received at the second transceiver.
• the transmitter power of either transceiver is adjusted according to the power control update.
• An advantage of the current invention is that interference is reduced. This is particularly important where multiple impulse radios are operating in close proximity (e.g. , a densely utilized network), and their transmissions interfere with one another. Reducing the transmitter power of each radio to a level that produces satisfactory reception increases the total number of radios that can operate in an area without excess interference.
• impulse radios can be more energy efficient. Reducing transmitter power to only the level required to produce satisfactory reception allows a reduction in the total power consumed by the transceiver, and thereby increases its efficiency.
• bit error rate is employed alone or in combination to form a power control update. These performance measurements vary by accuracy and time required to achieve an update. An appropriate performance measurement can be chosen based on the particular environment and application.
• the output power of a transceiver is controlled by controlling the quantity N tra ⁇ n of pulses according to the power control update.
• the quantity N tra ⁇ n of pulses includes a quantity N pe ⁇ od of periods, and each period includes a quantity N pulses . per - penod of pulses
• the output power of a transceiver can be controlled by controlling the quantity N penod of periods.
• the output power can be controlled by controlling the quantity N pulses . per . per ⁇ od of pulses.
• the power control update is determined at the second transceiver and then sent from the second transceiver to the first transceiver.
• the second transceiver sends at least one performance measurement to the first transceiver, and the first transceiver then determines the power control update based on the performance measurement(s).
• Fig. 1 A illustrates a representative Gaussian Monocycle waveform in the time domain
• Fig. IB illustrates the frequency domain amplitude of the Gaussian
• Fig. 2A illustrates a pulse train comprising pulses as in Fig. 1A;
• Fig. 2B illustrates the frequency domain amplitude of the waveform of Fig. 2A
• Fig. 3 illustrates the frequency domain amplitude of a sequence of time coded pulses
• Fig. 4 illustrates a typical received signal and interference signal
• Fig. 5A illustrates a typical geometrical configuration giving rise to multipath received signals
• Fig. 5B illustrates exemplary multipath signals in the time domain
• Fig. 6 illustrates a representative impulse radio transmitter functional diagram that does not include power control
• Fig. 7 illustrates a representative impulse radio receiver functional diagram that does not include power control
• Fig. 8 A illustrates a representative received pulse signal at the input to the correlator
• Fig 8B illustrates a sequence of representative impulse signals in the correlation process
• Fig 8C illustrates the potential locus of results as a function of the various potential template time positions
• Fig. 9 illustrates an example environment of an impulse radio communication system
• Fig 10 is an exemplary flow diagram of a two transceiver system employing power control according to one embodiment of the present invention.
• Fig. 11 is an exemplary diagram of an impulse receiver including power control functions according to one embodiment of the present invention.
• Fig. 12 is a detailed representation of one embodiment of the detection process in Fig. 10;
• Fig. 13 is a detailed block diagram of one embodiment of the signal evaluation process in Fig. 11 ;
• Fig. 14 illustrates an alternate processing method for Fig. 13
• Fig. 15 is a detailed block diagram of one embodiment of the signal evaluation process in Fig. 11 ;
• Fig. 16 illustrates an alternate processing method for Fig. 15;
• Fig. 17 illustrates a lock detection and signal combination function used by the signal evaluation function of Fig. 11;
• Fig. 18 is a flowchart that describes a method of power control according to the present invention
• Fig. 19 is a flowchart that describes controlling the transmitter power of a first transceiver according to the power control updates
• Fig. 20 is a flow diagram illustrating the control dynamics of one embodiment of the present invention
• Fig. 21 is a flow diagram illustrating the control dynamics of a system including Signal to Noise Ratio measurement
• Fig. 22 is a flow diagram illustrating the control dynamics of a system including Bit Error Rate measurement
• Fig. 23 is a flow diagram illustrating the control dynamics of a system employing log mapping of Bit Error Rate measurements
• Fig.24 is a flow diagram illustrating the control dynamics of a system that incorporates auto power control and cross power control;
• Fig.25 illustrates an embodiment of a power control algorithm employing auto-control with power level messaging
• Fig. 26 illustrates an embodiment of a power control algorithm where auto-control and cross control are implemented in combination
• Fig. 27 illustrates two signals having different pulse peak power
• Fig. 28 illustrates periods of two subcarriers
• Fig. 29 is a flow diagram illustrating the control dynamics of a system employing gain expansion power control.
• This section is directed to technology basics and provides the reader with an introduction to impulse radio concepts, as well as other relevant aspects of communications theory.
• This section includes subsections relating to waveforms, pulse trains, coding for energy smoothing and channelization, modulation, reception and demodulation, interference resistance, processing gain, capacity, multipath and propagation, distance measurement, and qualitative and quantitative characteristics of these concepts. It should be understood that this section is provided to assist the reader with understanding the present invention, and should not be used to limit the scope of the present invention.
• Impulse radio refers to a radio system based on short, low duty cycle pulses.
• An ideal impulse radio waveform is a short Gaussian monocycle. As the name suggests, this waveform attempts to approach one cycle of radio frequency (RF) energy at a desired center frequency. Due to implementation and other spectral limitations, this waveform may be altered significantly in practice for a given application. Most waveforms with enough bandwidth approximate a Gaussian shape to a useful degree.
• RF radio frequency
• Impulse radio can use many types of modulation, including AM, time shift (also referred to as pulse position) and M-ary versions.
• the time shift method has simplicity and power output advantages that make it desirable.
• the time shift method is used as an illustrative example.
• the pulse-to-pulse interval can be varied on a pulse-by-pulse basis by two components: an information component and a pseudo-random code component.
• an information component e.g., a digital signal processor
• a pseudo-random code component e.g., a pseudo-random code component
• conventional spread spectrum systems make use of pseudo-random codes to spread the normally narrow band information signal over a relatively wide band of frequencies.
• a conventional spread spectrum receiver correlates these signals to retrieve the original information signal.
• the pseudo- random code for impulse radio communications is not necessary for energy spreading because the monocycle pulses themselves have an inherently wide bandwidth. Instead, the pseudo-random code is used for channelization, energy smoothing in the frequency domain, resistance to interference, and reducing the interference potential to nearby receivers.
• the impulse radio receiver is typically a direct conversion receiver with a cross correlator front end in which the front end coherently converts an electromagnetic pulse train of monocycle pulses to a baseband signal in a single stage.
• the baseband signal is the basic information signal for the impulse radio communications system. It is often found desirable to include a subcarrier with the baseband signal to help reduce the effects of amplifier drift and low frequency noise.
• the subcarrier that is typically implemented alternately reverses modulation according to a known pattern at a rate faster than the data rate. This same pattern is used to reverse the process and restore the original data pattern just before detection.
• This method permits alternating current (AC) coupling of stages, or equivalent signal processing to eliminate direct current (DC) drift and errors from the detection process. This method is described in detail in U.S. Patent No. 5,677,927 to Fullerton et al.
• each data bit typically time position modulates many pulses of the periodic timing signal. This yields a modulated, coded timing signal that comprises a train of identically shaped pulses for each single data bit.
• the impulse radio receiver integrates multiple pulses to recover the transmitted information.
• Impulse radio refers to a radio system based on short, low duty cycle pulses.
• the resulting waveform approaches one cycle per pulse at the center frequency.
• each pulse consists of a burst of cycles usually with some spectral shaping to control the bandwidth to meet desired properties such as out of band emissions or in-band spectral flatness, or time domain peak power or burst off time attenuation.
• Fig. 1 A This waveform is representative of the transmitted pulse produced by a step function into an ultra-wideband antenna.
• the basic equation normalized to a peak value of 1 is as follows:
• ⁇ is a time scaling parameter
• t is time
• f mo / is me waveform voltage
• e is the natural logarithm base.
• the center frequency (f), or frequency of peak spectral density is:
• pulses may be produced by methods described in the patents referenced above or by other methods that are known to one of ordinary skill in the art. Any practical implementation will deviate from the ideal mathematical model by some amount. In fact, this deviation from ideal may be substantial and yet yield a system with acceptable performance. This is especially true for microwave implementations, where precise waveform shaping is difficult to achieve.
• These mathematical models are provided as an aid to describing ideal operation and are not intended to limit the invention. In fact, any burst of cycles that adequately fills a given bandwidth and has an adequate on-off attenuation ratio for a given application will serve the purpose of this invention.
• Impulse radio systems can deliver one or more data bits per pulse; however, impulse radio systems more typically use pulse trains, not single pulses, for each data bit. As described in detail in the following example system, the impulse radio transmitter produces and outputs a train of pulses for each bit of information.
• Prototypes built by the inventors have pulse repetition frequencies including 0.7 and 10 megapulses per second (Mpps, where each megapulse is 10 6 pulses).
• Figs. 2A and 2B are illustrations of the output of a typical 10 Mpps system with uncoded, unmodulated, 0.5 nanosecond (ns) pulses 102.
• Fig. 2A shows a time domain representation of this sequence of pulses 102.
• Fig 2B which shows 60 MHZ at the center of the spectrum for the waveform of Fig. 2A, illustrates that the result of the pulse train in the frequency domain is to produce a spectrum comprising a set of lines 204 spaced at the frequency of the 10 Mpps pulse repetition rate.
• the envelope of the line spectrum follows the curve of the single pulse spectrum 104 of Fig. IB.
• the power of the pulse train is spread among roughly two hundred comb lines. Each comb line thus has a small fraction of the total power and presents much less of an interference problem to receiver sharing the band.
• impulse radio systems typically have very low average duty cycles resulting in average power significantly lower than peak power.
• the duty cycle of the signal in the present example is 0.5%, based on a 0.5 ns pulse in a 100 ns interval.
• Fig. 3 is a plot illustrating the impact of a pseudo-noise (PN) code dither on energy distribution in the frequency domain (A pseudo-noise, or PN code is a set of time positions defining the pseudo-random positioning for each pulse in a sequence of pulses).
• PN code is a set of time positions defining the pseudo-random positioning for each pulse in a sequence of pulses.
• Fig. 3 when compared to Fig. 2B, shows that the impact of using a PN code is to destroy the comb line structure and spread the energy more uniformly. This structure typically has slight variations which are characteristic of the specific code used.
• the PN code also provides a method of establishing independent communication channels using impulse radio.
• PN codes can be designed to have low cross correlation such that a pulse train using one code will seldom collide on more than one or two pulse positions with a pulses train using another code during any one data bit time. Since a data bit may comprise hundreds of pulses, this represents a substantial attenuation of the unwanted channel.
• any aspect of the waveform can be modulated to convey information.
• Amplitude modulation, phase modulation, frequency modulation, time shift modulation and M-ary versions of these have been proposed. Both analog and digital forms have been implemented. Of these, digital time shift modulation has been demonstrated to have various advantages and can be easily implemented using a correlation receiver architecture.
• Digital time shift modulation can be implemented by shifting the coded time position by an additional amount (that is, in addition to PN code dither) in response to the information signal. This amount is typically very small relative to the PN code shift. In a 10 Mpps system with a center frequency of 2 GHz., for example, the PN code may command pulse position variations over a range of 100 ns; whereas, the information modulation may only deviate the pulse position by l50 ps.
• each pulse is delayed a different amount from its respective time base clock position by an individual code delay amount plus a modulation amount, where n is the number of pulses associated with a given data symbol digital bit. Modulation further smooths the spectrum, minimizing structure in the resulting spectrum.
• impulse radios are able to perform in these environments, in part, because they do not depend on receiving every pulse.
• the impulse radio receiver performs a correlating, synchronous receiving function (at the RF level) that uses a statistical sampling and combining of many pulses to recover the transmitted information.
• Impulse radio receivers typically integrate from 1 to 1000 or more pulses to yield the demodulated output.
• the optimal number of pulses over which the receiver integrates is dependent on a number of variables, including pulse rate, bit rate, interference levels, and range. 1.6. Interference Resistance
• the PN coding also makes impulse radios highly resistant to interference from all radio communications systems, including other impulse radio transmitters. This is critical as any other signals within the band occupied by an impulse signal potentially interfere with the impulse radio. Since there are currently no unallocated bands available for impulse systems, they must share spectrum with other conventional radio systems without being adversely affected.
• the PN code helps impulse systems discriminate between the intended impulse transmission and interfering transmissions from others.
• Fig.4 illustrates the result of a narrow band sinusoidal interference signal 402 overlaying an impulse radio signal 404.
• the input to the cross correlation would include the narrow band signal 402, as well as the received ultrawide-band impulse radio signal 404.
• the input is sampled by the cross correlator with a PN dithered template signal 406. Without PN coding, the cross correlation would sample the interfering signal 402 with such regularity that the interfering signals could cause significant interference to the impulse radio receiver.
• the transmitted impulse signal is encoded with the PN code dither (and the impulse radio receiver template signal 406 is synchronized with that identical PN code dither)
• the correlation samples the interfering signals pseudo-randomly.
• the samples from the interfering signal add incoherently, increasing roughly according to square root of the number of samples integrated; whereas, the impulse radio samples add coherently, increasing directly according to the number of samples integrated.
• integrating over many pulses overcomes the impact of interference.
• Impulse radio is resistant to interference because of its large processing gain.
• processing gain which quantifies the decrease in channel interference when wide-band communications are used, is the ratio of the bandwidth of the channel to the bit rate of the information signal.
• a direct sequence spread spectrum system with a 10 kHz information bandwidth and a 10 MHZ channel bandwidth yields a processing gain of 1000 or 30 dB.
• far greater processing gains are achieved with impulse radio systems, where for the same 10 KHz information bandwidth is spread across a much greater 2 GHz. channel bandwidth, the theoretical processing gain is 200,000 or 53 dB.
• V 2 tot is the total interference signal to noise ratio variance, at the receiver; N is the number of interfering users; ⁇ 2 is the signal to noise ratio variance resulting from one of the interfering signals with a single pulse cross correlation; and Z is the number of pulses over which the receiver integrates to recover the modulation.
• This relationship suggests that link quality degrades gradually as the number of simultaneous users increases. It also shows the advantage of integration gain. The number of users that can be supported at the same interference level increases by the square root of the number of pulses integrated.
• impulse radio is its resistance to multipath fading effects.
• Conventional narrow band systems are subject to multipath through the Rayleigh fading process, where the signals from many delayed reflections combine at the receiver antenna according to their relative phase. This results in possible summation or possible cancellation, depending on the specific propagation to a given location. This also results in potentially wild signal strength fluctuations in mobile applications, where the mix of multipath signals changes for every few feet of travel.
• Figs. 5 A and 5B Three propagation paths are shown.
• the direct path is the shortest. It represents the straight line distance between the transmitter and the receiver.
• Path 1 represents a grazing multipath reflection, which is very close to the direct path.
• Path 2 represents a distant multipath reflection.
• elliptical (or, in space, ellipsoidal) traces that represent other possible locations for reflections with the same time delay.
• Fig. 5B represents a time domain plot of the received waveform from this multipath propagation configuration.
• This figure comprises three doublet pulses as shown in Fig. 1 A.
• the direct path signal is the reference signal and represents the shortest propagation time.
• the path 1 signal is delayed slightly and actually overlaps and enhances the signal strength at this delay value. Note that the reflected waves are reversed in polarity.
• the path 2 signal is delayed sufficiently that the waveform is completely separated from the direct path signal. If the correlator template signal is positioned at the direct path signal, the path 2 signal will produce no response. It can be seen that only the multipath signals resulting from very close reflectors have any effect. The bulk of the multipath signals, which are substantially delayed, are removed from the correlation process and are ignored.
• the multipath signals delayed less than one quarter wave are the only signals that will attenuate the direct path signal. This is the reflection from the first Fresnel zone, and this property is shared with narrow band signals; however, impulse radio is highly resistant to all other Fresnel zone reflections. The ability to avoid the highly variable attenuation from multipath gives impulse radio significant performance advantages.
• Impulse systems can measure distances to extremely fine resolution because of the absence of ambiguous cycles in the waveform.
• Narrow band systems are limited to the modulation envelope and cannot easily distinguish precisely which RF cycle is associated with each data bit because the cycle-to-cycle amplitude differences are so small they are masked by link or system noise. Since the impulse radio waveform has no multi-cycle ambiguity, this allows positive determination of the waveform position to less than a wavelength - potentially, down to the noise floor of the system.
• This time position measurement can be used to measure propagation delay to determine link distance, and once link distance is known, to transfer a time reference to an equivalently high degree of precision.
• the inventors of the present invention have built systems that have shown the potential for centimeter distance resolution, which is equivalent to about 30 ps of time transfer resolution. See, for example, commonly owned, co-pending applications 09/045,929, filed March 23, 1998, titled “Ultrawide-Band Position Determination System and Method", and 09/083,993, filed May 26, 1998, titled “System and Method for Distance Measurement by Inphase and Quadrature Signals in a Radio System", both of which are incorporated herein by reference.
• the transmitter 602 comprises a time base 604 that generates a periodic timing signal 606.
• the time base 604 typically comprises a voltage controlled oscillator (NCO), or the like, having a high timing accuracy and low jitter, on the order of picoseconds (ps).
• the voltage control to adjust the VCO center frequency is set at calibration to the desired center frequency used to define the transmitter's nominal pulse repetition rate.
• the periodic timing signal 606 is supplied to a precision timing generator 608.
• the precision timing generator 608 supplies synchronizing signals 610 to the code source 612 and utilizes the code source output 614 together with an internally generated subcarrier signal (which is optional) and an information signal 616 to generate a modulated, coded timing signal 618.
• the code source 612 comprises a storage device such as a random access memory (RAM), read only memory (ROM), or the like, for storing suitable P ⁇ codes and for outputting the P ⁇ codes as a code signal 614.
• RAM random access memory
• ROM read only memory
• maximum length shift registers or other computational means can be used to generate the P ⁇ codes.
• An information source 620 supplies the information signal 616 to the precision timing generator 608.
• the information signal 616 can be any type of intelligence, including digital bits representing voice, data, imagery, or the like, analog signals, or complex signals.
• a pulse generator 622 uses the modulated, coded timing signal 618 as a trigger to generate output pulses.
• the output pulses are sent to a transmit antenna 624 via a transmission line 626 coupled thereto.
• the output pulses are converted into propagating electromagnetic pulses by the transmit antenna 624.
• the electromagnetic pulses are called the emitted signal, and propagate to an impulse radio receiver 702, such as shown in Fig. 7, through a propagation medium, such as air, in a radio frequency embodiment.
• the emitted signal is wide-band or ultrawide-band, approaching a monocycle pulse as in Fig. 1A.
• the emitted signal can be spectrally modified by filtering of the pulses. This bandpass filtering will cause each monocycle pulse to have more zero crossings (more cycles) in the time domain.
• the impulse radio receiver can use a similar waveform as the template signal in the cross correlator for efficient conversion.
• receiver An exemplary embodiment of an impulse radio receiver (hereinafter called the receiver) for the impulse radio communication system is now described with reference to Fig. 7.
• the receiver 702 comprises a receive antenna 704 for receiving a propagated impulse radio signal 706.
• a received signal 708 is input to a cross correlator or sampler 710 via a receiver transmission line, coupled to the receive antenna 704, and producing a baseband output 712.
• the receiver 702 also includes a precision timing generator 714, which receives a periodic timing signal 716 from a receiver time base 718. This time base 718 is adjustable and controllable in time, frequency, or phase, as required by the lock loop in order to lock on the received signal 708.
• the precision timing generator 714 provides synchronizing signals 720 to the code source 722 and receives a code control signal 724 from the code source 722.
• the precision timing generator 714 utilizes the periodic timing signal 716 and code control signal 724 to produce a coded timing signal 726.
• the template generator 728 is triggered by this coded timing signal 726 and produces a train of template signal pulses 730 ideally having waveforms substantially equivalent to each pulse of the received signal 708.
• the code for receiving a given signal is the same code utilized by the originating transmitter to generate the propagated signal.
• the timing of the template pulse train matches the timing of the received signal pulse train, allowing the received signal 708 to be synchronously sampled in the correlator 710.
• the correlator 710 ideally comprises a multiplier followed by a short term integrator to sum the multiplier product over the pulse interval.
• the output of the correlator 710 is coupled to a subcarrier demodulator 732, which demodulates the subcarrier information signal from the subcarrier.
• the purpose of the optional subcarrier process when used, is to move the information signal away from DC (zero frequency) to improve immunity to low frequency noise and offsets.
• the output of the subcarrier demodulator is then filtered or integrated in the pulse summation stage 734.
• a digital system embodiment is shown in Fig. 7.
• a sample and hold 736 samples the output 735 of the pulse summation stage 734 synchronously with the completion of the summation of a digital bit or symbol.
• the output of sample and hold 736 is then compared with a nominal zero (or reference) signal output in a detector stage 738 to determine an output signal 739 representing the digital state of the output voltage of sample and hold 736.
• the baseband signal 712 is also input to a lowpass filter 742 (also referred to as lock loop filter 742).
• a control loop comprising the lowpass filter 742, time base 718, precision timing generator 714, template generator 728, and correlator 710 is used to generate an error signal 744.
• the error signal 744 provides adjustments to the adjustable time base 718 to time position the periodic timing signal 726 in relation to the position of the received signal 708.
• FIGS. 8A-8C illustrate the cross correlation process and the correlation function.
• Fig. 8 A shows the waveform of a template signal.
• Fig. 8B shows the waveform of a received impulse radio signal at a set of several possible time offsets.
• Fig. 8C represents the output of the correlator (multiplier and short time integrator) for each of the time offsets of Fig. 8B .
• this graph does not show a waveform that is a function of time, but rather a function of time-offset.
• Fig. 9 depicts an example communications environment within which the present invention is used.
• Two or more impulse radio transceivers 902 A, 902B communicate with one another, possibly in the presence of an interfering transmitter 908.
• Each transceiver 902 A, 902B includes an impulse radio receiver 702 and an impulse radio transmitter 602.
• Fig. 9 depicts two transceivers 902A and 902B, separated by a distance dl .
• transmitter 602A transmits a signal SI that is received by receiver 702B.
• Transmitter 602B transmits a signal S2 that is received by receiver 702A.
• Interfering transmitter 908, if present, transmits an interfering signal S3 that is received by both receiver 702A and receiver 702B.
• Interfering transmitter 908 is situated a distance d2 from transceiver 902B.
• the output power of transmitters 602 A. 602B is adjusted, according to a preferred embodiment of the present invention, based on a performance measurement(s) of the received signals.
• the output power of transmitter 602B is adjusted based on a performance measurement of signal S2 as received by receiver 702 A.
• the output power of transmitter 602B is adjusted based on a performance measurement of signal SI received by receiver 702B. In both cases, the output power of transmitter 602B is increased when the performance measurement of the received signal drops below a threshold, and is decreased when the performance measurement rises above a threshold.
• Power control refers to the control of the output power of a transmitter. However, it is noted that this is usually implemented as a voltage control proportional to the output signal voltage.
• Different measurements of performance can be used as the basis for calculating a power control update.
• examples of such performance measurements include signal strength, signal-to-noise ratio (SNR), and bit error rate (BER), used either alone or in combination.
• SNR signal-to-noise ratio
• BER bit error rate
• Transceiver 902 can represent any transceiver employing impulse radio technology (for examples, see U.S. Patent No.
• Transceiver 902 can be a hand-held unit, or mounted in some fashion, e.g., a transceiver mounted in a base station.
• transceiver 902A can represent a hand-held phone communicating a transceiver 902B that is part of a base station.
• both transceivers 902A and 902B can represent hand-held phones communicating with each other.
• a plethora of further alternatives are envisioned.
• Interfering transmitter 908 includes transmitter 910 that transmits electromagnetic energy in the same or a nearby frequency band as that used by transceivers 902 A and 902B, thereby possibly interfering with the communications of transceivers 902A and 902B.
• Interfering transmitter 908 might also include a receiver, although the receiver function does not impact interference analysis.
• interfering transmitter 908 could represent an impulse radio communicating with another impulse radio (not shown).
• interfering transmitter 908 could represent any arbitrary transmitter that transmits electromagnetic energy in some portion of the frequency spectrum used by transceivers 902. Those skilled in the art will recognize that many such transmitters can exist, given the ultra-wideband nature of the signals transmitted by transceivers 902.
• interfering transmitter 908 represents an impulse radio transceiver similar in design to transceivers 902 A and 902B.
• Lowering the output power of interfering transmitter 908 reduces the extent to which S3 interferes with the communication between transceivers 902A and 902B.
• lowering the power of transmitters 602A and 602B reduces the extent to which S 1 and S2 interfere with the communications of transmitter 908.
• each transmitter (602A, 602B, and 910 in those situations where interfering transmitter 908 represents an impulse radio) maintains its output power to achieve a satisfactory signal reception.
• the present invention is therefore particularly well suited to a crowded impulse radio environment.
• impulse radio power control methods utilize a performance measurement indicative of the quality of the communications process where the quality is power dependent. This quality measurement is compared with a quality reference in order to determine a power control update.
• Various performance measurements can be used, individually or in combination. Each has slightly different characteristics, which can be utilized in different combinations to construct an optimum system for a given application. Specific performance measurements that are discussed below include signal strength, signal to noise ratio (SNR), and bit error rate (BER). These performance measurements are discussed in an idealized embodiment. However, great accuracy is generally not required in the measurement of these values. Thus, signals approximating these quantities can be substituted as equivalent. Other performance measurements related to these or equivalent to these would be apparent to one skilled in the relevant art. Accordingly, the use of other measurements of performance are within the spirit and scope of the present invention.
• Fig. 10 illustrates a typical two transceiver system comprising transceiver 902A and transceiver 902B and utilizing power control according to an embodiment of the present invention.
• receiver 702A receives the transmission 1008 from transmitter 602B of transceiver 902B.
• Signal evaluation function 1011 A evaluates the signal quality, and quality measurement(s) 1012A are provided to the power control algorithm 1014A.
• Power control algorithm 1014A determines a power control update 1016 according to the current received signal quality measurement(s) 1012A determined by signal evaluation function 1011 A.
• This update 1016 is added to the signal data stream in the transmitter data multiplexer 1018A and then transmitted via transmitter 602 A to transceiver 902B.
• Receiver 702B of transceiver 902B receives a data stream and demultiplexer 1020B separates the user data and power control command 1016, sending the power control command
• Transmitter 602B (or power control function 1126) then adjusts the transmission output level of signal 1008 according to the power control command, which is based on the received signal quality measurement(s) 1012A determined by transceiver 902A.
• a similar control loop operates to control transmitter 602A according to the received signal quality measurement(s) 1012B determined by signal evaluation function 101 IB of transceiver 902B.
• Fig. 11 illustrates a transceiver 902 modified to measure signal strength, SNR, and BER according to an embodiment of the present invention.
• an originating transmitter transmits the RF signal 706, which is received by the antenna 704.
• the resulting received signal 708 is then provided to the correlator 710 where it is multiplied according to the template signal 730 and then short term integrated (or alternatively sampled) to produce a baseband output 712.
• This baseband output is provided to the optional subcarrier demodulator 732, which demodulates a subcarrier as applied to the transmitted signal 706.
• This output is then long term integrated in the pulse summation stage 734, which is typically an integrate and dump stage that produces a ramp shape output waveform when the receiver 702 is receiving a transmitted signal 706, or is typically a random walk type waveform when receiving pure noise.
• This output 735 (after it is sampled by sample and hold state 736) is fed to a detector 738 having an output 739, which represents the detection of the logic state of the transmitted signal 706.
• the output of the correlator 710 is also coupled to a lock loop comprising a lock loop filter 742, an adjustable time base 718, a precision timing generator 714, a template generator 728, and the correlator 710.
• the lock loop maintains a stable quiescent operating point on the correlation function in the presence of variations in the transmitter time base frequency and variations due to Doppler effects.
• the adjustable time base 718 drives the precision timing generator 714, which provides timing to the code generator 722, which in turn, provides timing commands back to the timing generator 714 according to the selected code.
• the timing generator 714 then provides timing signals to the template generator 728 according to the timing commands, and the template generator 728 generates the proper template waveform 730 for the correlation process. Further examples and discussion of these processes can be found in the patents incorporated by reference above.
• coding is optional. Accordingly, it should be appreciated that the present invention covers non-coded implementations that do not incorporate code source 722.
• the output 735 of the pulse summation stage 734 is sampled by the sample and hold stage 736 producing an output 1102 which is then processed by a signal evaluation stage 1011 that determines a measure of the signal strength 1106, received noise 1108, and SNR 1110.
• These values are passed to the power control algorithm 1014, which may combine this information with a BER measurement 1112 provided by a BER evaluation function 1116.
• the power control algorithm 1014 generates a power control update 1016 value according to one or more of the performance measurements. This value is combined with the information signal 616 and sent to the transceiver which is originating the received signal 706.
• One method of combining this information is to divide the data stream into time division blocks using a multiplexer 1018.
• a portion of the data stream 1122 contains user data (i.e., information signal 616) and a portion contains control information, which includes power control update information 1016.
• the combined data stream 1122 is then provided to the transmitter precision timing generator 608, which may optionally include a subcarrier modulation process.
• This timing generator is driven by a transmitter time base 604 and interfaces with a code generator 612, which provides pulse position commands according to a PN code.
• the timing generator 608 provides timing signals 618 to the pulse generator 622, which generates pulses 626 of proper amplitude and waveform according to the timing signals 618. These pulses are then transmitted by the antenna 624.
• BER 1112 is a measure of signal quality that is related to the ratio of error bits to the total number of bits transmitted.
• the system functions such as power command 1124 and power control 1126 can be implemented into either the transmitter 602 or receiver 702 of a transceiver, at the convenience of the designer.
• power control 1126 is shown as being part of transmitter 602 in Fig. 10.
• the transceiver originating the RF signal 706 has a similar architecture.
• the received data stream 739 contains both user data and power control commands, which are intended to control the pulse generator 622.
• These power control commands are selected from the data stream by a power command function 1124, which includes the function of receive data demultiplexer 1020, and delivered to a power control function 1126 that controls the output power of the pulse generator 622.
• the output 1102 of the sample and hold stage 736 is evaluated to determine signal performance criteria necessary for calculation of power control updates 1016.
• the signal performance criteria can include signal strength, noise, SNR and/or BER.
• the output 735 of the pulse summation stage 734 is provided to the input of the sample and hold 736, which is clocked by a sample clock signal 1202 at the end of the integration period (pulse summation period) for a data bit.
• the output 1102 of this sample and hold 736. is supplied to an averaging function 1204, which determines the average value 1206 of this signal 1102.
• This average function 1204 may be a running average, a single pole low pass filter, a simple RC filter (a filter including a resistor(s) and capacitor(s)), or any number of equivalent averaging functions as would be known by one of ordinary skill in the art.
• This average value 1206 represents the DC (direct current) value of the output 1102 of sample and hold 736 and is used as the reference for comparator 1208 in the determination of the digital value of the instant signal which is output as Received Data 739.
• the advantage of averaging function 1204 is to eliminate DC offsets in the circuits leading up to sample and hold 736. This function, however, depends on a relatively equal number of ones and zeroes in the data stream.
• An alternative method is to evaluate the average only when no signal is in lock, as evidenced by low signal strength, and then to hold this value when a signal is in lock. This will be discussed later in detail with reference to Fig. 17. This depends on the assumption that the DC offset will be stable over the period of the transmission.
• a further alternative is to build low offset circuits such that a fixed value, e.g. zero, may be substituted for the average. This is potentially more expensive, but has no signal dependencies.
• a fourth alternative is to split the difference between the average voltage detected as a data "one” and the average voltage detected as a data "zero" to determine a reference value for bit comparison. This difference is available from a signal strength measurement process, which is now described in greater detail in the discussion of Fig. 13
• This function determines signal strength by measuring the difference between the average voltage associated with a digital "one” and the average voltage associated with a digital “zero” .
• Noise is determined by measuring the variation of these signals, and "signal to noise” is determined by finding the ratio between the signal strength and the noise.
• Fig. 13 includes two signal paths, each for determining the average characteristics of the output voltage associated with a detected digital "one" or “zero” respectively.
• the upper path comprising switch 1302, average function 1304, square function 1306, filter 1308, and square root function 1310 operates when the receive data detects a digital "one.”
• the lower path comprising switch 1312, average function 1314, square function 1316, filter 1318, and square root function 1320 operates when the receive data detects a digital "zero" according to inverter 1322. It would be appreciated by one skilled in the art that multiple such paths may be implemented corresponding to multiple states of modulation, should such multiple states be implemented in the particular transceiver system. It should also be noted that a single path might be sufficient for many applications, resulting in possible cost savings with potentially some performance degradation.
• the output 1102 of the sample and hold 736 is fed to either average function 1304 or average function 1314, according to the receive data 739 and inverter 1322, which determines whether the instant signal summation (i.e., the instant of receive data 739) is detected as a "one" or a "zero". If the signal is detected as a digital "one", switch 1302 is closed and average function 1304 receives this signal, while average function 1314 receives no signal and holds its value. If the signal is detected as a digital "zero”, switch 1312 is closed and average function 1314 receives this signal, while average function 1304 receives no signal and holds its value.
• Average functions 1304 and 1314 determine the average value of their respective inputs over the number of input samples when their respective switch is closed. This is not strictly an averaging over time, but an average over the number of input samples. Thus, if there are more ones than zeroes in a given time interval, the average for the ones would reflect the sum of the voltage values for the ones over that interval divided by the number of ones detected in that interval rather than simply dividing by the length of the interval or number of total samples in the interval. Again this average may be performed by running average, or filter elements modified to be responsive to the number of samples rather than time.
• the average over the number of samples represents the best mode in that it corrects for an imbalance between the number of ones and zeroes
• a simple average over time or filter over time may be adequate for many applications.
• a number of averaging functions including, but not limited to, running average, boxcar average, low pass filter, and others can be used or easily adapted to be used in a manner similar to the examples by one of ordinary skill in the art.
• averaging functions 1304 and 1314 are combined to achieve a signal strength measurement 1324.
• the voltage associated with digital "one" is positive, and the voltage associated with digital zero is negative, thus the subtraction indicated in the diagram, is equivalent to a summation of the two absolute values of the voltages. It should also be noted that this summation is equal to twice the average of these two values. A divide by two at this point would be important only in a definitional sense as this factor will be accommodated by the total loop gain in the power control system.
• the purpose of square functions 1306 and 1316, filters 1308 and 1318, and square root functions 1310, 1320 shall be described below in the following section relating to noise measurements.
• Fig. 15 and Fig. 13 illustrate a noise measurement process in accordance with an embodiment of the present invention.
• This noise measurement process is contained within the signal evaluation function 1011 of Fig. 11.
• the noise measurement is combined with the signal strength measurement to derive a signal to noise measurement 1110.
• the samples from sample and hold 736 are evaluated for statistical standard deviation, i.e. the RMS (root mean square) AC (alternating current) voltage. This value is then averaged by an average function to provide a stable measure of the noise. The averaged value can then be used as an initial value for the noise after a signal is captured and locked.
• the output 1102 of sample and hold 736 is averaged in the average function 1204 to remove any DC offset that may be associated with the signal.
• the output of average function 1204 is then subtracted from the sample and hold output producing a zero mean signal 1502.
• the zero mean signal 1502 is then squared by square function 1504 and filtered by filter 1506. This result (the output of filter 1506) represents the variance 1512 of the noise.
• a square root function 1508 is also applied, resulting in the RMS value 1510 of the noise.
• Fig. 16 illustrates an alternate processing method which may afford some implementation economies.
• the zero mean signal 1502 is provided to an absolute value function 1602 which is then filtered by filter 1604, resulting in an output 1606 that may be used in place of the RMS value 1510.
• the second mode to be considered occurs when the receiver is locked to a received signal.
• the pulse summation function is generating a generally ramp shaped time function signal due to the coherent detection of modulated data "ones" and "zeroes".
• the desired noise value measurement is the statistical standard deviation of the voltage associated with either the data “ones” or data “zeros".
• the absolute value of the voltage associated with either the data “ones” or data “zeros” can be used in place of standard deviation.
• the output of average function 1304 is subtracted from each sample resulting in a value 1326 that is then squared by square function 1306, and filtered by filterl308.
• the filtered result is then processed by square root function 1310, resulting in an RMS AC value 1325 representing the noise associated with the "ones" .
• a similar process is performed on the output of average function 1314 by the square function 1316, filter 1318, and square root function 1320, resulting in a value 1328 representing the noise associated with the data "zeroes".
• These two values 1325 and 1328 are combined resulting in a value 1330 representing the noise in the reception process. If the noise for the "ones" is equal to the noise for the "zeroes", then this method of adding the values results in a sum equivalent to twice the average of the noise value for the "ones".
• the noise value 1330 is combined with the signal strength value 1324 in a divide function 1332 to derive a signal-to-noise value 1334 result.
• computational economies may be achieved by using only the result of the data "ones” or data “zeros" processing for the standard deviation computation, or by using average absolute value in the place of standard deviation.
• Fig 14 illustrates an alternate solution to the square function 1306, filter 1308, and square root function 1310 sequence identified as 1336 in Fig. 13.
• the output of average function 1304 is subtracted from each sample resulting in a value 1326 that is provided to the absolute value function 1402 and the result is then filtered by filter 1308 to produce an alternative to the RMS value 1325.
• Other methods of achieving computational efficiency would be apparent to one of ordinary skill in the art.
• the terminology data “ones” and data “zeroes” refers to the logic states passed through the impulse radio receiver. In a typical system, however, there may be a Forward Error Correction ( hereinafter called FEC) function that follows the impulse receiver.
• FEC Forward Error Correction
• the data "ones" and “zeroes" in the impulse receiver would not be final user data, but instead would be symbol “ones” and “zeros” which would be input to the FEC function to produce final user data “ones” and "zeros".
• FIG. 17 An output combiner for the two noise measurement modes together with a mode logic method is shown with reference to Fig. 17.
• the output of the noise measurement 1510 from the algorithm of Fig. 15 which is valid for the unlocked case and the output of the noise measurement 1330 from the algorithm of Fig. 13, which is valid for the locked case, are provided to the two alternative inputs of a selector switch 1702.
• the switch 1702 is controlled by the output of a lock detector 1704, which determines the mode.
• the selected output is then supplied to the noise output 1106 of the signal evaluation block 1011 of Fig. 11.
• the lock detector 1704 comprises a comparator 1706 connected to the signal strength output 1324 of Fig. 13.
• a reference value 1708 supplied to the comparator 1706 is a value that is slightly higher than the ambient noise. For an impulse radio, and for digital radios in general, a 10 dB signal to noise ratio is generally required in order to achieve acceptable reception. Thus, it is feasible to place a threshold (that is, the reference value 1708) between the no-signal and the acceptable-signal level.
• the reference value 1708 may be fixed. In a more advanced radio, the reference value 1708 may be determined by placing the receiver in a state where lock is not possible due to, for instance, a frequency offset, and then setting the reference value 1708 such that the lock detector 1704 shows a stable unlocked state. In another embodiment, the reference value 1708 is set to a factor (e.g., two) times the unlocked noise value 1510.
• the output of lock detector 1704 is also shown switching (enabling) the outputs of the signal strength 1324 and signal to noise 1334 signals using switches 1712 and 1714, since these outputs are not meaningful until a significant signal is received and in lock.
• These outputs 1324, 1334 are then supplied to the outputs 1108, 11 10 of the signal evaluation function 1011 of Fig. 11. IV.2.C. Bit Error Rate (BER)
• the Bit Error Rate is measured directly from the received data stream 739.
• the result 1112 is provided to the power control algorithm 1014.
• BER can be measured by a number of methods depending on the configuration of the system. In an embodiment adaptable for a block oriented data transmittion system, BER is measured periodically, by sending a known bit pattern and determining the number of bits in error. For example, a known one-thousand bit message could be sent ten times a second, and the result examined for errors. The error rate could be directly calculated as the number of errors divided by the total bits sent.
• This block of known BER pattern data may be broken into sub-blocks and sent as part of the data contained in block or packet headers. Both of these methods require considerable overhead in the form of known data sent on the link in order to calculate the error rate.
• the error correction rate can be used as the raw BER measurement representative of signal quality.
• Suitable algorithms including Reed Soloman, Viterbi, and other convolutional codes, or generally any FEC method that yields an error correction rate can be used.
• parity or check sums are used as a measure of errors, even though they alone are insufficient to correct errors.
• the user data is used to measure the error rate and a very small overhead of one percent or less is required for the parity to detect normal error rates.
• one parity bit added to each block of 128 data bits could measure error rates to 10 " 2 . which would be sufficient to control to a BER of 10 3 .
• double bit errors within a block will go unnoticed, this is not of much consequence since the average of many blocks is the value used in the power control loop.
• the signal strength measurement 1324 could be fairly responsive, i.e. have very little averaging or filtering, in fact it may have no filtering and depend on the power control loop or algorithm 1014 to provide the necessary filtering.
• the signal to noise measurement 1334 also could be fairly responsive to power changes because the signal measurement is simply propagated through the signal to noise divide operation 1332.
• the noise measurement 1330 typically needs significant filtering 1308 to provide a stable base for the divide operation 1332. Otherwise, the SNR value 1334 will vary wildly due to fluctuations in the noise measurement 1330.
• Fig. 18 is a flowchart that describes a method of power control according to the present invention. Fig. 18 is described with reference to the example environment depicted Figs. 9 and 10.
• transceiver 902A transmits a signal SI.
• transceiver 902B receives signal SI .
• a power control update 1016 is calculated according to a performance measurement(s) of received signal SI.
• performance measurements are discussed below, such as received signal strength, BER, and SNR, can be used either alone or in combination.
• step 1808A and 1808B the output power of either transmitter 602A of transceiver 902A or transmitter 602B of transceiver 902B (or both) is controlled according to the power control update 1016.
• step 1808 A the power of transmitter 602A of transceiver 902A is controlled according to the power control update 1016, which is preferably calculated (in step 1806) at transceiver
• Step 1808A is described in additional detail in Fig. 19.
• transceiver 902B transmits a power control update, in step 1902.
• transceiver 902A receives the power control update from transceiver 902B.
• transceiver 902A adjusts its output power (Of transmitter 602 A) according to the received power control update 1016.
• the power control for a particular transceiver is therefore determined by the performance (measured by another transceiver receiving the signals) of signals it transmits.
• the output power of transmitter 602B of transceiver 902B is controlled according to the power control update 1016.
• the power control for a particular transceiver is therefore determined by the performance of signals it receives from another transceiver.
• This embodiment assumes that the propagation path between transceivers in communication is bilaterally symmetric, i.e., that signals transmitted between the pair of transceivers undergo the same path loss in both directions.
• the propagation path between transceivers 902A and 902B is bilateral symmetric if signal SI undergoes the same path loss as signal S2.
• the path loss of SI therefore provides an accurate estimate of the path loss of S2 to the extent that the propagation path approaches bilateral symmetry.
• the power control of transceiver 902B is determined by the performance of received signal SI (which is transmitted by transceiver 902A and received by transceiver 902B) in lieu of evaluating received signal S2 (which is transmitted by transceiver 902B and received by transceiver 902 A).
• Impulse radio provides a unique capability for implementing this kind of system.
• the multipath signals are delayed from the direct path signal.
• the first received pulse in a multipath group will be the direct path signal. If both transceivers in a transceiver system are configured to find and lock on the earliest signal in a multipath group, then the symmetry will be assured, assuming the direct path exists. If the direct path does not exist because of obstruction, then both transceivers will still likely lock on the same early multipath reflection - resulting in a bilateral symmetric propagation configuration.
• a power control update is calculated according to a performance measurement(s) of received signal SI .
• the signal strength of the received signal is used as a performance measurement.
• the power control update, dP is given by:
• K is a gain constant
• P S1 is the signal strength of received signal SI ;
• P ref is a signal strength reference
• dP is the power control update (which is preferable in the unit of Volts).
• the output level of transmitter 602A (of transceiver 902A) is therefore increased when P S) falls below P ref , and decreased when P s ⁇ rises above P ref .
• the magnitude of the update is linearly proportional to the difference between these two signals. Note that the power control update can be equivalently expressed as an absolute rather than a differential value. This can be achieved by accumulating the differential values dP and communicating the resulting output level P as follows:
• P n is the output level (e.g., voltage level or power level) to be transmitted during the next evaluation interval;
• P n _ is the output level transmitted during the last evaluation interval; and dP is the output level increment computed as a result of the signal evaluation during the last interval.
• the power control update could be quantized to two or more levels.
• a control loop diagram illustrating this embodiment will now be described with reference to Fig. 20.
• a signal 2002 (e.g., signal 2002 is transmitted by transmitter 602A of transceiver 902A) having a transmitted output level is disturbed by the propagation path according to a disturbance 2004.
• This disturbance 2004 may be modeled as either an additive process or a multiplicative process.
• the multiplicative process is generally more representative of the attenuation process for large disturbances 2004.
• transmitter 602A of transceiver 902 A if the embodiment including step 1808 A is implemented, or transmitter 602B of transceiver 902B if the embodiment including step 1808B is implemented
• a signal 2002 having a new output level (e.g., voltage level or power level).
• a new output level e.g., voltage level or power level.
• the receiver contains an automatic gain control (AGC)
• AGC automatic gain control
• the operation of this AGC must be taken into account in the measurement of signal strength. Indeed, some AGC control signals are suitable for use as a signal strength indicator.
• the embodiment of 1808B is implemented, the integrating step
• the gain K 1 should be low such that the gain correction is less than sufficient to fully level the power, preferably less than half of what would level the power. This will prevent instability in the system.
• Such low gain KI would likely be discarded as unworkable in conventional spread spectrum systems, but because of the potentially very high processing gain available in an impulse radio systems, and impulse radio system can tolerate gain control errors of much greater magnitude than conventional spread spectrum systems, making this method potentially viable for such impulse radio systems.
• the system functions including the reference 2010, the K[ scaling function 2012, and the integrator 2014 can be partitioned into either the transmitter or receiver at the convenience of the designer.
• the performance measurement might be compared against one or more threshold values. For example, if one threshold value is used the output power is increased if the measurement falls below the threshold and decreased if the measurement rise above the threshold.
• the performance measurement is compared against two threshold values, where output power is increased if the measurement falls below a low threshold, decreased if the measurement rises above a high threshold, or held steady if between the two thresholds. This alternative method is often referred to as being based on hysteresis. These two thresholding methods could also be used with the remaining performance measurements discussed below.
• transceiver 902A does not evaluate the signal.
• Transceiver 902B evaluates the signal strength of SI and computes a power control update command for transmitter 602B and for transmitter 602A.
• the power control update (dP) command for transmitter 602 A is sent to transceiver
• the SNR of the received signal is used as a performance measurement.
• the power control update, dP is given by:
• K is a gain constant
• SNR S is the signal-to-noise ratio of received signal SI ;
• SNR ref is a signal-to-noise ratio reference.
• the power of transmitter 602A (of transceiver 902A) is therefore increased when SNR S1 falls below SNR ref , and decreased when SNR S , rises above
• a control loop diagram illustrating the functionality of this embodiment will now be described with reference to Fig. 21.
• a signal 2002 e.g. signal 2002 is transmitted by transmitter 602A of transceiver 902A
• This disturbance 2004 may be modeled as either an additive process or a multiplicative process; however, the multiplicative process is generally more representative of the attenuation process for large disturbances 2004.
• the resulting signal 2006 is then combined with additive noise 2102 representing thermal and interference effects to yield a combined signal 2104 which is received by the receiver (receiver 702B of transceiver 902B), where signal strength 2008 and noise 2106 are measured. These values are combined 2108 to yield a signal to noise measurement 2110 (SNR SI ).
• the signal to noise measurement 2110 is then compared with a signal to noise reference value 2112 (SNR,. ef ).
• the result is then scaled by K, 2012 (K) to produce power control update 2013 (dP).
• Power control update (dP) is summed or integrated 2014 over time to produce a power control command signal 2016 to command the power control function 2018 (1126 in Fig.
• the BER of the received signal is used as a performance measurement.
• the power control update, dP is given by:
• K is a gain constant
• BER S1 is the bit error rate of received signal SI
• BER ref is a bit error rate reference.
• the sign is reversed in this case because the performance indicator, BER is reverse sensed, i.e. a high BER implies a weak signal.
• the power of transmitter 602A (of transceiver 902A) is therefore decreased when BER S] falls below BER ref , and increased when BER S1 rises above BER ref .
• the magnitude of the update is linearly proportional to the difference between these two signals.
• the power control update can be equivalently expressed as an absolute rather than a differential value. As described above, many alternative formulations are possible for calculating a power control update according to received signal BER.
• BER measurements span a large dynamic range, e.g., from 10 "6 to 10 " ', even where the received signal power may vary by only a few dB. BER measurements are therefore preferably compressed to avoid the wide variation in control loop responsiveness that would otherwise occur.
• One method of compressing the range is given by:
• log() is the logarithm function and the other variables are defined above.
• five orders of dynamic range are compressed into the range from -1 to -6, which makes the control loop stability manageable for typical systems.
• An alternative compression function can be generated by mapping BER into equivalent dB gain for a given system. This function can be based on theoretical white Gaussian noise, or can be based on measurements of environmental noise for a given system.
• BER As the measure of performance provides meaningful power control in digital systems.
• SNR is not as meaningful as BER, but may be determined more quickly.
• Signal strength is less meaningful still because it does not account for the effects of noise and interference, but may be determined with only a single sample.
• Those skilled in the art will recognize that one would use these performance measurements to trade accuracy for speed, and that the particular environment in which the transceivers will be used can help determine which measurement is the most appropriate. For example, received signal variations in a mobile application due to attenuation and multipath signals demand high update rates, whereas high noise environments tend to need more filtering to prevent erratic behavior.
• Combining BER, SNR, and/or signal strength can produce other useful performance measurements.
• BER and signal strength are combined to form a performance measurement, where the power control update, dP, is given by:
• K, and K 2 are gain constants
• BER S1 is the bit error rate of received signal S 1 ;
• BER ref is a bit error rate reference;
• P S1 is the signal strength of received signal SI.
• P ref a signal strength reference
• P ref a signal strength reference
• BER and SNR are combined to form a performance measurement.
• the power control update, dP is given by:
• K, and K 2 are gain constants
• BER SI is the bit error rate of received signal SI
• BER ref is a bit error rate reference
• SNR S] is the signal-to-noise ratio of received signal SI .
• the power control update, dP is given by:
• K, and K 2 are gain constants
• BER S ⁇ is the bit error rate of received signal SI
• BER ref is a bit error rate reference
• SNR S1 is the signal-to-noise ratio of received signal SI.
• SNR ref a signal-to-noise ratio reference
• a control loop simulation diagram illustrating the functionality of an embodiment based on BER and SNR will now be described with reference to Fig. 22.
• a signal 2002 e.g., signal 2002 is transmitted by transmitter 602A of transceiver 902 A
• a disturbance 2202. which may include both propagation and noise effects as in Fig. 21 yielding a combined signal 2104 which is received by the receiver (receiver 702B of transceiver 902B).
• This signal 2104 is evaluated for signal to noise ratio 2204 (combined functions of 2008, 2106 and 2108) and then compared with a reference 2206 to yield a result 2210.
• This result 2210 is then scaled by scaling function K, 2012 (K,) and summed or integrated over time by integrator 2014 to produce a power control command signal 2016 to command the power control function 2018 (1126 in Fig. 11) of the transmitter (transmitter 602A of transceiver 902A if the embodiment including step 1808 A is implemented, or transmitter 602B of transceiver 902B if the embodiment including step 1808B is implemented) to output a signal 2002 having a new power level.
• the embodiment including step 1808A is preferred, because the embodiment including step 1808B is susceptible to errors from non-symmetrical noise and interference as in the case where interfering transmitter 910 is closer to receiver 702B than to receiver 702A.
• the embodiment including step 1808B may be used in applications that do not need precise power control by using low gain factors (K, and K 2 ).
• Reference 2206 is based on BER measurement 2208 (BER S1 ) of signal 2104. More specifically, signal 2104 is evaluated for BER 2208 and then compared to desired BER reference 2209 (RER re/ ). The result is then scaled by
• K, and K 2 are gain constants;
• BER S1 is the bit error rate of received signal SI ;
• BER ref is a bit error rate reference;
• dSNR ref is an incremental change in SNRref ;
• SNR ref is a calculated reference used in the SNR loop
• SNR S1 is the signal-to-noise ratio of received signal SI.
• FIG.23 A control loop simulation diagram illustrating the addition of the log(BER) function will now be described with reference to Fig.23. It can be seen that this Figure is substantially similar to Fig. 22 except that the BER measurement 2208 is processed by a log function 2302 (log(BER sl ))and compared with a reference 2304 ( og(RER re )suitable for the log(B ⁇ R) value before being scaled by scaling function K 2 2212 (K 2 ) and integrated or filtered by integrator 2214 and used as the reference 2206 (SNR re f) for the SNR control loop .
• a log function 2302 log(BER sl )
• og(RER re ) suitable for the log(B ⁇ R) value before being scaled by scaling function K 2 2212 (K 2 ) and integrated or filtered by integrator 2214 and used as the reference 2206 (SNR re f) for the SNR control loop .
• K 2 2212 K 2
• power control for a particular transceiver can be determined based on the performance (i.e., signal strength, SNR and/or BER) of signals transmitted by the particular transceiver and received by another transceiver (e.g., transceiver 902B), as specified in step 1808A of Fig. 18.
• step 1808 A the power of transmitter 602A of transceiver 902A is controlled according to a power control update, which is preferably calculated at transceiver 902B and transmitted from transceiver 902B to transceiver 902A.
• each of the above discussed embodiments for performing power control for a particular transceiver can be determined based on the performance (i.e., signal strength, SNR and/or BER), of signals it receives, as in step 1808B of Fig. 18. More specifically, according to this embodiment, the power control for a particular transceiver (e.g., transceiver 902A) is determined by the performance of signals it receives from another transceiver (e.g., signals transmitted from transceiver 902B and received by transceiver 902 A). This power control embodiment assumes that the propagation path between transceivers in communication is bilaterally symmetric.
• the performance i.e., signal strength, SNR and/or BER
• an interfering transmitter e.g., transmitter 908
• interfering transmitter 908 when present, will disturb the system asymmetrically when it is nearer to one transceiver.
• interfering transmitter 908 is nearer to transceiver 902B.
• the noise level at transceiver 902B will increase more than the noise level at transceiver 902A.
• the response of the power control system can vary depending on the performance measurement utilized. If the power control system is using signal strength, the control system would be unaffected by the interference, but if the system is using signal to noise ratio, the nearby transceiver 902B would increase power to overcome the performance degradation. In this case, it is an unnecessary increase in power. This increase in power would be seen as a propagation improvement at transceiver 902A, which would decrease power, resulting in an even lower SNR at 902B, which would increase power further. Clearly this is not workable.
• this can be overcome by communicating to transceiver 902B the power (e.g., relative power or absolute power) transmitted by transceiver 902A.
• This allows transceiver 902B to separate power changes due to power control from changes due to propagation.
• This communication can be accomplished according to conventional techniques, such as transmitting a digital message in a link control header, or transmitting a periodic power reference.
• transceiver 902B may adjust its power based only on propagation changes and not on power control adjustments made by transceiver 902A.
• Multi-path environments can also disturb system symmetry.
• a transceiver 902 can lock onto various multi-path signals as the transceivers in communication move in relation to one another.
• FIG. 24 A more general block diagram of a transceiver power control system including power control of both transmitters (i.e., transmitter 602A of transceiver 902A and transmitter 602B of transceiver 902B) from signal evaluations from both transceivers (i.e., transceivers 902A and 902B) is shown in Fig. 24.
• auto-power control refers to power control of a first transceiver's
• step 1808B refers to the control of a first transceiver's (e.g., transceiver 902A) output according to the evaluation of the first transceiver's transmitted signal as received at a second transceiver (e.g., transceiver 902B).
• cross power control relates to step 1808 A discussed above.
• transmitter 602A transmits a signal 2402 to receiver 702B of transceiver 902B.
• This signal 2402 is evaluated by signal evaluation function 101 IB resulting in performance measurement(s) 1012B (e.g., signal strength, SNR and/or BER) which are delivered to the power control algorithm 1014B.
• the power control algorithm 1014B also receives power control messages 2404 from transmitter 602 A via the receiver data demultiplexer 1020B, which separates user data and power control messages 2404.
• These power control update messages 2404 can comprise data related to the power level of transmitter 602A and/or signal evaluations (e.g., signal strength, SNR, and/or BER) of signals 1008 received by receiver 702A (i.e., signals transmitted by transceiver 902B and received by transceiver 902A).
• the power control algorithm 1014B then computes a new power level
• Power control algorithm 1014B can also deliver signal evaluations 2408, which are based on measurements determined by signal evaluation function 101 IB. to the TX data multiplexer 1018B .
• signal evaluation function 1011 B can deliver this information 2408 directly to TX data multiplexer 1018B .
• This signal evaluation data 2408 is then added to the input data stream and transmitted at the commanded power level 2406B.
• Fig. 25 illustrates an embodiment of the power control algorithm 1014B (of transceiver 902B) employing auto-control with power level messaging.
• the received signal (transmitted by transmitter 702A and received by receiver 602B) is evaluated for signal strength 1106B by signal evaluation function 1011 B .
• receive data demultiplexer 1020B (See Fig. 24) separates user data and power control messages 2404 and delivers the power control messages 2404 to subtract function 2502B.
• the power control message value 2404 (representing the output level of transmitter 602A) is then subtracted by subtractor 2502 from the signal strength measurement 1106 (which is based on the strength of a signal transmitted by transceiver 902A).
• the result 2406 is used to deviate (e.g., decrease or increase) the transmitter output from a nominal output level. Additionally, a message value that represents the transmitted output level is generated and sent to the other transceiver 902 A.
• Fig. 26 illustrates an embodiment where auto and cross control are implemented in combination. Referring to Fig.
• the received signal is evaluated by signal evaluation function 101 IB for signal strength 1106B and SNR 1110B.
• the output level signal 2404 (representing the output level of transmitter 602 A) is subtracted from the signal strength 1106B resulting in an auto control signal 2406.
• This auto control signal 2406 is combined with a signal strength 1106A or SNR measurement 1110A determined by the signal evaluation function 1011 A of the other transceiver 902 A and further filtered by combiner/filer 2602 to produce an output level value 2604 used to control the output level of transmitter 602B.
• This output level value 2604 is combined with the signal strength 1106B and SNR 1110B measurements by multiplexer 2606, and then further combined with the transmitted data stream by transmit data multiplexer 1018B.
• This system takes full advantage of both the auto and cross power control methods, with the auto power control generally offering speed of response, and the cross power control offering precision together with tolerance of link imbalance and asymmetry.
• the power control update is calculated at the transceiver receiving the signals upon which the update is based.
• the data required to calculate the power control update may be transmitted to another transceiver and calculated there.
• step 1808 A and 1808B the output power of either transceiver 902A or 902B (or both) is controlled according to the power control update calculated in step 1806.
• step 1808A the power of transmitter 602A of transceiver 902A is controlled according to the power control update.
• Fig. 19, briefly discussed above, is a flowchart that depicts step 1808 A in greater detail according to a preferred embodiment.
• transceiver 902B transmits the power control update calculated in step 1806 (assuming that, according to a preferred embodiment, the power control update is calculated at transceiver 902B).
• transceiver 902A receives the power control update.
• step 1906 transceiver 902A adjusts its output level (e.g., voltage level or power level) according to the received power control update, as described in detail below.
• step 1808B the power of transmitter 602B of transceiver 902B is controlled according to the power control update.
• the power level of the signal S 1 (sent by transceiver 902 A and received by transceiver 902B) is used to control the output level of transmitter 602B.
• transceiver 902B preferably calculates the power control update and adjusts the power of its transmitter 602B accordingly.
• power control refers to the control of the output power of a transmitter, this is usually implemented as a voltage control proportional to the output signal voltage.
• power control of a transmitter 902 can be accomplished by controlling any parameter that affects power.
• the pulse peak power e.g., the height of pulses
• the timing parameters For example, Fig. 27 shows two signals 2702 and 2704 having different pulse peak powers but the same timing parameters. Note that signal 2702 has a greater pulse height and thus corresponds to a greater transmitter power than signal 2704.
• the number of pulses per bit is controlled, thereby controlling the integration gain while keeping pulse peak power constant.
• Integration gain relates to (e.g., is proportional to) the number of pulses summed or integrated in the receiver for each data bit.
• the transmitted power is directly proportional to the number of pulses per bit transmitted.
• the number of pulses may be found by first, summing the differential commands, and then computing the number of pulses based on this summation, as in the following:
• P n is the present commanded output level (e.g., voltage level or power level); P-..] is the output level transmitted during the just completed evaluation interval; dP is the output level increment commanded (also referred to as the power update command 1016) as a result of the just completed evaluation interval; N tra , n is the number of pulses per data bit (also referred to as the number of pulses in a pulse train) to be transmitted during the present evaluation interval; and
• K p is a constant relating power to number of pulses per bit.
• N tra ⁇ n cannot be greater than full power, nor can N tram be less than one. In some cases, N tram must be an even integer or some other quantized level.
• type A pulses 2802 shall be defined as pulses delayed from nominal by V2 modulation time and type B pulses 2804 shall be defined as pulses advanced from nominal by V2 modulation time.
• the difference between type A pulses 2802 and type B pulses 2804 is one full modulation time.
• each period 2806 comprises eight type A pulses 2802 followed by eight type B pulses 2804 when a data "one" is transmitted.
• Power can be reduced by adjusting the subcarrier cycle length by, for example, changing to eight periods 2808 of 14 pulses each (i.e., N pulses . per . period is reduced from 16 pulses/period to 14 pulses/period), where each period 2808 comprises seven type A pulses 2802 followed by seven type B pulses 2804 and two empty pulses 2810.
• each period 2808 can comprise seven type A pulses
• the power may be reduced by reducing the number of subcarrier cycles.
• the example system could transmit seven (instead of eight) periods of 16 pulses (i.e. N period is reduced from 8 periods to 7 periods), where each period comprises eight type A pulses followed by eight type B pulses when a data "one" is transmitted. This would result in a total of 1 12 pulses per data bit, as opposed to 128 pulses per data bit (i.e., N train is reduced from 128 pulses/bit to 112 pulses/bit).
• a subcarrier cycle can be reduced from eight periods 2806 of 16 pulses to seven periods 2806 of 16 pulses.
• Patterns may be generated that balance the pulse types over the data bit, wherein one or more subcarrier periods may be unbalanced. Some systems may even tolerate an unbalance of pulse types over a data bit, but this will usually come with some performance degradation. Other patterns can be easily implemented by one of ordinary skill in the art following the principles outlined in these examples.
• the receiver integration gain should ideally track the number of pulses transmitted. If these values are not coordinated, loss of performance may result. For example, if the receiver is receiving 128 pulses for each data bit and the transmitter is only transmitting the first 64 of these pulses, the receiver will be adding noise without signal for the second half of the integration time. This will result in a loss of receiver performance and will result in more power transmitted than necessary. This can be prevented by coordinating the number of pulses between the transmitter and receiver. In one embodiment, this information is placed in the headers or other control signals transmitted so that the receiver can determine exactly how many pulses are being sent.
• the receiver employs multiple parallel bit summation evaluations, each for a different possible integration gain pulse configuration.
• the SNR 1110 is evaluated for each summation evaluation path, and the path with the best SNR is selected for data reception. In this way, the receiver can adaptively detect which pulse pattern is being transmitted and adjust accordingly.
• Power control can be improved by expanding the gain control sensitivity at high levels relative to low levels.
• an unexpanded gain control function would be one where the voltage or power output would be simply proportional to the voltage or power control input signal:
• V ou , K cll V cll
• V out is the pulse voltage output
• K ctl is a gain constant (within power control block 1014, not to be confused with K,);
• V ct is the control voltage input (power control command signal).
• V m ⁇ l K a , exp(V
• This function can be implemented with a exponential gain control device, or a separate exponential function device together with a linear gain control device.
• An embodiment using a exponential gain control device is described in relation to Fig.20. In this embodiment, operation is much the same as previously described for the linear power control case except that now the power control function 2018 controls the power output in a manner such that the power output, expressed in decibels (dB), is substantially proportional to the power control input voltage 2016 (V /I ) (also referred to as, the power control command signal).
• V /I power control input voltage
• a signal 2002 (V m ⁇ l ) having a transmitted power level is disturbed by the propagation path according to a disturbance 2202.
• the resulting received signal 2104 is evaluated for signal to noise ratio 2204 and compared with the desired signal to noise reference 2112.
• the result is then scaled by K x 2012 and summed or integrated over time by integrator 2014 to produce an output 2902.
• 2902 drives an exponential function 2904 to yield a power control command signal 2906 to command the power control function 2018 (1126 in Fig. 11) of a transmitter to output a signal 2002 (V m ⁇ ! ) having a new power level.
• Np 2 KP p
• Np is the number of pulses per data bit to be transmitted
• P is the power control command
• Kp is a scaling constant
• Np is the only value in the above equation that is rounded to an integer.
• greater implementation simplicity may be achieved by rounding the product KpP to an integer value.
• a command for lower power results in half of the present number of pulses per data bit being transmitted.
• a command for more power results in twice the present number of pulses per data bit being transmitted.
• the average interference is the same whether a high power/high data rate message is transmitted for a short time or whether a low power/low data rate message is transmitted over a longer time.
• the user of a computer system would usually prefer the message to be transmitted in a short time.
• the power can be minimized during low data rate intervals to minimize interference.
• it is also possible to maintain maximum power and maximum data rate, but to turn off the transmitter for intervals when no data is available.
• the transmission wave may be electromagnetic or acoustic.

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