WO1999063722A1 - Logique de demodulation - Google Patents

Logique de demodulation Download PDF

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Publication number
WO1999063722A1
WO1999063722A1 PCT/FI1999/000457 FI9900457W WO9963722A1 WO 1999063722 A1 WO1999063722 A1 WO 1999063722A1 FI 9900457 W FI9900457 W FI 9900457W WO 9963722 A1 WO9963722 A1 WO 9963722A1
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WO
WIPO (PCT)
Prior art keywords
difference
vectors
signal sample
signal
data
Prior art date
Application number
PCT/FI1999/000457
Other languages
English (en)
Inventor
Hang Zhang
Marko HEINILÄ
Original Assignee
Nokia Networks Oy
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Nokia Networks Oy filed Critical Nokia Networks Oy
Priority to AU45171/99A priority Critical patent/AU4517199A/en
Publication of WO1999063722A1 publication Critical patent/WO1999063722A1/fr

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
    • H04L27/22—Demodulator circuits; Receiver circuits
    • H04L27/233—Demodulator circuits; Receiver circuits using non-coherent demodulation
    • H04L27/2332—Demodulator circuits; Receiver circuits using non-coherent demodulation using a non-coherent carrier
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/06—Dc level restoring means; Bias distortion correction ; Decision circuits providing symbol by symbol detection
    • H04L25/061—Dc level restoring means; Bias distortion correction ; Decision circuits providing symbol by symbol detection providing hard decisions only; arrangements for tracking or suppressing unwanted low frequency components, e.g. removal of dc offset

Definitions

  • the invention is directed to a demodulation method for tamed frequency modulated signals.
  • the invention is further directed to a demodulation method according to the preamble of the independent claim directed to a demodulation method.
  • Continuous phase modulation is a class of modulation techniques, which are efficient in power and bandwith at the same time.
  • the RF signal envelope is constant and phase varies in a continuous manner.
  • the constant envelope allows the use of nonlinear amplifiers, simplifying receiver and transmitter design.
  • Tamed frequency modulation is one continuous phase modulation scheme.
  • One of the main benefits of TFM is the very narrow bandwith needed in comparison with most of the other modulation schemes.
  • CPM signals can be described by
  • n 2 ⁇ a l h l q(t -iT b ), nT b ⁇ t ⁇ (n+l)T b (2)
  • h t is the modulation index, which may vary from interval to interval
  • q(t) is the phase response function
  • g(t) is the frequency response
  • E b is the bit energy
  • T b is the bit interval
  • f 0 is the carrier frequency
  • ⁇ 0 is an arbitrary initial phase.
  • Characteristic for TFM modulation is, that the phase shift of the modulated carrier over one bit interval is determined not only by the current bit but by three consecutive input binary signals in accordance with the encoding rule:
  • phase changes of ⁇ /2 are obtained, if three successive bits have the same polarity, and the phase remains constant for three bits of alternating polarity.
  • Phase changes of ⁇ /4 are connected with the bit configurations ++ -, + — , — + and — +.
  • TFM The signal space diagram of TFM is shown in Figure 1.
  • TFM signal provides a narrower power spectrum than for example a MSK signal, since the changes of phase are smoother in TFM.
  • TMF modulation is described further in the article entitled "Tamed
  • Direct conversion receivers are receivers, which do not use intermediate frequencies for filtering and detection of received signals.
  • the received RF signal is mixed with a local oscillator signal, whose frequency corresponds to the carrier frequency of the RF signal.
  • Direct conversion receivers have many advantages. For example, the bandwith filtering can be performed at low i.e. audio frequencies, allowing realisation of very narrow bandwiths with sharp edges. Also, intermediate frequency filters are not needed.
  • direct conversion receivers have not been used for receiving TFM signals due to an inherent problem of direct conversion receiver structures, namely the existence of a DC offset at the mixer output as a result of imperfections of the mixer structure.
  • the DC offset results from leaking of the local oscillator signal to the RF port of the mixer and subsequent mixing of the leaked signal with the local oscillator signal itself.
  • the random fluctuations in phase of the leakage signal result in a relatively slowly and randomly changing DC offset signal.
  • Low frequency phase noise resulting from transmitter phase noise or local oscillator phase noise can be presented as an example.
  • the DC offset problem can be thought of as a very low frequency phase noise.
  • Figure 2 illustrates the problem created by the DC offset in the detection of TFM modulated data. Without any DC offset, the received signal has the constellation shown with white circles.
  • the vectors Si and s 2 drawn with a thick dashed line, represent signals detected at two consecutive sampling times. Without any DC offset, the detection of vectors Sj and s 2 is straightforward. However, DC offset changes the situation considerably.
  • DC offset has the effect of moving the signal constellation in the IQ-diagram, as depicted by the black circles.
  • the vectors s ⁇ and s' 2 show the corresponding sampled signal vectors in the presence of DC offset.
  • any detector optimized for detecting the vectors Si and s 2 has difficulties in recognizing vectors s ⁇ and s' 2 . Therefore, the presence of DC offset results easily in a high detection error rate.
  • phase difference ⁇ n between A n and ⁇ personally_ j is defined as
  • the phase does not change with certain information bit patterns. Therefore, the quantity ⁇ n can also be zero for some bit patterns, which makes it impossible to make decisions about the bits based on the value of the quantity ⁇ n .
  • An object of the invention is to realize a demodulation method for demodulating TFM signals.
  • a further object of the invention is to realize a demodulation method, which is able to demodulate a TFM signal combined with DC offset or other types of low frequency noise. It is also an object of the invention to realize a demodulator structure capable of demodulating a TFM signal under DC offset.
  • the objects are reached by using the length of a differential vector between two received signal sample vectors as the decision variable for detecting the transmitted data symbols.
  • the demodulation method according to the invention is characterized by that, which is specified in the characterizing part of the independent method claim directed to a demodulation method.
  • the data transmission method according to the invention is characterized by that, which is specified in the characterizing part of the independent method claim directed to a data transmission method.
  • the demodulator system according to the invention is characterized by that, which is specified in the characterizing part of the independent claim directed to a demodulator system.
  • the mobile communication means according to the invention is characterized by that, which is specified in the characterizing part of the independent claim directed to a mobile communication means.
  • the base station according to the invention is characterized by that, which is specified in the characterizing part of the independent claim directed to a base station.
  • the radio link system according to the invention is characterized by that, which is specified in the characterizing part of the independent claim directed to a radio link system.
  • ⁇ 1 * is the differential vector between two received vectors over one bit interval
  • ⁇ 2 W is the differential vector between received vectors over two bit intervals
  • ⁇ n is the phase difference between two differential vectors ⁇ 1 ,, and ⁇ i .
  • the phase change over one bit interval is determined by three consecutive transmitted bits. Further, the quantities ⁇ 2 personally and ⁇ scenicwill be determined by four consecutive information bits.
  • Figure 3 illustrates the possible state changes resulting from two consecutive symbols. Since according to equation (3) the phase change of one symbol can be 0, ⁇ /4, or ⁇ /2, the concatenation of two symbols result in a phase change of either one of ⁇ 0, ⁇ /4, ⁇ /2, 3 ⁇ /4, ⁇ or one of ⁇ 0, - ⁇ /4, - ⁇ /2, -3 ⁇ /4, - ⁇ . Therefore, the difference of two consecutive signal vectors may be one of the vectors shown in Figure 3.
  • Figure 3 illustrates the case of ⁇ 0, ⁇ /4, ⁇ /2, 3 ⁇ /4, ⁇ .
  • the length of the difference vector may be one of z, s, l lt l 2 , or l 3 .
  • Length z refers to a zero differential vector
  • length 5 to a differential vector from one to a neighbouring constellation point
  • length lj refers to the length of a differential vector bypassing one constellation point on the constellation point circle
  • length l 2 refers to the length of a differential vector bypassing two constellation points on the constellation point circle
  • length U refers to the length of a differential vector bypassing three constellation points on the constellation point circle.
  • bit patterns which are complements of each other have the same differential vector lengths, i.e. have equal values of and ⁇ 2 .
  • the differential vectors corresponding to bit pattern 0010 and have the same values as the corresponding differential vectors corresponding to bit pattern 1101, as can be seen from the previous table. Therefore, based on the differential vector lengths alone, the complement bit patterns cannot be distinguished from each other at the receiver. This problem can be alleviated using a differential encoding. In differential encoding, two consecutive bits having the same value correspond to a "0" bit, and two consecutive bits having differing values correspond to a "1" bit.
  • the differentially encoded bit patterns 0010 and 1101 correspond to the same decoded bit pattern, 011.
  • any bit patterns which are complements of each other produce the same differentially decoded bit patterns.
  • the bit patterns 0010 and 1101 corresponding to the decoded bit pattern 011 have equal values of the differential vector lengths. Consequently, when the transmitted data is encoded with differential encoding, the length of a differential vector provides enough data for detecting the received symbols. Therefore, in various advantageous embodiments of the invention, the quantities and ⁇ 2 can be used as a decision variable to detect the received symbols. Also, any quantity substantially dependent on ⁇ l or or both can be used as a decision variable.
  • ⁇ 2 is used as the decision variable in combination with the 4-state trellis of Figure 4.
  • the length of the corresponding differential vector over two bit intervals, i.e. the value of ⁇ 2 resort is denoted on each path of the trellis.
  • the value of the weight of the correct path should be close to zero, and ideally would be zero. Therefore, the correct path can be found in the Viterbi algorithm by finding the path with the smallest accumulated weight.
  • these weights can be formulated in many other ways as well. For example, one can formulate the weights in such a way, that the weight of a particular path has a maximum, when the path is the correct path. In that case, a maximum of the cumulative sum of the path weights would be sought in the Viterbi algorithm.
  • the exponent k may be adjusted for obtaining an optimum performance.
  • the value of k may be for example 1, in which case the weights are the absolute value of the difference between the length of the differential vector between received vectors over two bit intervals and the ideal length of the vector corresponding to the path in question.
  • the value of k may further be for example 2, in which case the weights are the square of the absolute value of the difference between the length of the differential vector between received vectors over two bit intervals and the ideal length of the vector corresponding to the path in question.
  • Other values of k are also possible, and the optimum value can be found out, for example, by simulation or experiment.
  • the invention is not limited to the use of as the decision variable, and the use of a 4-state trellis.
  • a 4-state trellis is used as the decision variable with a 2-state trellis.
  • Figure 1 illustrates the symbol constellation of tamed frequency modulation
  • FIG. 3 illustrates the difference vectors calculated over two bit intervals
  • Figure 4 illustrates the 4-state trellis used in an advantageous embodiment of the invention
  • FIG. 5 illustrates a receiver structure according to an advantageous embodiment of the invention
  • FIG. 6 illustrates a block diagram of an advantageous embodiment of the invention
  • Figure 7 illustrates a block diagram of a further advantageous embodiment of the invention.
  • FIG. 5 illustrates a block diagram of a receiver structure according to an advantageous embodiment of the invention.
  • a receiver structure can be used, for example, in mobile communication means and in base stations of cellular mobile communication networks.
  • the receiver structure comprises a sampling part 200 and a demodulator part 300.
  • the sampling part receives the RF signal, and converts the received signal to baseband I and Q signals with mixers 210a, 210b.
  • a local oscillator 205 provides the signal to be mixed with the received RF signal, and a phase shifter 206 produces a 90° phase shift to the local oscillator signal taken to the mixer 210b.
  • the output signals of the mixers 210a and 210b are taken to matched filters 215a, 215b, whose filtering properties are optimized for filtering TFM signals.
  • the output signals from the filters 215a, 215b is taken to switch elements 225, which produce samples of the quadrature I and Q signals. Each corresponding pair of I and Q samples define a signal sample vector.
  • the switch elements 225 are controlled by a sampling oscillator 220.
  • the sampling part 200 is an example of a structure, which can be used in a receiver structure according to the invention. As a man skilled in the art knows, many other structures can be used to produce a sampled, downconverted signal. The invention is not limited to using the sampling part structure 200 shown in Figure 5.
  • the switch elements 225 performing the sampling comprise typically A D converters or other types of sampling means.
  • Both I and Q signal paths comprise two adding units 260a, 260b, 260c, 260d and two delay units 250a, 250b, 250c, 250d.
  • the first pair of adding and delay units 250a,260a; 250c,260c produce the difference vector between consecutive bit intervals, after which the second pair of adding and delay units 250b,260b; 250d,260d add two consecutive difference vectors for producing the difference vector ⁇ 2 administrat over two bit intervals.
  • the structure of the adding and delay units shown in figure 5 realizes a combination of equations (4) and (7):
  • the difference vector A n over two bit intervals can as well be calculated using one adding block and one delay block in each of the I and Q signal paths, the delay block producing a delay of 2 T b , and the delayed sample stream being subtracted from the not delayed sample stream in the adding block according to equation (11).
  • the differences over two bit intervals of the I and Q signals are taken to a vector length determining block 270, which calculates the value of ⁇ 2 resort .
  • the resulting sequence of values of ⁇ 2 are taken to a Viterbi detector block 280, which determines the actual transmitted data bits, and produces the transmitted data bits at the output OUT of the structure.
  • the Viterbi detector block may advantageously use the 4-state trellis of Figure 4 and path weights given by equation (9).
  • Fig. 6 shows a block diagram of a digital mobile communication means according to an advantageous embodiment of the invention.
  • the mobile communication means comprises a microphone 301, keyboard 307, display 306, earpiece 314, antenna duplexer or switch 308, antenna 309 and a control unit 305, which all are typical components of conventional mobile communication means.
  • the mobile communication means contains typical transmission and receiver blocks 304, 311.
  • Transmission block 304 comprises functionality necessary for speech and channel coding, encryption, and modulation, and the necessary RF circuitry for amplification of the signal for transmission.
  • Receiver block 311 comprises the necessary amplifier circuits and functionality necessary for demodulating and decryption of the signal, and removing channel and speech coding.
  • the signal produced by the microphone 301 is amplified in the amplifier stage 302 and converted to digital form in the A/D converter 303, whereafter the the signal is taken to the transmitter block 304.
  • the transmitter block encodes the digital signal and produces the modulated and amplified RF-signal, whereafter the RF signal is taken to the antenna 309 via the duplexer or switch 308.
  • the receiver block 311 demodulates the received signal and removes the encryption and channel coding.
  • the resulting speech signal is converted to analog form in the D/A converter 312, the output signal of which is amplified in the amplifier stage 313, whereafter the amplified signal is taken to the earpiece 314.
  • the control unit 305 controls the functions of the mobile communication means, reads the commands given by the user via the keypad 307 and displays messages to the user via the display 307.
  • the mobile communication means comprises a demodulating part 300 performing demodulation according to the invention.
  • the demodulating part 300 advantageously has the structure shown in Figure 5. However, other structures for demodulating a TFM signal according to the invention can also be used.
  • Figure 7 shows an example of an embodiment of the invention.
  • a demodulator 300 according to the invention is used in at least some base stations 360 of a mobile communication network for demodulating TFM signals received from the mobile communication means 350.
  • Figure 7 further illustrates a base station controller 370 controlling the base stations 360 and two radio link units 371 for connecting the base station controller 370 to the rest of the mobile communication network 380.
  • Figure 7 also illustrates a further advantageous embodiment of the invention, namely the use of demodulators 300 according to the invention in radio links.
  • the demodulation method according to the invention is very suitable to be used in continuous high speed communications, where a continuous demodulation is needed. High speed radio links are one example of such an advantageous application of the invention.
  • the method according to the invention can advantageously be used to alleviate the effects of DC offset. Further, the method can also be used for demodulating TFM signals in presence of other types of low frequency noises such as low frequency phase noise or low frequency thermal noise.
  • the invention can advantageously be applied in direct downconverting receivers.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Power Engineering (AREA)
  • Digital Transmission Methods That Use Modulated Carrier Waves (AREA)

Abstract

Ce brevet se rapporte à une logique de démodulation de signaux en modulation de fréquence asservie ou 'TFM' (tamed frequency modulation). En l'occurrence, l'invention constitue un procédé permettant de démoduler des signaux TFM sous décalage du niveau continu. Le procédé permet également de détecter des symboles émis malgré le décalage du niveau continu ou la présence d'autres types de bruits basse fréquence dans le signal en bande de base. Le procédé consiste à prendre un signal reçu et à former des vecteurs d'échantillonnage du signal, à calculer des vecteurs différentiels sur la base des vecteurs d'échantillonnage du signal considéré, et à détecter les symboles de données reçus sur la base des longueurs des vecteurs différentiels considérés.
PCT/FI1999/000457 1998-05-29 1999-05-27 Logique de demodulation WO1999063722A1 (fr)

Priority Applications (1)

Application Number Priority Date Filing Date Title
AU45171/99A AU4517199A (en) 1998-05-29 1999-05-27 Demodulation method

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
FI981207 1998-05-29
FI981207A FI105751B (fi) 1998-05-29 1998-05-29 Demodulointimenetelmä

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WO1999063722A1 true WO1999063722A1 (fr) 1999-12-09

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
DE10318643B4 (de) * 2002-05-01 2010-09-02 Futaba Corp., Mobara-shi Mehrwertiges FSK Frequenzmodulationssystem

Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO1994028662A1 (fr) * 1993-06-02 1994-12-08 Nokia Telecommunications Oy Procede de demodulation d'un signal module numeriquement et demodulateur
US5406218A (en) * 1993-02-19 1995-04-11 Hitachi, Ltd. Phase demodulator receiving inputs from phase detector and binary phase detector
US5574399A (en) * 1994-11-10 1996-11-12 Hideto Oura Coherent PSK detector not requiring carrier recovery
GB2306085A (en) * 1995-10-02 1997-04-23 Secr Defence Demodulation in digital communication systems
WO1997016001A2 (fr) * 1995-10-25 1997-05-01 Siemens Aktiengesellschaft Procede de modulation a phase continue
US5684832A (en) * 1993-06-04 1997-11-04 Ntt Mobile Communications Network Maximum likelihood differential detecting method and differential detector thereof

Patent Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5406218A (en) * 1993-02-19 1995-04-11 Hitachi, Ltd. Phase demodulator receiving inputs from phase detector and binary phase detector
WO1994028662A1 (fr) * 1993-06-02 1994-12-08 Nokia Telecommunications Oy Procede de demodulation d'un signal module numeriquement et demodulateur
US5684832A (en) * 1993-06-04 1997-11-04 Ntt Mobile Communications Network Maximum likelihood differential detecting method and differential detector thereof
US5574399A (en) * 1994-11-10 1996-11-12 Hideto Oura Coherent PSK detector not requiring carrier recovery
GB2306085A (en) * 1995-10-02 1997-04-23 Secr Defence Demodulation in digital communication systems
WO1997016001A2 (fr) * 1995-10-25 1997-05-01 Siemens Aktiengesellschaft Procede de modulation a phase continue

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
BELLINI SANDRO ET AL: "Noncoherent Detection of Tamed Frequency Modulation", IEEE TRANSACTIONS ON COMMUNICATIONS, vol. COM-32, no. 3, March 1984 (1984-03-01), USA, pages 218 - 224, XP000758560 *

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
DE10318643B4 (de) * 2002-05-01 2010-09-02 Futaba Corp., Mobara-shi Mehrwertiges FSK Frequenzmodulationssystem

Also Published As

Publication number Publication date
AU4517199A (en) 1999-12-20
FI105751B (fi) 2000-09-29
FI981207A (fi) 1999-11-30
FI981207A0 (fi) 1998-05-29

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