KR100659250B1 - Control apparatus and module of permanent magnet synchronous motor - Google Patents

Abstract

The present invention provides a permanent magnet synchronous motor that can be commonly applied to a weakened field region, to realize a high-precision, high-response motor torque, and to provide a low-cost current detection system or a system in which a magnetic pole position detector is omitted. It is an object of the present invention to provide a weakened field vector control device.

In the present invention, for this purpose, the control gain of the calculation unit that prepares the d-axis current command is automatically corrected by the frequency command value of the motor, and the d-axis current command value is obtained by calculating in advance the d-axis current command generated at no load. Is added to the output of the operation that creates.

Description

CONTROL APPARATUS AND MODULE OF PERMANENT MAGNET SYNCHRONOUS MOTOR}

1 is a block diagram of a weakened field vector control device of a permanent magnet synchronous motor according to an embodiment of the present invention;

2 is an explanatory diagram of a weakened field command calculation unit 8 in the control device of FIG. 1;

3 is an example of the voltage saturation characteristic diagram when there is no weakened field command calculation unit 8;

4 is an example of the voltage saturation characteristic diagram when the weakened field command calculation unit 8 is inserted;

5 is a configuration diagram of a weakened field vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention;

FIG. 6 is an example of explanatory drawing of the weakened field command calculating part 8a in the control apparatus of FIG.

7 is a configuration diagram of a weakened field vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention;

FIG. 8 is an example of explanatory drawing of the weakened field command calculating part 8b in the control apparatus of FIG.

9 is a configuration diagram of a weakened field vector control device for a permanent magnet synchronous motor according to another embodiment of the present invention;

FIG. 10 is an example of explanatory drawing of the weakened field command calculating part 8c in the control apparatus of FIG.

11 is a block diagram of a weakened field vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention;

12 is a configuration diagram of a weakened field vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention;

13 is a configuration diagram when an embodiment of the present invention is applied to a module.

※ Explanation of symbols for main parts of drawing

1: Permanent magnet synchronous motor 2: Power converter

3: current detector 4: magnetic pole position detector

5: frequency calculating unit 6: phase calculating unit

7, 13: coordinate conversion unit

8, 8a, 8b, 8c: weakened field command calculation unit

9: d-axis current command calculator 10: q-axis current command calculator

11: voltage vector calculator 12: output voltage calculator

14 current estimation unit 15 phase error calculation unit

21: DC power

IDC: Input DC bus current detection value Id * : First d-axis current command value

Id ** : 2nd d-axis current command value Iq * : 1st q-axis current command value

Iq **: 2nd q-axis current command value

V ₁ * _ref : Output voltage command value in weakened field area

V ₁ *: Output voltage value θc *: Rotational phase command

ω ₁ *: Frequency command

The present invention relates to a vector control method of a weakened field region of a permanent magnet synchronous motor.

As a conventional technique of the vector control method of the weakened field region, a d-axis current command value described in Japanese Patent Application Laid-Open No. 8-182398 is tabulated and the current control of the d-axis and q-axis is used as a proportional calculation method. As described in Japanese Laid-Open Patent Publication No. 2002-95300, the terminal voltage of the motor is obtained from the d- and q-axis current control units, and the d-axis current command value is calculated by proportional and integral calculation of the deviation between the terminal voltage command value and the terminal voltage. There is a way to compute.

[Patent Document 1]

Japanese Patent Application Laid-Open No. 8-182398

[Patent Document 2]

Japanese Patent Application Laid-Open No. 2002-95300

However, in the method described in Japanese Patent Application Laid-Open No. 8-182398, since current control is a proportional operation method, the torque accuracy is deteriorated because current does not occur according to the current command value, and the method described in Japanese Patent Laid-Open No. 2002-95300. In the d-axis current command, the torque response tends to deteriorate.

An object of the present invention is to provide a "field weakening device for permanent magnet synchronous motors" which can realize "high precision and high response torque control" even in a weak field control area.

According to the present invention, the d-axis current command value can be generated with a high response by automatically correcting the integral control gain of the weakened field command calculation unit for creating the d-axis current command value by the frequency command value of the motor.

The d-axis current command generated at no load is calculated in advance and added to the output of the calculation unit for preparing the d-axis current command value.

EMBODIMENT OF THE INVENTION Hereinafter, the Example of this invention is described in detail using drawing.

Example 1

Figure 1 shows an example of the configuration of a weakened field vector control device of a permanent magnet synchronous motor of an embodiment of the present invention. 1 is a permanent magnet synchronous motor, 2 is a power converter that outputs a voltage proportional to the voltage command values Vu *, Vv *, Vw * of a 3-phase AC, 21 is a DC power supply, 3 is a 3-phase AC current (Iu, Iv, A current detector capable of detecting Iw), 4 is a magnetic pole position detector capable of detecting a position detection value θi for every 60 degrees of electric angle of the motor, and 5 is a frequency command value ω ₁ from the position detection value θi. Frequency calculating section for calculating *), 6 is a phase calculating section for calculating the rotational phase command (θc *) of the motor from the position detection value (θi) and the frequency command value (ω ₁ *), and 7 is the three-phase alternating current (Iu). Is a coordinate conversion unit for outputting the current detection values Idc and Iqc on the d-axis and q-axis from the detected values Iuc, Ivc, Iwc and the rotational phase command θc *. The weakened field command calculation unit 9 calculates the first d-axis current command value Id * from a deviation between the output voltage command value V ₁ * _ref and the output voltage value V ₁ * in FIG. The d-axis current command outputting the second d-axis current command value Id ** according to the deviation between the first d-axis current command value Id * and the d-axis current detection value Idc, which are outputs of the Kin field command calculation unit. The calculating unit 10 is a q-axis current command calculating unit that outputs the second q-axis current command value Iq ** according to the deviation between the first q-axis current command value Iq * and the q-axis current detection value Iqc. Is a voltage vector for calculating the voltage command values Vd * and Vq * based on the electric constant of the motor 1, the second current command values Id ** and Iq ** and the frequency command value ω ₁ *. The calculating unit 12 is an output voltage calculating unit for calculating the output voltage value V ₁ * of the power converter from the voltage command values Vd * and Vq *, and 13 is the voltage command values Vd * and Vq * and the rotation phase command ( It is a coordinate conversion unit that outputs voltage command values (Vu *, Vv *, Vw *) of three-phase AC from θc *).

First, the basic operations of voltage control and phase control in the vector control method in the case of using the weakened field command operation, which is a feature of the present invention, will be described.

In the voltage control, the output voltage calculating unit 12 in FIG. 1 uses the voltage command values Vd * and Vq * on the d-axis and q-axis as shown in Equation 1 to output voltage values V ₁ *. Is computed.

In which the field-control calculation (8) weakens the output voltage value (V ₁ *) is the output voltage command value at which the field region of weakening (V ₁ * _ref), to match the claim 1 d-axis current command value ( Calculate Id *).

In addition, the voltage vector calculation unit 11 calculates the voltage command values Vd * and Vq * on the d-axis and q-axis by using the current command values and motor constants of the second d-axis and q-axis, which are previously shown in equation (2). To control the converter output voltage.

Here, R ₁ * is a set value of the resistor, Ld * is a set value of the d-axis inductance, Lq * is a set value of the q-axis inductance, and Ke * is a set value of the induced voltage constant.

On the other hand, in the phase control, the magnetic pole position detector 4 can grasp the magnetic pole position for every 60 degrees of electric angle. In this embodiment, the position detection value θi at this time is

Here, i = 0, 1, 2, 3, 4, 5.

In the frequency calculating section 5, the average rotation frequency ω ₁ * (hereinafter referred to as frequency command value) in the shortest 60-degree section is calculated from the position detection value θ i.

Here, Δθ = θ i-θ (i-1), and Δt is the time until the position detection signal in the 60 degree section is detected.

In addition, the phase calculating unit 6 controls the reference phase of the motor 1 by calculating the rotational phase command θc * as shown in Equation 5 using the position detection value θi and the frequency command ω ₁ *. .

The above is the basic operation of voltage control and phase control.

Next, the field command calculation unit 8, which is weakened by the feedback control method, which is a feature of the present invention, will be described with reference to FIG.

In the weakened field command calculation section 8, an integral calculation section having a constant of the integral gain K in which the deviation between the output voltage command value V ₁ * ref and the output voltage value V ₁ * in the weakened field region is constant. It is inputted to 81, and integral operation is performed. The operation value is input to the limiter calculation unit 82 which limits the + side to "zero", and the output value becomes the first d-axis current command Id *.

Next, the working effect brought about by the present invention will be described by the present embodiment.

In the control device of Fig. 1, the case where the first d-axis current command value Id * is controlled to "zero" is considered (when the weakened field command calculation is not performed).

When V ₁ * outputted from the voltage vector calculating unit 11 substitutes Equation 2 into Equation 1,

Further, if the saturation value of V ₁ * V ₁ * _max la, in the voltage saturation range is the relationship of Equation (7).

Here, by arranging Equation 7, a quadratic equation for the frequency command (ω ₁ *) can be obtained.

Here,

From Equation 8, ω ₁ * when V ₁ * is saturated can be obtained.

Here, Fig. 3 shows the relationship between the motor torque τ and the frequency command ω ₁ * in the case where Id ** = Id * = 0 and Iq ** = tau / KT.

Is the motor torque and KT is the torque coefficient.

The solid line shown in FIG. 3 is a boundary line in which V ₁ * is saturated, and the upper side of the boundary line is in a saturated region, and the lower side is in an unsaturated region, and becomes a range that can be actually operated.

For this reason, in vector control in which the d-axis current command value Id * is set to "zero", there is a problem that the operating range in the high speed region is limited low.

In this embodiment, the output voltage value (V ₁ *) this, and calculates the first 1 d-axis current command value (Id *) to match the output voltage command value (V ₁ * _ref) in which the field area of weakening Using this Id *, the second d-axis current command value Id ** is created to calculate the voltage vector.

Here, the output voltage command value V ₁ * _ref of the weakened field region is set as in Equation (10).

As a result, the voltage command values Vd * and Vq * are calculated by the voltage vector calculating unit 11 so that the output voltage value V ₁ * is not saturated (it becomes smaller than V ₁ * _max ). The operating range in the area can be expanded.

According to the present invention, since the current can be generated according to the current command value, high-precision torque control can be realized, and the operating range can be expanded as shown in FIG.

In addition, when high torque is required in the torque control operation, a large current suitable for the torque needs to flow. When high torque is required for a continuous time, the winding resistance value R inside the motor increases with time due to heat generation by the motor current. As a result, since the resistance set value calculated by the voltage vector calculating unit does not coincide with the actual resistance value, the voltage required for the motor cannot be supplied. As a result, a current required for generating torque does not flow, resulting in a lack of torque.

Therefore, by having the current command calculating section upstream of the vector calculating section as shown in FIG. 1 of the present embodiment, the output voltage is controlled to match the electric current of the motor with the current command value. It is possible to provide an AC motor control device that is not affected and does not cause torque shortage from the low speed region.

(Example 2)

5 shows another embodiment of the present invention.

This embodiment is a control device for a permanent magnet synchronous motor in which the integral gain of the field command calculating section weakened by the feedback control method is changed to the frequency command (ω ₁ *).

In FIG. 5, 1-7, 9-13, and 21 are the same as FIG. 8a is a weakened field command calculation unit that automatically corrects the integral gain when integrating the deviation between V ₁ * _ref and V ₁ * according to the frequency command (ω ₁ *).

Next, the weakened field command calculation unit 8a, which is a feature of the present invention, will be described with reference to FIG.

In the weakened field command calculation unit 8a, an integral calculation unit having a constant of the integral gain K has a deviation between the output voltage command value V ₁ * _ref and the output voltage value V ₁ in the weakened field region. 8a1), the integral operation is performed. At that time, the integral gain K is automatically corrected by the frequency ω ₁ . The output value of the integral calculating section 8a1 is input to the limiter calculating section 8a2 which restricts the + side to "zero", and the output value becomes the first d-axis current command Id *.

The second current command value Id ** is created using this current command value Id *, and the voltage output values Vd * and Vq * are calculated to control the converter output voltage.

Here, the effect of the present invention will be described.

If the integral gain (K) used in the weakened field command operation is constant, the closed loop transfer function [G _φ (S)] from V ₁ * _ref to Id * at no load (Iq * = 0) is

Where s is the Laplus operator. It can be seen from Equation 11 that Id * occurs at the first order delay, the response time constant T _φ becomes Equation 12, and T _φ is changed by the frequency command ω ₁ *.

Therefore, the integral gain K of 8a1 is computed as shown by Formula (13).

Here, ω c is the control response angular frequency (rad / s) of the field command operation weakened. Then the new transfer function [G _φ '(s)] is

Becomes Here, the new response time constant (T _φ ') is

to be. From this, T _φ 'can be set irrespective of the frequency command (ω ₁ *), and the effect of higher response can be obtained.

In addition, the control device of the permanent magnet synchronous motor of the method of changing the integral gain of the field command calculating section weakened by the feedback control method of the present embodiment to the frequency command (ω ₁ *) is an upstream section of the voltage vector calculating section as shown in FIG. The present invention can also be applied to control systems other than the control system having a current command calculating section.

(Example 3)

7 shows another embodiment of the present invention. This embodiment is a weakened field vector control device for a permanent magnet synchronous motor when the feedforward method is used in the weakened field command calculation unit.

In FIG. 7, 1-7, 9-13, and 21 of a component are the same as that of FIG.

Referring to Fig. 8, the weakened field command calculation unit 8b by the feedforward control method, which is a feature of the present invention, will be described.

Except for the present embodiment, the d-axis current command generated at no load is calculated in advance by calculation.

In the weakened field command calculating section 8b, the calculating section 8b1 subtracts the induced voltage command value (= ω ₁ * Ke *) from the output voltage command value V ₁ * ref in the weakening field region. Then, the subtracted value is multiplied by the multiplication value of ω ₁ * and Ld *. The output value of the calculating part 8b1 is input to the 1st order delay filter 8b2. Moreover, the output value of 8b2 is input to the limiter calculating part 8b2 which limits the + side to "zero", and the output value becomes a 1st d-axis current command Id *.

Using this current command value Id *, a second current command value Id ** is created, and voltage command values Vd * and Vq * are calculated to control the converter output voltage.

In the high speed region, even when the torque is "zero", V ₁ * is saturated only with the induced voltage command value (= ω ₁ * Ke *) of Vq *.

If the d-axis current command value required to exit the voltage saturation region is Id * _ff0 ,

As a result, by setting the first delay filter time constant T of 8b2 as in Equation 17, the same effects as those in the second embodiment can be obtained in the feedforward control method.

The weakened field vector control device of the permanent magnet synchronous motor when the feedforward method is used in the weakened field command calculation unit of the present embodiment has a control system other than a control system having a current command calculating unit upstream of the voltage vector calculating unit as shown in FIG. It is possible to apply also in.

(Example 4)

9 shows another embodiment of the present invention. This embodiment is a control device for a permanent magnet synchronous motor when the feed forward control method and the feedback control method are used in the weakened field command calculation unit.

In FIG. 9, 1-7, 9-13, and 21 of a component are the same as that of FIG. 10, the field command calculation unit 8c, which is weakened by the feedforward control method and the feedback control method, which is a feature of the present invention, will be described.

In the weakened field command calculating section 8c, the calculating section 8c1 subtracts the induced voltage command value (= ω ₁ * Ke *) from the output voltage command value V ₁ * _ref in the weakening field region. Then, the subtraction is performed by multiplying the subtracted value by ω ₁ * and Ld *.

The output value of the calculating section 8c1 is input to the primary delay filter 8c2. Moreover, the output value of 8c2 is input into the limiter calculation part 8c3 which restricts the + side to "zero", and the output value becomes Id * _ff .

At the same time, the output voltage command value V ₁ * _ref and the output voltage value V ₁ are input to the integral calculation unit 8c4 having an integer of the integral gain K, and the integral operation is performed. At that time, the integral gain K is automatically corrected by the frequency ω ₁ .

The output value of the integral calculation unit 8c4 is input to the limiter calculation unit 8c5 which restricts the + side to "zero", and the output value becomes Id * _fb .

Therefore, as shown in equation (18), the first d-axis current command Id * is calculated based on the sum of the output value Id * _ff of the feedforward control and the output value Id * _fb of the feedback control.

In this manner as well, the operation is performed in the same manner as in the above embodiment, and the effect of higher response can be obtained.

Similarly, the control device of the permanent magnet synchronous motor in the case where the feed forward control method and the feedback control method are used in the weakened field command operation unit of the present embodiment is other than the control system having a current command operation unit upstream of the voltage vector operation unit as shown in FIG. It is also applicable to the control system.

Example 5

In the first to fourth embodiments, the three-phase alternating currents (Iu to Iw) detected by the expensive current detector 3 were detected. However, the present invention can also be applied to a control device that performs inexpensive current detection.

This embodiment is shown in FIG. In FIG. 11, components 1, 2, 4-7, 8a, 9-13, and 21 are the same as that of FIG.

14 is a current estimation unit for estimating the three-phase alternating currents Iu, Iv, and Iw flowing in the motor 1 from the direct current current IDC flowing in the input bus of the power converter.

Using the estimated current values Iu ^, Iv ^, Iw ^, the coordinate detection unit 7 calculates the current detection values Idc, Iqc on the d-axis and q-axis.

In this current sensorless control system, since Id *, Idc, Iq *, and Iqc coincide with each other, it is obvious that the same effects can be obtained by operating in the same manner as in the above embodiment.

In addition, although the method of FIG. 6 is used in the field command calculation part which was weakened in this embodiment, the same effect can be acquired even if the method of FIG. 2, FIG. 8, and FIG. 10 is used.

Example 6

12 shows another embodiment of the present invention.

This embodiment is applied to a control device in which current detection is omitted by performing inexpensive current detection.

The components 1, 2, 7, 8a, 9-13, and 21 in FIG. 12 are the same as that of FIG.

6 'is a go phase computing unit for computing the rotational phase reference (θc *) by integrating the frequency command (ω _l *).

14 is a current estimator for estimating the three-phase alternating currents (Iu, Iv, Iw) flowing through the synchronous motor from the direct current (IDC) flowing through the input bus of the power converter.

By using the estimated current values Iu ^, Iv ^, Iw ^, the coordinate detection unit 7 calculates the current detection values Idc, Iqc on the d-axis and q-axis.

15 is a phase error that is a deviation between the rotational phase command θc * and the rotational phase θ of the motor 1 based on the voltage command values Vd * and Vq * and the current detection values Idc and Iqc. It is a phase error calculating section that estimates [Δθc (= θc * θ)].

16 is a frequency estimating unit that calculates ω ₁ ** so that the phase error Δθ c is “zero”. It is clear that the same effect can be obtained by operating in the same manner as in the above embodiment even in this position and current sensorless control method.

In addition, although the method of FIG. 6 is used in the field command calculation part which was weakened in this embodiment, the same effect can be acquired also when using the method of FIG. 2, FIG.

Example 7

An example in which the present invention is applied to a module will be described with reference to FIG. 13. This example shows an embodiment of Example 1. FIG. Here, the frequency calculating section 5, the phase calculating section 6, the coordinate converting section 7, the weakened field command calculating section 8, the d-axis current command calculating section 9, the q-axis current command calculating section 10, and the voltage vector calculating section. (11), the output voltage calculating section 12 and the coordinate converting section 13 are configured by using a one-chip microcomputer. In addition, the one-chip microcomputer and the power converter are in the form of one module configured on the same substrate. The module here means "standardized structural unit" and is composed of removable hardware / software components. In manufacturing, it is preferable that the structure is formed on the same substrate, but is not limited to the same substrate. Rather, it may be configured on a plurality of circuit boards built in the same box body. In the other embodiments, the same configuration can be taken.

As described above, according to the present invention, high accuracy and high response motor torque can be realized even in the weakened field region, and the present invention can be commonly applied to a system that performs inexpensive current detection or a system in which the magnetic pole position detector is omitted. It is possible to provide a field vector control device for weakening the permanent magnet synchronous motor.

According to the present invention, even in the weakened field region, motor torque with high precision and high response can be realized.

Claims (11)

The output voltage value of the power converter driving the permanent magnet synchronous motor is controlled according to the current command value and the frequency command value of the second d-axis and q-axis calculated by the current command value and the current detection value of the first d-axis and q-axis. In the control device of a permanent magnet synchronous motor, A control device for a permanent magnet synchronous motor, characterized in that it has a weakened field command calculating unit that makes an integral calculation value of a deviation between an output voltage command value and the output voltage value the first d-axis current command value. The method of claim 1, The integral operation of the deviation between the output voltage command value and the output voltage value corrects the integral gain in accordance with the frequency command value. In the control device of the permanent magnet synchronous motor for controlling the output voltage value of the power converter for driving the permanent magnet synchronous motor in accordance with the current command value and the frequency command value of the d-axis and q-axis, A weakened field command operation unit that makes the integral operation value of the deviation between the output voltage command value and the output voltage value a d-axis current command value, The integral operation of the deviation between the output voltage command value and the output voltage value is A control device for a permanent magnet motor, characterized in that the integral gain is corrected according to the frequency command value. In the control device of the permanent magnet synchronous motor for controlling the output voltage value of the power converter for driving the permanent magnet synchronous motor in accordance with the current command value and the frequency command value of the d-axis and q-axis, A control device for a permanent magnet synchronous motor, characterized in that the d-axis current command value is calculated from the weakened field voltage output voltage command value, frequency command value, and motor constant. The method of claim 4, wherein The integrated operation value of the deviation between the output voltage command value and the output voltage value and the sum of the output voltage command value, the frequency command value and the value calculated by the motor constant are d-axis current command values. Control device for permanent magnet synchronous motors. The method of claim 5, The integral operation of the deviation between the output voltage command value and the output voltage value corrects the integral gain in accordance with the frequency command value. The method of claim 4, wherein The calculation based on the output voltage command value, the frequency command value and the motor constant subtracts the induced voltage command value of the motor from the output voltage command value, and divides the subtracted value by the multiplication value of the frequency command value and the d-axis inductance. Control device of a permanent magnet synchronous motor, characterized in that. The method of claim 1, The current detection value is a control device for a permanent magnet synchronous motor, characterized in that the electric current is reproduced from the input DC bus current detection value of the power converter. The method according to any one of claims 1 to 8, The frequency command value is calculated such that the deviation between the rotational phase command and the rotational phase of the motor is calculated based on the voltage command values on the d-axis and q-axis, the detected motor current or the reproduced current, and the deviation is zero. Control device of a permanent magnet synchronous motor, characterized in that. A module comprising the control device according to any one of claims 1 to 8 and a power converter for converting direct current into alternating current. A module comprising the control device according to claim 9 and a power converter for converting direct current into alternating current.

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