CN1716758A

CN1716758A - Control device and module for permanent magnet synchronous motor

Info

Publication number: CN1716758A
Application number: CNA2005100822384A
Authority: CN
Inventors: 户张和明; 远藤常博; 白滨秀文; 伊藤佳树; 青柳滋久
Original assignee: Hitachi Ltd
Current assignee: Hitachi Power Semiconductor Device Ltd
Priority date: 2004-07-01
Filing date: 2005-07-01
Publication date: 2006-01-04
Anticipated expiration: 2025-07-01
Also published as: JP4380437B2; KR20060048738A; JP2006020411A; KR100659250B1; CN100365927C

Abstract

To provide the weakening field vector controller of a permanent magnet synchronous motor which realizes a high accuracy and high response motor torque even in a weakening field region and which is applicable commonly even in a system which performs a cheap current detection or a system which omits a pole position detector. The control gain of an operating part for forming a d-axis current command is automatically corrected according to the frequency command value of the motor. Further, the d-axis current command generated at the time of no load is obtained previously by the operation, and added to the output of the operation part for forming the d-axis current command value.

Description

Control device and module for permanent magnet synchronous motor

Technical Field

The invention relates to a vector control mode of a weak magnetic field region of a permanent magnet synchronous motor.

Background

As a conventional technique of a vector control method of a weak magnetic field region, there are known two methods of controlling currents of a d-axis and a q-axis to a proportional operation method by using a d-axis current command value list (table) as described in japanese unexamined patent publication No. 8-182398; and a method of calculating the d-axis current command value by obtaining the terminal voltage of the motor by the d-axis and q-axis current control units and calculating the deviation between the command value of the terminal voltage and the terminal voltage by proportional-integral calculation, as described in japanese unexamined patent application publication No. 2002-95300.

[ patent document 1] Japanese patent application laid-open No. Hei 8-182398

[ patent document 2] Japanese patent application laid-open No. 2002-95300

However, in the method described in japanese unexamined patent publication No. 8-182398, since the current control is a proportional operation method, a current according to the current command value does not occur, and the torque accuracy tends to deteriorate; in the method described in japanese unexamined patent publication No. 2002-95300, the generation of the d-axis current command is slow, and therefore the torque response tends to deteriorate.

The invention aims to provide a 'weak magnetic field control device of a permanent magnet synchronous motor', which can realize 'high-precision and high-response torque control' even in a weak magnetic field control area.

Disclosure of Invention

The present invention can generate a d-axis current command value with high response by automatically correcting the integral control gain of a weak magnetic field command operation unit for generating the d-axis current command value by using a frequency command value of a motor.

Furthermore, the d-axis current command generated at the time of no load is obtained by performing calculation in advance, and is added to the output of the calculation unit that generates the d-axis current command value.

The present invention can realize a high-precision and high-response motor torque even in a weak magnetic field region.

Drawings

Fig. 1 is a configuration diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to an embodiment of the present invention.

Fig. 2 is an explanatory diagram of the weak magnetic field command arithmetic unit 8 in the control device of fig. 1.

Fig. 3 is an example of a voltage saturation characteristic diagram in the case where the weak magnetic field command arithmetic unit 8 is not provided.

Fig. 4 is an example of a voltage saturation characteristic diagram in the case where the weak magnetic field command arithmetic unit 8 is added.

Fig. 5 is a configuration diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention.

Fig. 6 is an explanatory diagram of the weak magnetic field command arithmetic unit 8a in the control device of fig. 5.

Fig. 7 is a configuration diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention.

Fig. 8 is an explanatory diagram of the weak magnetic field command arithmetic unit 8b in the control device of fig. 7.

Fig. 9 is a configuration diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention.

Fig. 10 is an explanatory diagram of the weak magnetic field command calculation unit 8c in the control device of fig. 9.

Fig. 11 is a configuration diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention.

Fig. 12 is a configuration diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention.

Fig. 13 is a structural diagram in the case where the embodiment of the present invention is applied to a module.

In the figure:

1-permanent magnet synchronous motor, 2-power converter, 3-current detector, 4-magnetic pole position detector, 5-frequency calculation unit, 6-phase calculation unit, 7, 13-coordinate conversion unit, 8a, 8b, 8 c-weak magnetic field command calculation unit, 9-d axis current command calculation unit, 10-q axis current command calculation unit, 11-voltage vector calculation unit, 12-output voltage calculation unit, 14-current estimation unit, 15-phase error calculation unit, 21-DC power supply, IDC-input DC bus current detection value, Id^*A first d-axis current command value, Id^**Second d-axis current command value, Iq^*-a first q-axis current command value, Iq^**-a second q-axis current command value, V₁ ^* _ref-an output voltage command value, V, in the field of weak magnetic field₁ ^*-value of output voltage, [ theta ] c^*-rotational phase command, ω₁ ^*-a frequency command.

Detailed Description

Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.

Fig. 1 shows a configuration example of a field weakening vector control device of a permanent magnet synchronous motor as an embodiment of the present invention. 1 is a permanent magnet synchronous motor; 2 is a voltage command value Vu for outputting an alternating current to three phases^*、Vv^*、Vw^*A power converter of proportional voltage; 21 is a direct current power supply; 3 is a current detector capable of detecting three-phase alternating currents Iu, Iv and Iw; a magnetic pole position detector 4 capable of detecting a position detection value θ i of the motor at an electrical angle of 60 °; 5 calculating a frequency command value omega according to the position detection value theta i₁ ^*The frequency calculating section of (1); 6 is a value obtained from the position detection value thetai and the frequency command value omega₁ ^*Calculating a rotational phase command θ c of the motor^*The phase operation unit of (1); 7 are based on the three-phase alternating currents Iu, Iv,Iw detection values Iuc, Ivc, Iwc, and a rotational phase command θ c^*A coordinate conversion unit for outputting the current detection values Idc, Iqc of the d-axis and q-axis; 8 is based on the output voltage command value V in the field of weak magnetic field₁ ^* _refAnd an output voltage value V₁ ^*Calculating a first d-axis current command value Id^*The weak magnetic field command operation unit; reference numeral 9 denotes a d-axis current command value Id based on the first d-axis current command value Id output from the weak magnetic field command operation unit^*And a deviation from the d-axis current detection value Idc to output a second d-axis current command value Id^**The d-axis current command calculation unit of (1); 10 is based on the first q-axis current command value Iq^*And a deviation from the q-axis current detection value Iqc to output a second q-axis current command value Iq^**The q-axis current command operation unit of (1); reference numeral 11 denotes a 2 nd current command value Id according to an electric constant of the motor 1^**、Iq^**And the frequency command value omega₁ ^*Calculating a voltage command value Vd^*、Vq^*A voltage vector calculation unit of (1); 12 is based on the voltage command value Vd^*、Vq^*Calculating the output voltage value V of the power converter₁ ^*An output voltage calculation unit of (1); 13 is based on the voltage command value Vd^*、Vq^*And a rotation phase command thetac^*Outputting a voltage command value Vu of the three-phase alternating current^*、Vv^*、Vw^*The coordinate transformation unit.

First, basic operations of voltage control and phase control in the vector control method when weak magnetic field command operation, which is a feature of the present invention, is used will be described.

For the voltage control, as shown in fig. 1, the output voltage calculation unit 12 in fig. 1 uses the voltage command values Vd on the d-axis and q-axis^*、Vq^*To calculate the output voltage value V₁ ^*。

[ number 1]

A weak magnetic field command calculation part 8 for calculating a first d-axis current command value Id^*To make the above-mentioned output voltage value V₁ ^*And the output voltage instruction value V in the weak magnetic field region₁ ^* _refAnd (5) the consistency is achieved.

The voltage vector calculation unit 11 calculates the voltage command values Vd on the d-axis and q-axis in advance using the motor constants and the current command values on the second d-axis and q-axis shown by the number 2^*、Vq^*And controls the converter output voltage.

[ number 2]

Vd^*＝R₁ ^*Id^**-ω₁ ^*·Lq^*·Iq^**

Vq^*＝R₁ ^*·Iq^**+ω₁ ^*·Ld^*·Id^**+ω₁ ^*·Ke^* …(2)

Here, R1^*Is a set value of resistance, Ld^*Is the set value of d-axis inductance, Lq^*Is the set value of the q-axis inductance, Ke^*Is the set value of the induced voltage constant.

On the other hand, for the phase control, the magnetic pole position at every 60 degrees of electrical angle can be grasped in the magnetic pole position detector 4. In the present embodiment, the position detection value θ i at this time is

[ number 3]

θi＝60i+30 …(3)

Wherein i is 0, 1, 2, 3, 4, 5. In the frequency calculation part 5, the position detection value is used as the basisθ i, calculating the average rotation frequency ω in the shortest 60 degree interval₁ ^*(hereinafter referred to as a frequency command value).

[ number 4]

ω₁ ^*＝Δθ/Δt …(4)

Here, Δ θ is θ i — θ (i-1), and Δ t is the time until the position detection signal in the 60-degree interval is detected.

The phase calculation unit 6 uses the position detection value θ i and the frequency command ω₁ ^*Calculating the rotation phase command θ c as in FIG. 5^*The reference phase of the motor 1 is controlled.

[ number 5]

θ^*＝θi+ω₁ ^*·Δt …(5)

The above is the basic operation of the voltage control and the phase control.

Next, the weak magnetic field command calculation unit 8 in the feedback control system, which is a feature of the present invention, will be described with reference to fig. 2.

In the weak magnetic field command operation unit 8, the output voltage command value V in the weak magnetic field region₁ ^* _refAnd an output voltage value V₁ ^*The deviation (2) is inputted to an integration operation unit 81 having an integration gain of a constant K, and is integrated. The calculated value is input to the limiter calculation unit 82 whose positive side is limited to zero, and the output value thereof is the first d-axis current command Id^*。

Next, the operation and effect of the present invention will be described by the present example.

The control device of FIG. 1 considers the first d-axis current command value Id^*The control is "zero" (the weak magnetic field command operation is not performed).

V output from voltage vector operation unit 11₁ ^*Substituting the number 2 into the number 1 to obtain：

[ number 6]

In addition, if V is set₁ ^*Has a saturation value of V₁ ^* _maxThe voltage saturation region forms a relationship of a number 7.

[ number 7]

V₁ ^* _max ²＝(R1^*Id^**-ω₁ ^*·Lq^*·Iq^**)²+(R1^*·Iq^**+ω₁ ^*·Ld^*·Id^**+ω₁ ^*·Ke^*)² …(7)

Here, by sorting the number 7, the frequency command ω can be obtained₁ ^*The second order equation of (a) is,

[ number 8]

Wherein,

A＝(Ld^*·Id^**)²+(Lq^*·Iq^**)²+(Ke^*+2·Ld^*·Id^**)

B＝2·R1^*·Iq^**·(Ke^*+(Ld^*-Iq^**)·Id^**)

from the number 8, V can be obtained₁ ^*Omega at saturation₁ ^*。

[ number 9]

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Here, when Id is set^**＝Id^*＝0、Iq^**Motor torque tau and frequency command omega at tau/KT₁ ^*The relationship of (c) is shown in fig. 3.

Here, τ is a motor torque, and KT is a torque coefficient.

The solid line shown in FIG. 3 is V₁ ^*The boundary of saturation is a region in which the upper side is saturated and the lower side is unsaturated, and is a range in which the operation can be performed practically.

Therefore, the d-axis current command value Id is set^*The vector control set to "zero" has a problem in that the operating range in the high speed region is limited to be low.

Therefore, in this embodiment, the output voltage value V is₁ ^*And the output voltage instruction value V in the weak magnetic field region₁ ^* _refIn a consistent manner, a first d-axis current command value Id is calculated^*And using the Id^*Generating a second d-axis current command value Id^**Then, the voltage vector is calculated.

Here, the output voltage command value V of the weak magnetic field region₁ ^* _refAnd is set as 10.

[ number 10]

V₁ ^* _ref＜V₁ ^* _max …(10)

As a result, the voltage vector operation part 11 outputs the voltage value V₁ ^*Unsaturated (to less than V)₁ ^* _maxValue of (d), the voltage command value Vd is calculated^*、Vq^*Therefore, the operating range in the high speed region can be expanded.

According to the present invention, since a current can be generated in accordance with a current command value, torque control with high accuracy can be realized, and the operating range can be expanded as shown in fig. 4.

Further, when a high torque is required in the torque control operation, a large current corresponding to the torque needs to be applied. If high torque is required for a continuous period of time, heat is generated by the motor current, causing the resistance R of the windings inside the motor to increase with time. In this way, the resistance set value calculated by the voltage vector calculation unit does not match the actual resistance value, and thus a necessary voltage cannot be supplied to the motor.

Therefore, by providing the current command calculation unit in the upstream portion of the vector calculation unit and controlling the output voltage so that the motor current matches the current command value as in fig. 1 of the present embodiment, it is possible to provide a control device for an ac motor in which torque shortage does not occur from a low speed region without being affected by a change in the motor constant or an attachment error of a hall element or the like.

[ example 2]

Fig. 5 shows another embodiment of the present invention.

In this embodiment, the frequency command ω is used₁ ^*Method for changing integral gain of weak magnetic field command operation part in feedback control modeA control device for a permanent magnet synchronous motor of the formula.

In FIG. 5, 1 to 7, 9 to 13, 21 are the same as those in FIG. 1. Reference numeral 8a denotes a weak magnetic field command calculation unit which calculates a frequency command ω from the frequency command ω₁ ^*Automatically correct to V₁ ^* _refAnd V₁ ^*The deviation of (2) is an integral gain in the integral operation.

Next, the weak magnetic field command calculation unit 8a, which is a feature of the present invention, will be described with reference to fig. 6.

In the weak magnetic field command operation unit 8a, the output voltage command value V in the weak magnetic field region₁ ^* _refAnd an output voltage value V₁ ^*The deviation (c) is inputted to the integration operation unit 8a1 having an integration gain of constant K, and is integrated. At this time, the integral gain K is derived from the frequency ω₁Is automatically corrected. The output value of the integral computation unit 8a1 is input to the limiter computation unit 82a that limits the positive side to "zero", and the output value thereof is the first d-axis current command Id^*。

Using the current command value Id^*Generating a second current command value Id^**Calculating a voltage command value Vd^*、Vq^*And controlling the output voltage of the converter.

The present embodiment will now be described with reference to the accompanying drawings.

When the integral gain K used for the weak magnetic field instruction operation is a certain time, the time from no load (Iq)^*0) V₁ ^* _refTo Id^*Closed loop transfer function G_Ф(s) is

[ number 11]

Here, s is the laplace operator. According to the number 11, Id^*Generated by a time delay, with a response time constant T_ФNumber 12, known as T_ФAccording to the frequency command omega₁ ^*And (3) varied.

[ number 12]

Therefore, the integral gain K of 8a1 is calculated as shown in fig. 13.

[ number 13]

K＝1/ω₁ ^*·ω_c/Ld^* …(13)

Wherein, ω is_cIs the control response angular frequency (rad/s) of the low field command operation. Thus, the new transfer function G_Ф'(s) become

[ number 14]

Wherein the new response time constant T_Ф' is

[ number 15]

T_Ф′＝1/ω_c …(15)

Thus, T_Ф' can be associated with the frequency command omega₁ ^*Independently set, an effect of higher response can be obtained.

In addition, the frequency command ω₁ ^*The control device of the permanent magnet synchronous motor of the method of changing the integral gain of the low-field command calculation unit in the feedback control method according to the present embodiment can be applied to a control system other than the control system having the current command calculation unit in the upstream portion of the voltage vector calculation unit shown in fig. 5.

[ example 3]

Fig. 7 shows another embodiment of the present invention. In the present embodiment, the field weakening vector control device of the permanent magnet synchronous motor is used in the case where the feed-forward method is used in the field weakening command calculation unit.

In FIG. 7, the constituent elements 1 to 7, 9 to 13, 21 are the same as those in FIG. 1.

The weak magnetic field command calculation unit 8b in the feedforward control method, which is a feature of the present invention, will be described with reference to fig. 8.

In addition to the present example, the d-axis current command generated in the no-load state is obtained by performing calculation in advance.

In the weak magnetic field command calculation unit 8b, the calculation unit 8b1 outputs the voltage command value V from the weak magnetic field region₁ ^* _refThe induced voltage command value (═ ω) is subtracted₁ ^*Ke^*) And dividing the subtracted value by ω₁ ^*And Ld^*The product of (a). The output value of the arithmetic unit 8b1 is input to the first-order delay filter 8b 2. The output value of 8b2 is input to the limiter arithmetic unit 8b3 that limits the positive side to zero, and the output value thereof is the first d-axis current command Id^*。

Using the current command value Id^*Generating a second current command value Id^**And calculates a voltage command value Vd^*、Vq^*To control the converter output voltage.

In the high speed region, even if the torque is "zero", only the torque is limited by Vq^*Induced voltage command value (═ ω)₁ ^*Ke^*)，V₁ ^*It will also saturate.

Id is a d-axis current command value necessary for the operation out of the voltage saturation region^* _ff0Then, then

[ number 16]

Id^* _ff0＝(V₁ ^*max-ω₁ ^*·Ke^*)/(ω₁ ^*·Ld^*) …(16)

By setting the time constant T of the primary delay filter of 8b2 as the number (17) in this way, the same effect as in example 2 can be obtained even in the feedforward control method.

[ number 17]

T＝1/ω_c …(17)

In the case where the feedforward method is used for the field weakening command calculation unit of the present embodiment, the field weakening torque control device of the permanent magnet synchronous motor can be applied to a control system other than the control system having the current command calculation unit in the upstream portion of the voltage vector calculation unit shown in fig. 7.

[ example 4]

Fig. 9 shows another embodiment of the present invention. In the present embodiment, the present invention is a control device for a permanent magnet synchronous motor in the case where a feedforward control method and a feedback control method are used for a low-field command calculation unit.

In FIG. 9, the constituent elements 1 to 7, 9 to 13, 21 are the same as those in FIG. 1. The weak magnetic field command calculation unit 8c in the feedforward control method and the feedback control method, which are features of the present invention, will be described with reference to fig. 10.

In the weak magnetic field command calculation unit 8c, the calculation unit 8c1 outputs the voltage command value V from the weak magnetic field region₁ ^* _refThe induced voltage command value (═ ω) is subtracted₁ ^*Ke^*) And dividing the subtracted value by ω₁ ^*And Ld^*The product of (a).

The output value of the arithmetic unit 8c1 is input to the first-order delay filter 8c 2. The output value of 8c2 is input to limiter operation unit 8c3 for limiting the positive side to zero, and the output value is Id^* _ff。

Further, a voltage command value V is outputted₁ ^* _refAnd an output voltage value V₁The signals are simultaneously input to the integral calculation unit 8c4, which has an integral gain of constant K, and integrated. At this time, the integral gain K is controlled by the frequency ω₁And (6) automatically correcting.

The output value of the integral computing unit 8c4 is input to the positive input terminalThe limiter calculation unit 8c5 whose side limit is "zero" has an output value Id^* _fb。

Therefore, the output value Id of the feedforward control is adjusted as shown in numeral (18)^* _ffAnd the output value Id of the feedback control^* _fbAdding the obtained values to calculate a first d-axis current command Id^*。

[ number 18]

Id^*＝Id^* _ff+Id^* _fb …(18)

In this embodiment, the same operation as in the above embodiment can be performed, and the effect of higher response can be obtained.

Similarly, the control device of the permanent magnet synchronous motor in the case where the feedforward control method and the feedback control method are used for the weak field command calculation unit according to the present embodiment can be applied to a control system other than the control system having the current command calculation unit in the upstream portion of the voltage vector calculation unit as shown in fig. 9.

[ example 5]

Although embodiments 1 to 4 are a system in which the 3-phase alternating currents Iu to Iw detected by the expensive current detector 3 are detected, they can be applied to a control device that detects inexpensive currents.

Fig. 11 shows this embodiment. In FIG. 11, the

constituent elements

1, 2, 4 to 7, 8a, 9 to 13, 21 are the same as those shown in FIG. 5.

Reference numeral 14 denotes a current estimating unit which estimates three-phase ac currents Iu, Iv, and Iw in the motor 1 from the dc current IDC on the input bus of the power converter.

Using the estimated current values Iu, Iv, Iw, the coordinate conversion unit 7 calculates current detection values Idc, Iqc for the d-axis and q-axis.

In this current-less sensor control method, Id is also individually commanded^*And Idc、Iq^*Since the measured value is equal to Iqc, the same operation and the same effect as those of the above-described embodiment can be obtained.

Although the embodiment of fig. 6 is used in the weak magnetic field command calculation unit in the present embodiment, similar effects can be obtained by using the embodiments of fig. 2, 8, and 10.

[ example 6]

Fig. 12 shows another embodiment of the present invention.

This embodiment is suitable for a control device that performs inexpensive current detection and omits a magnetic pole position detector.

In fig. 12, the

constituent elements

1, 2, 7, 8a, 9 to 13, 21 are the same as those shown in fig. 5.

6' is a phase operation part for outputting a frequency command omega₁ ^*Integral to calculate the rotation phase command thetac^*。

Reference numeral 14 denotes a current estimating unit which estimates three-phase ac currents Iu, Iv, and Iw in the synchronous motor from a dc current IDC on an input bus of the power converter.

Using the estimated current values Iu, Iv, Iw, the current detection values Idc, Iqc of the d-axis and q-axis are calculated in the coordinate conversion unit 7.

Reference numeral 15 denotes a phase error calculation unit which calculates a phase error from the voltage command value Vd^*、Vq^*And current detection values Idc, Iqc are added to estimate the rotation phase command θ c^*Phase error Δ θ c (═ θ c) of deviation from rotational phase θ of motor 1^*-θ)。

Reference numeral 16 denotes a frequency estimation unit which calculates ω so that the phase error Δ θ c becomes zero₁ ^**. The same operation as in the above-described embodiment can be performed in the position-less and current-sensor control method, and the same effects can be obtained.

[ example 7]

An example in which the present invention is applied to a module will be described with reference to fig. 13. This example shows an embodiment of example 1. Here, the frequency calculation unit 5, the phase calculation unit 6, the coordinate conversion unit 7, the weak magnetic field command calculation unit 8, the d-axis current command calculation unit 9, the q-axis current command calculation unit 10, the voltage vector calculation unit 11, the output voltage calculation unit 12, and the coordinate conversion unit 13 are configured by a single chip microcomputer. The single chip and the power converter are housed in a single module formed on the same substrate. Here, the module means a "normalized unit of construction" and is composed of separable hardware/software components. In addition, although it is preferable to be formed on the same substrate in terms of manufacturing, the present invention is not limited to the same substrate. Therefore, the present invention can be configured to a plurality of circuit boards incorporated in the same housing. In other embodiments, the same configuration may be adopted.

As described above, the present invention can provide a low-field vector control device for a permanent magnet synchronous motor, which can realize a high-precision and high-response motor torque even in a low-field region; further, the present invention can be used in a system for detecting a current at low cost and a system for omitting a magnetic pole position detector.

Claims

1. A control device for a permanent magnet synchronous motor, which controls an output voltage value of a power converter for driving the permanent magnet synchronous motor according to a frequency command value and current command values of a second d-axis and a q-axis calculated from current command values and current detection values of a first d-axis and a q-axis, characterized in that:

the d-axis current control device includes a weak magnetic field command calculation unit configured to set an integral calculation value of a deviation between an output voltage command value and the output voltage value as the first d-axis current command value.

2. The control device of a permanent magnet synchronous motor according to claim 1,

the integral operation of the deviation between the output voltage command value and the output voltage value corrects the integral gain according to the frequency command value.

3. A control device for a permanent magnet synchronous motor, which controls the output voltage value of a power converter for driving the permanent magnet synchronous motor according to d-axis and q-axis current command values and a frequency command value, characterized in that:

the device comprises a weak magnetic field command calculation unit for calculating an integral calculation value of a deviation between an output voltage command value and the output voltage value as a d-axis current command value,

and integrating the deviation between the output voltage command value and the output voltage value, and correcting the integral gain according to the frequency command value.

4. A control device for a permanent magnet synchronous motor, which controls the output voltage value of a power converter for driving the permanent magnet synchronous motor according to d-axis and q-axis current command values and a frequency command value,

a d-axis current command value is calculated using the output voltage command value and the frequency command value of the field weakening region and a motor constant.

5. The control device of a permanent magnet synchronous motor according to claim 4,

the d-axis current command value is defined as a sum of an integral value of a deviation between the output voltage command value and the output voltage value and a value calculated using the output voltage command value, the frequency command value, and a motor constant.

6. The control device of a permanent magnet synchronous motor according to claim 5,

7. The control device of a permanent magnet synchronous motor according to claim 4,

the output voltage command value, the frequency command value, and the motor constant are used to calculate, the induced voltage command value of the electric motor is subtracted from the output voltage command value, and the subtracted value is divided by the product of the frequency command value and the d-axis inductance.

8. The control device of a permanent magnet synchronous motor according to claim 1,

the current detection value is a current for reproducing a motor current based on a detected value of an input dc bus current of the power converter.

9. The control device of a permanent magnet synchronous motor according to any one of claims 1 or 8,

the deviation between the rotational phase command and the rotational phase of the motor is calculated using the voltage command values of the d-axis and the q-axis and the detected motor current or the reproduced current, and the frequency command value is calculated so that the deviation is zero.

10. A module, comprising:

the control device of any one of claims 1 to 9; and,

and a power converter for converting the direct current into the alternating current.