CN100365927C - Control device and module for permanent magnet synchronous motor - Google Patents
Control device and module for permanent magnet synchronous motor Download PDFInfo
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- CN100365927C CN100365927C CNB2005100822384A CN200510082238A CN100365927C CN 100365927 C CN100365927 C CN 100365927C CN B2005100822384 A CNB2005100822384 A CN B2005100822384A CN 200510082238 A CN200510082238 A CN 200510082238A CN 100365927 C CN100365927 C CN 100365927C
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
- H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
- H02P25/024—Synchronous motors controlled by supply frequency
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
- H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
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- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Control Of Ac Motors In General (AREA)
- Control Of Motors That Do Not Use Commutators (AREA)
Abstract
To provide the weakening field vector controller of a permanent magnet synchronous motor which realizes a high accuracy and high response motor torque even in a weakening field region and which is applicable commonly even in a system which performs a cheap current detection or a system which omits a pole position detector. The control gain of an operating part for forming a d-axis current command is automatically corrected according to the frequency command value of the motor. Further, the d-axis current command generated at the time of no load is obtained previously by the operation, and added to the output of the operation part for forming the d-axis current command value.
Description
Technical Field
The present invention relates to a vector control method for a weak magnetic region of a permanent magnet synchronous motor.
Background
As a conventional technique of a vector control method of a weak magnetic region, there are known two methods of controlling currents of a d-axis and a q-axis to a proportional operation method by using a d-axis current command value list (table) as described in japanese unexamined patent publication No. 8-182398; and a method of calculating the d-axis current command value by obtaining the terminal voltage of the motor by the d-axis and q-axis current control units and calculating the deviation between the command value of the terminal voltage and the terminal voltage by proportional-integral calculation, as described in japanese unexamined patent application publication No. 2002-95300.
[ patent document 1] Japanese patent application laid-open No. Hei 8-182398
[ patent document 2] Japanese patent application laid-open No. 2002-95300
However, in the method described in japanese unexamined patent publication No. 8-182398, since the current control is a proportional operation method, a current according to the current command value does not occur, and the torque accuracy tends to deteriorate; in contrast, in the method described in japanese patent application laid-open No. 2002-95300, the generation of the d-axis current command is slow, and therefore, the torque response tends to deteriorate.
Disclosure of Invention
The invention aims to provide a flux weakening control device of a permanent magnet synchronous motor, which can realize high-precision and high-response torque control even in a flux weakening control area.
A control device for a permanent magnet synchronous motor controls an output voltage value of a power converter for driving the permanent magnet synchronous motor according to a second d-axis current command value, a second q-axis current command value, and a frequency command value, which are calculated from a first d-axis current command value, a first q-axis current command value, and a current detection value, by a voltage vector calculation unit.
In another control device for a permanent magnet synchronous motor according to the present invention, an output voltage value of a power converter for driving the permanent magnet synchronous motor is controlled by a voltage vector operation unit in accordance with current command values of d-axis and q-axis and a frequency command value, and the control device includes a field weakening command operation unit for performing an integral operation of a deviation between an output voltage command value in a field weakening region and the output voltage value as the d-axis current command value and correcting an integral gain based on the frequency command value by performing the integral operation of the deviation between the output voltage command value and the output voltage value.
In still another control device for a permanent magnet synchronous motor according to the present invention, the output voltage value of the power converter for driving the permanent magnet synchronous motor is controlled by the voltage vector calculation unit in accordance with the d-axis and q-axis current command values and the frequency command value, and the d-axis current command value is calculated by the voltage vector calculation unit using the output voltage command value and the frequency command value in the field weakening region and the motor constant.
A module of the invention comprises: the control device described above; and a power converter for converting the direct current to the alternating current.
The present invention can realize a high-precision and high-response motor torque even in a weak magnetic region.
Drawings
Fig. 1 is a block diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to an embodiment of the present invention.
Fig. 2 is an explanatory diagram of the field weakening command calculation unit 8 in the control device of fig. 1.
Fig. 3 is an example of a voltage saturation characteristic diagram in the case where the field weakening command calculation unit 8 is not provided.
Fig. 4 is an example of a voltage saturation characteristic diagram in the case where the field weakening command calculation unit 8 is added.
Fig. 5 is a configuration diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention.
Fig. 6 is an explanatory diagram of the field weakening command calculation unit 8a in the control device of fig. 5.
Fig. 7 is a configuration diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention.
Fig. 8 is an explanatory diagram of the field weakening command calculation unit 8b in the control device of fig. 7.
Fig. 9 is a configuration diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention.
Fig. 10 is an explanatory diagram of the field weakening command calculation unit 8c in the control device of fig. 9.
Fig. 11 is a configuration diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention.
Fig. 12 is a configuration diagram showing a field weakening vector control device of a permanent magnet synchronous motor according to another embodiment of the present invention.
Fig. 13 is a structural diagram in the case where the embodiment of the present invention is applied to a module. In the figure:
1-permanent magnet synchronous motor, 2-power converter, 3-current detector, 4-magnetic pole position detector, 5-frequency operation unit, 6-phase operation unit, 7, 13-coordinate conversion unit, 8a, 8b, 8 c-weak magnetic instruction operation unit, 9-d axis current instruction operation unit, 10-q axis current instruction operation unit, 11-voltage vector operation unit, 12-output voltage operation unit, 14-current push operation unitA determination unit 15-a phase error calculation unit 21-a DC power supply, IDC-an input DC bus current detection value, id * A first d-axis current command value, id ** -second d-axis current command value, iq * -a first q-axis current command value, iq ** -a second q-axis current command value, V 1 * ref -a command value of output voltage in weak magnetic region, V 1 * -value of output voltage, [ theta ] c * -rotational phase command, ω 1 * -a frequency command.
Detailed Description
Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.
Fig. 1 shows a configuration example of a field weakening vector control device of a permanent magnet synchronous motor as an embodiment of the present invention. 1 is a permanent magnet synchronous motor; 2 is a voltage command value Vu for outputting an alternating current to three phases * 、Vv * 、Vw * A proportional voltage power converter; 21 is a direct current power supply; 3 is a current detector capable of detecting three-phase alternating currents Iu, iv and Iw; a magnetic pole position detector capable of detecting a position detection value θ i of the motor per 60 ° electrical angle; 5 calculating a frequency command value omega according to the position detection value theta i 1 * The frequency calculating section of (1); 6 is a value obtained from the position detection value thetai and the frequency command value omega 1 * Calculating a rotational phase command θ c of the motor * The phase operation unit of (1); reference numeral 7 denotes detected values Iuc, ivc, iwc of the three-phase alternating currents Iu, iv, iw and a rotation phase command θ c * A coordinate conversion unit for outputting the current detection values Idc, iqc of the d-axis and q-axis; 8 is based on the output voltage command value V in the weak magnetic region 1 * ref And an output voltage value V 1 * Calculating a first d-axis current command value Id * The field weakening instruction operation unit; reference numeral 9 denotes a d-axis current command value Id based on the first d-axis current command value Id output from the flux weakening command calculation unit * And a deviation from the d-axis current detection value Idc, and outputs a secondd-axis current fingerOrder value Id ** The d-axis current command calculation unit of (1); 10 is based on the first q-axis current command value Iq * And a deviation from the q-axis current detection value Iqc to output a second q-axis current command value Iq ** The q-axis current command operation unit of (1); reference numeral 11 denotes a 2 nd current command value Id according to an electric constant of the motor 1 ** 、Iq ** And the frequency command value omega 1 * Calculating a voltage command value Vd * 、Vq * A voltage vector operation unit of (2); 12 is based on the voltage command value Vd * 、Vq * Calculating the output voltage value V of the power converter 1 * An output voltage calculating section of (1); 13 is based on the voltage command value Vd * 、Vq * And a rotation phase command thetac * Outputting a voltage command value Vu of the three-phase alternating current * 、Vv * 、Vw * The coordinate transformation unit.
First, basic operations of voltage control and phase control in the vector control method when the field weakening command operation, which is a characteristic of the present invention, is used will be described.
For the voltage control, as shown in fig. 1, the output voltage calculation unit 12 in fig. 1 uses the voltage command values Vd on the d-axis and q-axis * 、Vq * To calculate the output voltage value V 1 * 。
[ number 1]
A weak magnetic command calculation part 8 for calculating a first d-axis current command value Id * To make the above-mentioned output voltage value V 1 * And the output voltage command value V in the weak magnetic region 1 * ref And (5) the consistency is achieved.
The voltage vector calculation unit 11 calculates the voltage command values Vd on the d-axis and q-axis in advance using the motor constants and the current command values on the second d-axis and q-axis shown by the number 2 * 、Vq * And controls the converter output voltage.
[ number 2]
Vd * ＝R 1 * Id ** -ω 1 * ·Lq * ·Iq **
Vq * ＝R 1 * ·Iq ** +ω 1 * ·Ld * ·Id ** +ω 1 * ·Ke * ...(2)
Here, R1 * Is a set value of resistance, ld * Is the set value of d-axis inductance, lq * Is the set value of the q-axis inductance, ke * Is the set value of the induced voltage constant.
On the other hand, for the phase control, the magnetic pole position can be grasped every 60 degrees of electrical angle in the magnetic pole position detector 4. In the present embodiment, the position detection value θ i at this time is
[ number 3]
θi＝60i+30...(3)
Wherein i =0, 1,2, 3, 4, 5.
The frequency calculation unit 5 calculates an average rotational frequency ω within the shortest 60-degree interval from the position detection value θ i 1 * (hereinafter referred to as a frequency command value).
[ number 4]
ω 1 * ＝Δθ/Δt...(4)
Here, Δ θ = θ i- θ (i-1), and Δ t is a time until the position detection signal of the 60-degree section is detected.
The phase calculation unit 6 uses the position detection value θ i and the frequency command ω 1 * Calculating the rotation phase command θ c as in FIG. 5 * The reference phase of the motor 1 is controlled.
[ number 5]
θ * ＝θi+ω 1 * ·Δt ...(5)
The above is the basic operation of the voltage control and the phase control.
Next, the field weakening command operation unit 8 in the feedback control method, which is a feature of the present invention, will be described with reference to fig. 2.
In the weak magnetic instruction operation part 8, the output voltage instruction value V in the weak magnetic region 1 * ref And an output voltage value V 1 * The deviation (2) is inputted to an integration operation unit 81 having an integration gain of a constant K, and integrated. The calculated value is input to the limiter calculation unit 82 whose positive side is limited to zero, and the output value thereof is the first d-axis current command Id * 。
Next, the operation and effects of the present invention will be described with reference to the present example.
The control device of FIG. 1 considers the first d-axis current command value Id * The control is "zero" (the weak magnetic instruction operation is not performed).
V output from the voltage vector operation unit 11 1 * Substituting the number 2 into the number 1 yields:
[ number 6]
In addition, if V is set 1 * Has a saturation value of V 1 * max The voltage saturation region forms a relationship of a number 7.
[ number 7]
V 1 * max 2 ＝(R1 * Id ** -ω 1 * ·Lq * ·Iq ** ) 2 +(R1 * ·Iq ** +ω 1 * ·Ld * ·Id ** +ω 1 * ·Ke * ) 2 ...(7)
Here, by sorting the number 7, the frequency command ω can be obtained 1 * The second order equation of (a) is,
[ number 8]
A·ω 1 *2 +B·ω 1 * +C＝0...(8)
Wherein, the first and the second end of the pipe are connected with each other,
A＝(Ld * ·Id ** ) 2 +(Lq * ·Iq ** ) 2 +(Ke * +2·Ld * ·Id ** )
B＝2·R1 * ·Iq ** ·(Ke * +(Ld * -Iq ** )·Id ** )
C＝R1 *2 ·(Id **2 +Iq **2 )-V 1 * max 2
from the number 8, V can be obtained 1 * Omega at saturation 1 * 。
[ number 9]
Here, when Id is set ** ＝Id * ＝0、Iq ** Motor torque tau and frequency command omega when t/KT 1 * The relationship of (c) is shown in fig. 3.
Here, τ is a motor torque, and KT is a torque coefficient.
The solid line shown in FIG. 3 is V 1 * The boundary of saturation is a region in which the upper side is saturated and the lower side is unsaturated, and is a range in which the operation can be performed practically.
Therefore, the d-axis current command value Id is set * The vector control set to "zero" has a problem that the operating range in the high speed region is limited to be low.
Therefore, in this embodiment, the output voltage value V is 1 * With the output voltage command value V in the weak magnetic region 1 * ref In a consistent manner, a first d-axis current command value Id is calculated * And using the Id * Generating a second d-axis current command value Id ** The voltage vector is calculated.
In this case, the weak magnetic regionOutput voltage command value V 1 * ref And is set as 10.
[ number 10]
V 1 * ref ＜V 1 * max ...(10)
As a result, the voltage vector operation part 11 outputs the voltage value V 1 * Unsaturated (to less than V) 1 * max Value (d) of the voltage command value Vd is calculated * 、Vq * Therefore, the operating range in the high speed region can be expanded.
Since the present invention can generate a current in accordance with a current command value, it is possible to realize a high-precision torque control and to expand an operation range as shown in fig. 4.
In addition, when a high torque is required during the torque control operation, a large current corresponding to the torque needs to be applied. If high torque is required for a continuous period of time, heat is generated by the motor current, causing the resistance R of the windings inside the motor to increase with time. In this way, the resistance set value calculated by the voltage vector calculation unit does not match the actual resistance value, and thus a necessary voltage cannot be supplied to the motor.
Therefore, by providing the current command calculation unit in the upstream portion of the vector calculation unit and controlling the output voltage so that the motor current matches the current command value as in fig. 1 of the present embodiment, it is possible to provide a control device for an alternating current motor in which torque shortage does not occur from a low speed region without being affected by a change in the motor constant or an attachment error of a hall element or the like.
[ example 2]
Fig. 5 shows another embodiment of the present invention.
In this embodiment, the frequency command ω is used 1 * Method for changing integral gain of weak magnetic instruction operation part in feedback control modeAnd a control device for a permanent magnet synchronous motor.
In FIG. 5, 1 to 7,9 to 13, and 21 are the same as those in FIG. 1. 8a is a weak magnetic instruction arithmetic unit based on the frequency instruction omega 1 * Automatically correct for V 1 * ref And V 1 * The deviation of (2) is an integral gain in the integral operation.
Next, the field weakening command calculation unit 8a, which is a feature of the present invention, will be described with reference to fig. 6.
In the weak magnetic command operation part 8a, the output voltage command value V in the weak magnetic region 1 * ref And an output voltage value V 1 * The deviation (b) is input to the integration operation unit 8a1 having an integration gain of constant K, and is integrated. At this time, the integral gain K is derived from the frequency ω 1 Is automatically corrected. The output value of the integration operation unit ga1 is input to the limiter operation unit 82a that limits the positive side to zero, and is the first d-axis current command Id * 。
Using the current command value Id * Generating a second current command value Id ** Calculating a voltage command value Vd * 、Vq * And controlling the output voltage of the converter.
The present embodiment is described with reference to the effects of the present invention.
When the integral gain K used by the weak magnetic instruction operation is a certain time, the time is from no load (Iq) * V of = 0) 1 * ref To Id * Closed loop transfer function G Φ (s) is
[ number 11]
Here, s is the laplace operator. According to the number 11,Id * Generated by a delay, the response time constant T of which Ф Number 12, known as T Φ According to the frequency command omega 1 * And (3) varied.
[ number 12]
Therefore, the integral gain K of 8a1 is calculated as shown in fig. 13.
[ number 13]
K＝1/ω 1 * ·ω c /Ld * ...(13)
Wherein, ω is c Is the control response angular frequency (rad/s) of the flux weakening command operation. Thus, the new transfer function G Φ '(s) become
[ number 14]
Wherein the new response time constant T Φ ' is
[ number 15]
T Φ ′＝1/ω c ...(15)
Thus, T Φ ' can be associated with the frequency command omega 1 * Independently set, a higher response effect can be obtained.
In addition, the frequency command ω 1 * The control device of the permanent magnet synchronous motor of the method of changing the integral gain of the field weakening command calculation unit in the feedback control method according to the present embodiment can be applied to a control system other than the control system having the current command calculation unit in the upstream portion of the voltage vector calculation unit shown in fig. 5.
[ example 3]
Fig. 7 shows another embodiment of the present invention. The present embodiment is a field weakening vector control device for a permanent magnet synchronous motor in which a feed-forward method is used in a field weakening command calculation unit.
In fig. 7, the constituent elements 1 to 7,9 to 13, and 21 are the same as those in fig. 1.
The field weakening command calculation unit 8b in the feedforward control system, which is a characteristic of the present invention, will be described with reference to fig. 8.
In addition to the present example, the d-axis current command generated in the no-load state is obtained by performing calculation in advance.
In the weak magnetic command calculation unit 8b, the calculation unit 8b1 outputs the voltage command value V from the weak magnetic region 1 * ref Minus the induction voltage command value (= ω) 1 * Ke * ) And dividing the subtracted value by ω 1 * And Ld * The product of (a) and (b). The output value of the computing unit 8b1 is input to the first-order delay filter8b2. The output value of 8b2 is input to the limiter arithmetic unit 8b3 that limits the positive side to zero, and the output value thereof is the first d-axis current command Id * 。
Using the current command value Id * Generating a second current command value Id ** And calculates a voltage command value Vd * 、Vq * To control the converter output voltage.
In the high speed region, even if the torque is "zero", only the torque is limited by Vq * Induced voltage command value (= ω) 1 * Ke * )，V 1 * It will also saturate.
Id is a d-axis current command value necessary for the operation out of the voltage saturation region * ff0 Then, then
[ number 16]
Id * ff0 ＝(V 1 * max -ω 1 * ·Ke * )/(ω 1 * ·Ld * ) ...(16)
In this way, by setting the time constant T of the primary delay filter of 8b2 as in numerical value (17), the same effect as in example 2 can be obtained even in the feedforward control method.
[ number 17]
T＝1/ω c ...(17)
In the case where the feedforward method is used for the field weakening command calculation unit of the present embodiment, the field weakening torque control device of the permanent magnet synchronous motor can be applied to a control system other than the control system having the current command calculation unit in the upstream portion of the voltage vector calculation unit shown in fig. 7.
[ example 4]
Fig. 9 shows another embodiment of the present invention. In the present embodiment, the present invention is a control device for a permanent magnet synchronous motor in a case where a feedforward control method and a feedback control method are used for a field weakening command calculation unit.
In fig. 9, components 1 to 7,9 to 13, and 21 are the same as those in fig. 1. The field weakening command calculation unit 8c in the feedforward control method and the feedback control method, which are the features of the present invention, will be described with reference to fig. 10.
In the weak magnetic command calculation unit 8c, the calculation unit 8c1 outputs the voltage command value V from the weak magnetic region 1 * ref Subtracting the induction voltage command value (= ω) from the induction voltage command value 1 * Ke * ) And dividing the subtracted value by ω 1 * And Ld * The product of (a) and (b).
The output value of the arithmetic unit 8c1 is input to the first-order delay filter 8c2. The output value of 8c2 is input to the limiter calculation part 8c3 for limiting the positive side to zero, and the output value is Id * ff 。
Further, a voltage command value V is outputted 1 * ref And an output voltage value V 1 The signals are simultaneously input to the integral calculation unit 8c4 having an integral gain of constant K, and are subjected to integral calculation. At this time, the integral gain K is controlled by the frequency ω 1 And (6) automatically correcting.
The output value of the integral operation unit 8c4 is input to the limiter operation unit 8c5 that limits the positive side to zero, and the output value is Id * fb 。
Thus, by the following equation (18)Shows the output value Id of the feedforward control * ff And the output value Id of the feedback control * fb Adding the first d-axis current command Id to calculate a first d-axis current command Id * 。
[ number 18]
Id * ＝Id * ff +Id * fb ...(18)
In this embodiment, the same operation as in the above embodiment can be performed, and the effect of higher response can be obtained.
Similarly, the control device of the permanent magnet synchronous motor in the case where the feedforward control method and the feedback control method are used for the field weakening command calculation unit according to the present embodiment can be applied to a control system other than the control system having the current command calculation unit in the upstream portion of the voltage vector calculation unit as shown in fig. 9.
[ example 5]
Although embodiments 1 to 4 are a system in which the 3-phase alternating currents Iu to Iw detected by the expensive current detector 3 are detected, they can be applied to a control device that detects inexpensive currents.
Fig. 11 shows this embodiment. In FIG. 11, the constituent elements 1,2,4 to 7,8a,9 to 13, and 21 are the same as those shown in FIG. 5.
Using the estimated current values Iu, iv, iw, the coordinate conversion unit 7 calculates current detection values Idc, iqc for the d-axis and q-axis.
In this current-less sensor control method, id is also individually commanded * And Idc, iq * Since the measured value is equal to Iqc, the same operation and the same effect as those of the above-described embodiment can be obtained.
Although the embodiment of fig. 6 is used in the field weakening command calculation unit in the present embodiment, similar effects can be obtained by using the embodiments of fig. 2, 8, and 10.
[ example 6]
Fig. 12 shows another embodiment of the present invention.
This embodiment is suitable for a control device that performs inexpensive current detection and omits a magnetic pole position detector.
In FIG. 12, the constituent elements 1,2,7,8a,9 to 13, and 21 are the same as those shown in FIG. 5.
6' is a phase operation part for instructing the frequency ω 1 * Integral to calculate the rotation phase command thetac * 。
Using the estimated current values Iu, iv, iw, the current detection values Idc, iqc of the d-axis and q-axis are calculated in the coordinate conversion unit 7.
Reference numeral 15 denotes a phase error calculation unit which calculates a phase error from the voltage command value Vd * 、Vq * And current detection values Idc and Iqc are added to estimate the rotation phase command θ c * Phase error Δ θ c (= θ c) of deviation from rotational phase θ of motor 1 * -θ)。
Reference numeral 16 denotes a frequency estimation unit which calculates ω so that the phase error Δ θ c becomes zero 1 ** . The same operation as in the above-described embodiment can be performed in the position-less and current-sensor control method, and the same effects can be obtained.
Although the embodiment of fig. 6 is used in the field weakening command calculation unit in the present embodiment, similar effects can be obtained by using the embodiments of fig. 2, 8, and 10.
[ example 7]
An example in which the present invention is applied to a module will be described with reference to fig. 13. This example shows an embodiment of example 1. Here, the frequency calculation unit 5, the phase calculation unit 6, the coordinate conversion unit 7, the field weakening instruction calculation unit 8, the d-axis current instruction calculation unit 9, the q-axis current instruction calculation unit 10, the voltage vector calculation unit 11, the output voltage calculation unit 12, and the coordinate conversion unit 13 are configured by a single chip microcomputer. The single chip and the power converter are housed in a single module formed on the same substrate. Here, the module means a "normalized unit of construction" and is composed of separable hardware/software components. Although it is preferable to form the substrate on the same substrate in terms of manufacturing, the substrate is not limited to the same substrate. Therefore, the present invention can be configured to a plurality of circuit boards incorporated in the same housing. In other embodiments, the same configuration may be adopted.
As described above, the present invention can provide a field weakening vector control device for a permanent magnet synchronous motor, which can realize a high-precision and high-response motor torque even in a field weakening region; further, the present invention can be used in a system for detecting a current at low cost and a system for omitting a magnetic pole position detector.
Claims (11)
1. A control device for a permanent magnet synchronous motor, wherein a voltage vector calculation unit controls an output voltage value of a power converter for driving the permanent magnet synchronous motor in accordance with a second d-axis current command value, a second q-axis current command value, and a frequency command value, which are calculated from a first d-axis current command value, a first q-axis current command value, and a current detection value, the control device being characterized in that:
the d-axis current control device includes a flux-weakening command calculation unit configured to set an integral calculation value of a deviation between an output voltage command value and the output voltage value in a flux-weakening region as the first d-axis current command value.
2. The control device of a permanent magnet synchronous motor according to claim 1,
and performing integral operation of the deviation between the output voltage command value and the output voltage value, and correcting integral gain according to the frequency command value.
3. A control device for a permanent magnet synchronous motor, which controls an output voltage value of a power converter for driving the permanent magnet synchronous motor according to current command values of a d-axis and a q-axis and a frequency command value by a voltage vector calculation unit, characterized in that:
the device comprises a weak magnetic instruction calculation unit for calculating an integral of a deviation between an output voltage instruction value and the output voltage value in a weak magnetic region as a d-axis current instruction value,
and integrating the deviation of the output voltage command value and the output voltage value, and correcting the integral gain according to the frequency command value.
4. A control device for a permanent magnet synchronous motor, which controls an output voltage value of a power converter for driving the permanent magnet synchronous motor according to current command values of d-axis and q-axis and a frequency command value by a voltage vector operation unit,
a d-axis current command value is calculated by a voltage vector calculation unit using the output voltage command value and the frequency command value of the weak magnetic region and a motor constant.
5. The control device of a permanent magnet synchronous motor according to claim 4,
the d-axis current command value is defined as a sum of an integral value of a deviation between the output voltage command value and the output voltage value and a value calculated using the output voltage command value, the frequency command value, and a motor constant.
6. The control device of a permanent magnet synchronous motor according to claim 5,
and integrating the deviation of the output voltage command value and the output voltage value, and correcting the integral gain according to the frequency command value.
7. The control device of a permanent magnet synchronous motor according to claim 4,
and subtracting the induced voltage command value of the electric motor from the output voltage command value by calculation using the output voltage command value, the frequency command value and the motor constant, and dividing the subtracted value by the product of the frequency command value and the d-axis inductance.
8. The control device of a permanent magnet synchronous motor according to claim 1,
the current detection value is a current for reproducing the motor current based on the input dc bus current detection value of the power converter.
9. The control device of a permanent magnet synchronous motor according to any one of claims 1 or 8,
the deviation between the rotational phase command and the rotational phase of the motor is calculated using the voltage command values of the d-axis and the q-axis and the detected or estimated motor current, and the frequency command value is calculated so that the deviation is zero.
10. A module, comprising:
the control device according to any one of claims 1 to 8; and the combination of (a) and (b),
and a power converter for converting the direct current into the alternating current.
11. A module, comprising:
the control device of claim 9; and the combination of (a) and (b),
and a power converter for converting the direct current into the alternating current.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2004195111 | 2004-07-01 | ||
| JP2004195111A JP4380437B2 (en) | 2004-07-01 | 2004-07-01 | Control device and module for permanent magnet synchronous motor |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN1716758A CN1716758A (en) | 2006-01-04 |
| CN100365927C true CN100365927C (en) | 2008-01-30 |
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Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CNB2005100822384A Expired - Fee Related CN100365927C (en) | 2004-07-01 | 2005-07-01 | Control device and module for permanent magnet synchronous motor |
Country Status (3)
| Country | Link |
|---|---|
| JP (1) | JP4380437B2 (en) |
| KR (1) | KR100659250B1 (en) |
| CN (1) | CN100365927C (en) |
Families Citing this family (18)
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|---|---|---|---|---|
| JP4634321B2 (en) * | 2006-02-28 | 2011-02-16 | 日立オートモティブシステムズ株式会社 | Control device for electric four-wheel drive vehicle |
| KR101053315B1 (en) * | 2006-09-26 | 2011-08-01 | 미쓰비시덴키 가부시키가이샤 | Vector control device of permanent magnet synchronous motor |
| CN100413207C (en) * | 2006-11-17 | 2008-08-20 | 清华大学 | Vector control system of alien frequencies power supplied permanent magnet synchronous motor |
| JP4654217B2 (en) * | 2007-04-25 | 2011-03-16 | 日立オートモティブシステムズ株式会社 | Field weakening control device for permanent magnet motor and electric power steering using the same |
| JP4194645B1 (en) * | 2008-03-28 | 2008-12-10 | 日立アプライアンス株式会社 | Refrigeration equipment |
| JP5576039B2 (en) * | 2008-12-17 | 2014-08-20 | 株式会社日立産機システム | Synchronous motor controller |
| KR101522285B1 (en) * | 2009-01-16 | 2015-05-21 | 삼성테크윈 주식회사 | Apparatus for controlling motor |
| CN103718451B (en) * | 2011-08-10 | 2016-06-15 | 松下电器产业株式会社 | The control device of electric motor |
| JP5509167B2 (en) | 2011-09-08 | 2014-06-04 | 株式会社日立産機システム | Synchronous motor control system |
| JP5575176B2 (en) * | 2012-04-23 | 2014-08-20 | 三菱電機株式会社 | Control device for rotating electrical machine |
| JP5633551B2 (en) * | 2012-11-05 | 2014-12-03 | 株式会社安川電機 | AC motor control device |
| CN105024604B (en) * | 2014-04-28 | 2017-09-29 | 青岛海信日立空调系统有限公司 | A kind of method and a device for controlling weak magnetism of permagnetic synchronous motor |
| CN105337548A (en) * | 2015-11-23 | 2016-02-17 | 南京金龙新能源汽车研究院有限公司 | Permanent magnet synchronous motor controller based on vector control and control method |
| US10008967B2 (en) * | 2016-10-25 | 2018-06-26 | Microchip Technology Inc. | Closed loop flux weakening for permanent magnet synchronous motors |
| CN107086544B (en) * | 2017-05-10 | 2019-03-08 | 广东美的暖通设备有限公司 | Overvoltage crowbar, three-phase drive motor and air conditioner |
| CN111404437A (en) * | 2020-03-17 | 2020-07-10 | 美的集团股份有限公司 | Boost-buck driving method and device, air conditioner and computer readable storage medium |
| WO2021234934A1 (en) | 2020-05-22 | 2021-11-25 | 三菱電機株式会社 | Control device for permanent magnet synchronous motor, and electric power steering device equipped with said control device |
| US20230402946A1 (en) * | 2020-12-17 | 2023-12-14 | Mitsubishi Electric Corporation | Motor driving apparatus, refrigeration cycle equipment, and air conditioner |
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| JPH08182398A (en) * | 1994-12-27 | 1996-07-12 | Fuji Electric Co Ltd | Driving device for permanent magnet-type synchronous motor |
| JP2002095300A (en) * | 2000-09-19 | 2002-03-29 | Meidensha Corp | Method of controlling permanent magnet synchronous motor |
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| JPH05284771A (en) * | 1992-03-31 | 1993-10-29 | Meidensha Corp | Automatic tuning method for pi controller |
| JP3366854B2 (en) | 1997-03-27 | 2003-01-14 | 松下電器産業株式会社 | Motor control device |
| JP2000245199A (en) | 1999-02-24 | 2000-09-08 | Matsushita Electric Ind Co Ltd | Motor controller |
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- 2004-07-01 JP JP2004195111A patent/JP4380437B2/en not_active Expired - Lifetime
-
2005
- 2005-06-30 KR KR1020050057564A patent/KR100659250B1/en active IP Right Grant
- 2005-07-01 CN CNB2005100822384A patent/CN100365927C/en not_active Expired - Fee Related
Patent Citations (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPH08182398A (en) * | 1994-12-27 | 1996-07-12 | Fuji Electric Co Ltd | Driving device for permanent magnet-type synchronous motor |
| JP2002095300A (en) * | 2000-09-19 | 2002-03-29 | Meidensha Corp | Method of controlling permanent magnet synchronous motor |
Also Published As
| Publication number | Publication date |
|---|---|
| KR100659250B1 (en) | 2006-12-19 |
| JP2006020411A (en) | 2006-01-19 |
| KR20060048738A (en) | 2006-05-18 |
| JP4380437B2 (en) | 2009-12-09 |
| CN1716758A (en) | 2006-01-04 |
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