JP2009095135A - Controller of synchronous electric motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To drive a permanent magnet synchronous electric motor with variable speed by position sensorless vector control while responsiveness of speed estimation with respect to speed fluctuation is improved. <P>SOLUTION: An adaptive current observer 110 estimates armature current based on a model of an embedded permanent magnet synchronous electric motor 101. An inverse transfer function matrix calculation means 111 separates a deflection of current in γδ axes into a component proportional to an angle error and a component proportional to a speed error based on an inverse matrix of a transfer function to the deflection of a detected value of the γδ axes of armature current and an estimation value, and outputs an angle estimation error Δθ<SB>#</SB>and a speed estimation error Δω<SB>#</SB>. An angle/speed/primary resistance estimator 112 calculates a speed estimation value ω<SB>r#</SB>=ω<SB>1</SB>based on the angle estimation error Δθ<SB>#</SB>and the speed estimation error Δω<SB>#</SB>, which are outputted from the inverse transfer function matrix calculation means 111. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a control apparatus for a synchronous motor, and is particularly suitable for application to a control method for driving a permanent magnet synchronous motor at a variable speed by position sensorless vector control.

  Permanent magnet synchronous motors have the advantage of being smaller and more efficient than induction motors, and are applied to transportation fields such as electric vehicles and railway vehicles as well as elevators. In particular, an electric motor drive device in a transportation field such as a railway vehicle is required to be capable of variable speed operation and torque control in a wide range from a very low speed to a weak field operation range. Also, in the field of railway vehicles, it is required that the estimated value follows the fluctuation of the wheel speed at high speed. As such a permanent magnet synchronous motor control method, Non-Patent Document 1 discloses a method of realizing permanent magnet synchronous motor speed sensorless vector control by using an observer for estimating magnetic flux.

FIG. 4 is a block diagram showing a schematic configuration of a conventional synchronous motor control device.
In FIG. 4, the embedded permanent magnet synchronous motor 001 is connected to a synchronous motor control device 000 that drives the embedded permanent magnet synchronous motor 001 at a variable speed, and is connected to the rotating shaft of the embedded permanent magnet synchronous motor 001. Is connected to a load 002.
Here, the synchronous motor control device 000 includes a power conversion device 004, a rotating two-phase / three-phase coordinate converting unit 005, a three-phase / rotating two-phase coordinate converting unit 006, a current control unit 007, and a current command value creating unit 008. , A speed PID adjuster 009, an armature current magnetic flux observer 010, an angle error calculating means 011, a speed / angle calculating means 012 and subtractors 013, 014, 015 are provided on the output side of the synchronous motor control device 000. Current detecting means 003 for detecting the UVW phase currents i _u , i _v , i _w supplied to the embedded permanent magnet synchronous motor 001 is provided.

The subtractor 013 can calculate a deviation between the speed command value ω ^* given to the control device 000 for the synchronous motor and the estimated speed value ω _{r #} output from the speed / angle calculation unit 012.
Speed PID controller 009, the deviation between the output velocity command value omega ^* and the speed estimated value omega _{r #} subtractor 013 so that the zero velocity command value omega ^* and the speed estimated value omega _{r #} and the The torque command value T ^* can be calculated by performing the PID calculation of the deviation.

The current command value creating means 008 can calculate the current command values iγ ^* and iδ ^* of the γδ axis based on the torque command value T ^* output from the speed PID adjuster 009. The axis estimated as the direction parallel to the magnetic pole of the magnet of the embedded permanent magnet synchronous motor 001 was taken as the γ axis, and the direction perpendicular to the γ axis was taken as the δ axis.
The three-phase / rotation two-phase coordinate conversion unit 006 performs a speed / angle calculation after the two-phase conversion of the detected values of the UVW phase currents i _u , i _v , i _w detected by the current detection unit 003 on the αβ axis. By performing rotational coordinate conversion of the estimated angle value θ _# output from the means 012, current detection values iγ and iδ on the γδ axis can be calculated.

The subtracters 014 and 015 deviate between the current command values iγ ^* and iδ ^* output from the current command value creation unit 008 and the detected current values iγ and iδ output from the three-phase / rotation two-phase coordinate conversion unit 006. Can be calculated respectively.
The current control unit 007 is configured to output a voltage command value vγ ^* on the γδ axis so that a deviation between the current command values iγ ^* and iδ ^* output from the subtracters 014 and 015 and the detected current values iγ and iδ becomes zero. vδ ^* can be calculated.

The rotating two-phase / three-phase coordinate conversion unit 005 performs reverse rotation conversion of the voltage command values vγ ^* and vδ ^* of the γδ axis based on the estimated angle value θ _# output from the speed / angle calculation unit 012 to obtain a fixed two-phase. The voltage command values v _u ^* , v _v ^* , and v _w ^* can be calculated by performing the two-phase three-phase conversion after the conversion to the above value.
The power conversion device 004 drives the embedded permanent magnet synchronous motor 001 based on the voltage command values v _u ^* , v _v ^* , v _w ^* output from the rotating two-phase / three-phase coordinate conversion means 005. The embedded permanent magnet synchronous motor 001 can be controlled at a variable speed.

The armature current magnetic flux observer 010 is based on the voltage command values vγ ^* and vδ ^* output from the current control means 007, and the value uγδ obtained by subtracting the voltage drop at the resistance. A child current magnetic flux (a magnetic flux generated by a current flowing through the stator winding) is estimated, and a deviation between the armature current estimated magnetic flux λγδ _# and the apparent armature current magnetic flux λγδ ^* can be calculated.

The angle error calculation means 011 calculates the angle error Δθ of the magnetic pole position based on the deviation between the armature current estimated magnetic flux λγδ _# output from the armature current magnetic flux observer 010 and the apparent armature current magnetic flux λγδ ^*. Can do.
The speed / angle calculating unit 012 can calculate the estimated speed value ω _{r #} and the estimated angle value θ _# based on the angular error Δθ output from the angular error calculating unit 011.

When the speed command value ω ^* is given to the synchronous motor control device 000, the difference between the speed command value ω ^* and the estimated speed value ω _{r #} output from the speed / angle calculation means 012 is the subtractor 013. And output to the speed PID adjuster 009. Then, the speed PID adjuster 009 performs PID calculation so that the deviation between the speed command value ω ^* and the speed estimated value ω _{r #} becomes zero, and sends the torque command value T ^* to the current command value creating means 008. Outputs the value T ^* . Then, the current command value preparing unit 008 receives the torque command value T ^* from the speed PID controller 009, a current command value i? ^* Based on the torque command value T ^*, calculates i? ^*, A subtracter 014,015 Respectively.

On the other hand, the current detection means 003 detects the UVW phase currents i _u , i _v , i _w supplied to the embedded permanent magnet synchronous motor 001 and sends the detected values to the three-phase / rotation two-phase coordinate conversion means 006. Output. Then, the three-phase / rotation two-phase coordinate conversion means 006 converts the detected values of the UVW phase currents i _u , i _v , i _w into the two-phase conversion of the αβ axis, and then outputs the angle output from the speed / angle calculation means 012 By performing rotational coordinate conversion of the estimated value θ _# , current detection values iγ and iδ are calculated and output to the subtracters 014 and 015, respectively.

Then, the subtracter 014,015, the current command value i? ^*, I? ^* And the current detection value i?, Receives the i?, Respectively, the current command value i? ^*, I? ^* And the current detection value i?, The deviation between the i?, Respectively Calculate and output to the current control means 007. Then, the current control unit 007, current command value i? ^{*,? *} And the current detection value i?, The voltage command value so that the deviation between the i? Is zero v? ^*, Calculates v? ^*, Rotating two-phase / three-phase It outputs to the coordinate conversion means 005.

The rotary two-phase / three-phase coordinate conversion unit 005, the voltage command values v? ^*, Receives the v? ^*, The voltage command value based on the output from the speed and angle calculating means 012 angle estimate theta _# v? ^*, The voltage command values v _u ^* , v _v ^* , and v _w ^* are calculated by performing two-phase three-phase conversion after reverse rotation conversion of vδ ^* and conversion to a fixed two-phase value, and the power conversion device 004 Output to. Then, the power conversion device 004 drives the embedded permanent magnet synchronous motor 001 based on the voltage command values v _u ^* , v _v ^* , v _w ^* to change the embedded permanent magnet synchronous motor 001 to a variable speed. Control.

In addition, the current detection values iγ and iδ calculated by the three-phase / rotation two-phase coordinate conversion unit 006 are output to the armature current magnetic flux observer 010 and the voltage command value vγ ^* calculated by the current control unit 007 ^. , Vδ ^* is output to the armature current magnetic flux observer 010. The armature current flux observer 010 calculates the following equation (1) based on the voltage command values vγ ^* and vδ ^* output from the current control means 007 and the value uγδ obtained by subtracting the voltage drop at the resistor. To estimate the armature current magnetic flux of the embedded permanent magnet synchronous motor 001 and output the deviation between the armature current estimated magnetic flux λγδ _# and the apparent armature current magnetic flux λγδ ^* to the angle error calculating unit 011 To do.

However,
uγδ: value obtained by subtracting the voltage drop due to resistance from the biaxial inverter voltage p: differential operator λγδ _# : biaxial armature current estimation magnetic flux ω ₁ : speed estimated value (= ω _{r #} ) (primary frequency and equal)
Φ _m : Magnet magnetic flux (αI−Jω ₁ ): Feedback gain λγδ ^* of armature current magnetic flux observer 010: Apparent armature current magnetic flux of two axes. In the expression (1), the complex number display is represented by a matrix display. Also,

である。ただし、
Ｌ_d：ｄ軸インダクタンス
Ｌ_q：ｑ軸インダクタンス
Ｒ_S：電機子抵抗
である。

It is. However,
L _d : d-axis inductance L _q : q-axis inductance R _S : armature resistance.

In equation (1), the armature current flux observer 010 receives as input the voltage command values vγ ^* and vδ ^* output from the current control means 007 and a value uγδ obtained by subtracting the voltage drop at the resistor. Magnetic flux λγδ _# can be calculated. In the equation (2), the armature current flux observer 010 calculates an apparent armature current flux λγδ ^* and calculates a deviation (λγδ _# −λγδ ^* ) from the armature current estimated flux λγδ _# as an angular error calculating means. 011 can be output.

Then, upon receiving the deviation (λγδ _# −λγδ ^* ) between the armature current estimated magnetic flux λγδ _# and the apparent armature current magnetic flux λγδ ^* , the angle error calculating means 011 calculates the angle error Δθ of the magnetic pole position, and the velocity Output to angle calculation means 012
Here, when the embedded permanent magnet synchronous motor 001 is in a steady state (when the angle error Δθ is small), the angle error calculation unit 011 uses the γ-axis component of the magnetic flux estimation error to obtain the following (3). The angle error Δθ can be obtained by the equation.

  Further, when the embedded permanent magnet synchronous motor 001 is in the starting state (when the angle error Δθ is large), the angle error calculation unit 011 can obtain the angle error Δθ by the following equation (4).

Upon receiving the angle error Δθ, the speed / angle calculation means 012 calculates the speed estimated value ω _{r #} and the angle estimated value θ _# , outputs the speed estimated value ω _{r #} to the subtractor 013, and also estimates the angle. By outputting the value θ _# to the rotating two-phase / three-phase coordinate converting means 005 and the three-phase / rotating two-phase coordinate converting means 006, an embedded permanent magnet synchronous motor can be used without using a position sensor for detecting the magnetic pole position. 001 can be driven at a variable speed.
Here, the speed / angle calculating means 012 can obtain the speed estimated value ω _{r #} = ω ₁ by the following equation (5).

Further, the speed / angle calculating means 012 can obtain the estimated angle value θ _# by the following equation (6).

As described above, when the speed calculation is performed using the angle error Δθ obtained by the expression (3) or (4), the speed error is obtained in the transfer function from the true value ω _r of the rotation speed to the estimated speed value ω _{r #.} A value obtained by integrating Δω is calculated as a proportional integral.
FIG. 5 is a block diagram showing a transfer function from the true value of the rotational speed of the conventional synchronous motor to the estimated speed value.
In FIG. 5, in the configuration of the transfer function from the true value of the rotational speed to the estimated speed value in the conventional synchronous motor control device 000, a subtractor 016 and blocks B11 to B13 are provided. Here, the transfer function of the block B11 is 1 / s, the transfer function of the block B12 (sK _p + K _i ) / s, and the transfer function of the block B13 is β / (s + β). Here, K _p is a proportional gain, K _i is an integral gain, and β is a time constant.

The true value omega _r and velocity estimates omega _{r #} rotational speed is inputted to the subtracter 016, by the speed estimated value omega _{r #} from the true value omega _r of the rotational speed is subtracted by the subtractor 016, A speed error Δω is calculated and input to the block B11. Then, the angle error Δθ is calculated by integrating the speed error Δω in the block B11. After the proportional-integral calculation is performed in the block B12, the estimated speed value ω _{r #} is output via the block B13.
Satoshi Murakami, Hiroki Kanno, Kenji Kamai, Yoichi Hayashi, Tetsuya Fukumoto: "Design method of permanent magnet synchronous motor speed sensorless vector control system using adaptive observer", IEEJ Semiconductor Power Conversion Study Group, SPC-02-85

However, in the method of calculating the estimated speed value ω _{r #} using the armature current magnetic flux observer 010 in FIG. 4, the angular error Δθ is proportionally integrated to obtain the estimated speed value ω _{r #.} When the true value ω _r of the motor fluctuates, it takes time until the speed estimated value ω _{r #} follows. For this reason, when applied to the field of railway vehicles, there is a problem in that when the wheel slips, the speed may change abruptly, and the idling of such a wheel cannot be detected quickly.
Accordingly, an object of the present invention is to provide a control apparatus for a synchronous motor that can drive a permanent magnet synchronous motor at a variable speed by position sensorless vector control while improving the responsiveness of speed estimation to speed fluctuation. is there.

In order to solve the above-described problem, according to the control apparatus for a synchronous motor according to claim 1, the armature current is estimated based on the current detection means for detecting the armature current of the synchronous motor and the model of the synchronous motor. Based on an adaptive current observer, and an inverse matrix of a transfer function from an angle estimation error of an axis deviation and a speed estimation error of an armature to a deviation between the detected value and the estimated value of the two axes of the armature current. Considering the inverse transfer function matrix calculation means for separating the current deviation of the current into a component proportional to the angle error and a component proportional to the speed error, and a component proportional to the speed error separated by the inverse transfer function matrix calculation means A speed estimator for estimating the rotational speed of the synchronous motor, and a voltage command value calculated by feeding back the speed estimated value estimated by the speed estimator to the speed command value. Characterized by comprising a power converter for variable speed control said synchronous motor.
Further, according to the control device for a synchronous motor according to claim 2, the speed estimator is a value obtained in consideration of both the angle estimation error and the speed estimation error calculated by the inverse transfer function matrix calculation means. The rotational speed of the synchronous motor is estimated by performing a proportional integral calculation.

  As described above, according to the present invention, the inverse matrix of the transfer function is used from the angle estimation error of the axis deviation and the speed estimation error of the armature to the deviation between the detected value and the estimated value of the two-axis armature current. Thus, the component proportional to the angular error and the component proportional to the speed error can be separated and estimated. For this reason, the rotational speed of the synchronous motor can be estimated while considering not only the angle error but also the instantaneous value of the speed error, and the position sensorless vector control is made permanent while improving the speed estimation responsiveness to the speed fluctuation. Since the magnet synchronous motor can be driven at a variable speed, idling of the wheel of the railway vehicle can be detected quickly.

Hereinafter, a control apparatus for a synchronous motor according to an embodiment of the present invention will be described with reference to the drawings.
FIG. 1 is a block diagram showing a schematic configuration of a synchronous motor control device according to a first embodiment of the present invention.
In FIG. 1, a controller 100 for a synchronous motor that drives the embedded permanent magnet synchronous motor 101 at a variable speed is connected to the embedded permanent magnet synchronous motor 101. Is connected to a load 102.

Here, the synchronous motor control device 100 includes a power conversion device 104, a rotating two-phase / three-phase coordinate converting means 105, a three-phase / rotating two-phase coordinate converting means 106, a current control means 107, and a current command value creating means 108. , A speed PID adjuster 109, an adaptive current observer 110, an inverse transfer function matrix calculating means 111, an angle / speed / primary resistance estimator 112, and subtractors 113, 114, 115 are provided on the output side of the synchronous motor control device 100 Are provided with current detection means 103 for detecting the UVW phase currents i _u , i _v , i _w supplied to the embedded permanent magnet synchronous motor 101.

  Note that the power converter 104, the rotating two-phase / three-phase coordinate converting means 105, the three-phase / rotating two-phase coordinate converting means 106, the current control means 107, the current command value creating means 108, the speed PID adjuster 109, and the subtractor 113. , 114, 115 are the power conversion device 004, rotating two-phase / three-phase coordinate converting means 005, three-phase / rotating two-phase coordinate converting means 006, current control means 007, current command value creating means 008, speed PID. The same operation as that of the adjuster 009 and the subtracters 013, 014, and 015 can be performed.

The adaptive current observer 110 can estimate the armature current based on the model of the embedded permanent magnet synchronous motor 101.
The inverse transfer function matrix calculating means 111 calculates the deviation from the angle estimation error Δθ _# of the axis deviation and the speed estimation error Δω _{# of the} armature to the deviation between the current detection values iγ, iδ and the current estimation values iγ _# , iδ _# on the γδ axis. Based on the inverse matrix of the transfer function, the current deviation on the γδ axis is separated into a component proportional to the angle error and a component proportional to the speed error, and an angle estimation error Δθ _# and a speed estimation error Δω _# can be output. it can.

Angle, speed and stator resistance estimator 112 based on the inverse transfer function matrix calculating unit 111 is output from the position estimation error [Delta] [theta] _# and speed estimation error [Delta] [omega _#, velocity estimate ω _{r #} = ω ₁ and the angle estimate θ _# can be calculated.
The detected current values iγ and iδ calculated by the three-phase / rotating two-phase coordinate conversion means 106 are output to the adaptive current observer 110 and the voltage command values vγ ^* and vδ calculated by the current control means 107 are output. ^* Is output to the adaptive current observer 110.

Then, the adaptive current observer 110 uses the state equations relating to the current detection values iγ, iδ on the γδ axis and the current estimation values iγ _# , iδ _# on the γδ axis, thereby detecting the current detection values iγ, iδ on the γδ axis and the γδ axis. Deviations (hereinafter also referred to as current errors) from the current estimated values iγ _# and iδ _# are calculated and output to the inverse transfer function matrix calculating unit 111.
Here, the equation of state regarding the current detection values iγ and iδ on the γδ axis and the current estimation values iγ _# and iδ _# on the γδ axis can be expressed by the following equation (7). In the present embodiment, it is assumed that the method is applied to a speed estimation method of a low speed or higher, and the error of the armature resistance can be ignored.

However,
i γ _# , i δ _# : current estimated value R _{s # of the} γδ axis: armature resistance estimated values g ₁₁ , g ₁₂ , g ₂₁ , g ₂₂ : feedback gain of the adaptive current observer 110. Further, the feedback gain of the adaptive current observer 110 can be given by the following equation (8).

However,
g _c : Control variable that determines the pole of (positive value)
It is.
When the inverse transfer function matrix calculation means 111 receives the deviation between the detected current values iγ and iδ on the γδ axis and the estimated current values iγ _# and iδ _{# on} the γδ axis, the estimated value and the detected value on the γδ axis of the armature current are estimated. Based on the inverse matrix of the transfer function up to the deviation from the value, the angle estimation error Δθ _# and the speed estimation error Δω _# are calculated and output to the angle / speed / primary resistance estimator 112.
Here, the inverse matrix of the transfer function from the angle estimation error Δθ _# and the speed estimation error Δω _# to the deviation between the current detection values iγ, iδ of the γδ axis and the current estimation values iγ _# , iδ _# of the γδ axis is (9).

However, the coefficients B _{11 to} B ₂₂ can be set as in the following equation (10).

Hereinafter, the point that the angle estimation error Δθ _# and the speed estimation error Δω _# can be estimated simultaneously by using the inverse matrix of the transfer function calculated by the inverse transfer function matrix calculating unit 111 will be described.
First, when the γδ axis of the synchronous motor control device 100 has an angle error Δθ from the dq axis of the embedded permanent magnet synchronous motor 001, the state equation of the embedded permanent magnet synchronous motor 001 viewed from the γδ axis is as follows: It can be given by equation (11).

  However, C is a rotational coordinate transformation matrix for converting the amount of the dq axis into the γδ axis, and can be given by the following equation (12). When Δθ> 0, the γδ axis is delayed by Δθ compared to the dq axis of the embedded permanent magnet synchronous motor 001.

  When the equation (7) is subtracted from the equation (11), an error equation can be given by the following equation (13).

Note that F ₁ (x) and F ₂ (x) are 2 × 2 matrices including a trigonometric function with the angle error Δθ as a variable.
Here, in F ₁ (x) and F ₂ (x), they are approximated as cos Δθ = cos ₂ Δθ = 1, sin Δθ = Δθ, and sin 2Δθ = 2Δθ. When Δω = d / dtΔθ and the value of the equation (8) is used as the feedback gain of the adaptive current observer 110, the equation (13) can be rewritten as the following equation (14).

However, the coefficients B _{11 to} B ₂₂ can be set as shown in Equation (10). In this case, the characteristic equation of the feedback gain of the adaptive current observer 110 can be given by the following equation (15), and the adaptive current observer 110 operates stably.

Then, by using the equation (14), as shown in the following equation (16), the deviation between the detected current values iγ, iδ of the γδ axis and the estimated current values iγ _# , iδ _# of the γδ axis, the angle error Δθ And a relationship of speed error Δω = ω _r −ω ₁ can be given.

Then, by substituting the equation (16) into the equation (9), the angle estimation error Δθ _# when the deviation between the current detection values iγ, iδ of the γδ axis and the current estimation values iγ _# , iδ _# of the γδ axis is eliminated, and When the speed estimation error Δω _# is obtained, it can be expressed by the following equation (17), and the output from the angle error Δθ and the speed error Δω to the inverse transfer function can be expressed by a first-order lag system.

When the angle / velocity / primary resistance estimator 112 receives the angle estimation error Δθ _# and the velocity estimation error Δω _# from the inverse transfer function matrix calculation means 111, the following equations (18) and (19) are used. The estimated speed value ω _{r #} = ω ₁ and the estimated angle value θ _# can be calculated.
When the angle / speed / primary resistance estimator 112 calculates the speed estimated value ω _{r #} = ω ₁ and the angle estimated value θ _# , the angle / speed / primary resistance estimator 112 outputs the speed estimated value ω _{r #} to the subtractor 113, and the angle estimated value θ _# Can be output to the rotating two-phase / three-phase coordinate converting means 105 and the three-phase / rotating two-phase coordinate converting means 106.

However,
Kθ _p : proportional gain of speed estimator Kθ _i : integral gain α ₁ and α _{2 of} speed estimator: control variables.
As a result, the inverse transfer function matrix calculation means 111 calculates the current detection values iγ and iδ of the γδ axis and the current estimation values iγ _# and iδ _# from the angle estimation error Δθ _# of the axis deviation and the speed estimation error Δω _{# of the} armature. By using the inverse matrix of the transfer function up to the deviation, the component proportional to the angle error Δθ and the component proportional to the speed error Δω can be separated and estimated. For this reason, the rotational speed of the permanent magnet synchronous motor 101 can be estimated in consideration of not only the angular error Δθ but also the instantaneous value of the speed error Δω, and position sensorless while improving the responsiveness of the speed estimation to the speed fluctuation. Since the permanent magnet synchronous motor 101 can be driven at a variable speed by vector control, the idling of the wheel of the railway vehicle can be detected quickly.

FIG. 2 is a block diagram showing a transfer function from the true value of the rotational speed of the synchronous motor according to the embodiment of the present invention to the estimated speed value.
2, in the configuration of the transfer function from the true value of the rotational speed to the estimated speed value in the synchronous motor control apparatus 100 of FIG. 1, a subtractor 116 and blocks B <b> 1 to B <b> 4 are provided. The blocks B1 to B4 can be provided in the angle / velocity / primary resistance estimator 112 of FIG. Here, the transfer function of the block B1 is α ₁ / s, the transfer function of the block B2 is 1 / (s + g _c ), the transfer function g _c α ₂ of the block B3 is (sK _p + K _i ). / S.

The true value omega _r and velocity estimates omega _{r #} rotational speed is inputted to the subtracter 116, by the speed estimated value omega _{r #} from the true value omega _r of the rotational speed is subtracted by the subtractor 116, A speed estimation error Δω _# is calculated and input to blocks B1 and B2. As the speed estimation error Δω _# input to the blocks B1 and B2, the value calculated by the inverse transfer function matrix calculating unit 111 in FIG. 1 can be used. Then, by integrating the speed estimation error Δω _# in the block B1, an angle estimation error Δθ _# is calculated and input to the adder 117, and the speed estimation error Δω _# is added to the adder via the blocks B2 and B3. The values input to 117 and output from the blocks B1 and B3 are added by the adder 117, and then a proportional-integral calculation is performed in the block B4 to be output as a speed estimated value ωr _# . The estimated speed value ω _{r #} is fed back to the subtractor 116 and also fed back to the subtractor 113 in FIG.

FIG. 3 is a diagram showing a response of the speed estimation system according to the embodiment of the present invention in comparison with a response of the speed estimation system of the conventional example. In FIG. 3, the response is obtained by setting the proportional gain K _p in the configuration of FIG. 2 to 10, the integral gain _Ki to 25, the control variable α ₁ to 3, and the control variable α ₂ to 0.75. In addition, by setting the time constant β in the configuration of FIG. 5 to 1000 and setting the cut-off frequency sufficiently higher than the cut-off frequency determined by the proportional-integral calculation, the first-order filter after the proportional-integral calculation has improved speed estimation performance. The influence was not made.

In FIG. 3, in the configuration of FIG. 5, since the angular error Δθ is simply proportionally integrated to obtain the speed estimated value ω _{r #} , it takes time to follow the speed fluctuation when the speed changes in a ramp shape. On the other hand, in the configuration of FIG. 2, since the speed integral value ω _{r #} is obtained by performing the proportional integration calculation on the value that considers not only the angle error Δθ but also the speed error Δω, even when the speed changes in a ramp shape, It is possible to quickly follow the speed fluctuation.

It is a block diagram which shows schematic structure of the control apparatus of the synchronous motor which concerns on one Embodiment of this invention. It is a block diagram which shows the transfer function from the true value of the rotational speed of the synchronous motor which concerns on one Embodiment of this invention to a speed estimated value. It is a figure which compares the response of the speed estimation system which concerns on one Embodiment of this invention with the response of the speed estimation system of a prior art example. It is a block diagram which shows schematic structure of the control apparatus of the conventional synchronous motor. It is a block diagram which shows the transfer function from the true value of the rotational speed of the conventional synchronous motor to a speed estimated value.

Explanation of symbols

DESCRIPTION OF SYMBOLS 100 Control apparatus of synchronous motor 101 Implantable permanent magnet synchronous motor 102 Load 103 Current detection means 104 Power converter 105 Rotation two phase / three phase coordinate conversion means 1066 Three phase / rotation two phase coordinate conversion means 107 Current control means 108 Current Command value creation means 109 Speed PID regulator 110 Adaptive current observer 111 Inverse transfer function matrix calculation means 112 Angle / speed / primary resistance estimator 113, 114, 115 Subtractor 117 Adder

Claims (2)

Current detection means for detecting the armature current of the synchronous motor;
An adaptive current observer for estimating an armature current based on the model of the synchronous motor;
Based on the inverse matrix of the transfer function from the angle estimation error of the axis deviation and the armature speed estimation error to the deviation between the detected value and the estimated value of the armature current, the deviation of the current of the two axes An inverse transfer function matrix calculating means for separating a component proportional to the error and a component proportional to the speed error;
A speed estimator for estimating the rotational speed of the synchronous motor by considering a component proportional to the speed error separated by the inverse transfer function matrix calculating means;
A control apparatus for a synchronous motor, comprising: a power converter that performs variable speed control of the synchronous motor.   The speed estimator estimates the rotational speed of the synchronous motor by performing a proportional-integral operation on the values obtained in consideration of both the angle estimation error and the speed estimation error calculated by the inverse transfer function matrix calculation means. The control apparatus for a synchronous motor according to claim 1, wherein:

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